MIC2171
100kHz 2.5A Switching Regulator
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (
408
) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
May 2007
1 M9999-051107
General Description
The MIC2171 is a complete 100kHz SMPS current-mode
controller with an internal 65V 2.5A power switch.
Although primarily intended for voltage step-up applica-
tions, the floating switch architecture of the MIC2171
makes it practical for step-down, inverting, and Cuk config-
urations as well as isolated topologies.
Operating from 3V to 40V, the MIC2171 draws only 7mA
of quiescent current, making it attractive for battery
operated supplies.
The MIC2171 is available in a 5-pin TO-220 or TO-263 for
–40°C to +85°C operation.
Data sheets and support documentation can be found on
Micrel’s web site at: www.micrel.com.
Features
• 2.5A, 65V internal switch rating
• 3V to 40V input voltage range
• Current-mode operation, 2.5A peak
• Internal cycle-by-cycle current limit
• Thermal shutdown
• Twice the frequency of the LM2577
• Low external parts count
• Operates in most switching topologies
• 7mA quiescent current (operating)
• Fits LT1171/LM2577 TO-220 and TO-263 sockets
Applications
• Laptop/palmtop computers
• Battery operated equipment
• Hand-held instruments
• Off-line converter up to 50W(requires external power
switch)
• Pre-driver for higher power capability
___________________________________________________________________________________________________________
Typical Application
+5V
(4.75V min.)
D1
1N5822
MIC2171
IN
SW
FB
C1*
47µF
R3
1k C3
1µF
R1
10.7k
1%
C2
470µF
V
OUT
+12V, 0.25
A
L1
15µH
* Locate near MIC2171 when supply leads > 2”
R2
1.24k
1%
GND
COMP
C4
470µF
D2
1N5818
C1
47µF
C2
1µF
R3
1k
R4*
V
OUT
5V, 0.5
A
* Optional voltage clipper (may be req’d if T1 leakage inductance too high)
C3*
D1*
R1
3.74k
1%
R2
1.24k
1%
V
IN
4V to 6V T1
1:1.25
L
PRI
= 12µH
MIC2171
IN
SW
FB
GND
COMP
Figure 1. MIC2171 5V to 12V Boost Co nverter Figure 2. MIC2171 5VFlyback Co nverter
Micrel, Inc. MIC2171
May 2007
2 M9999-051107
Ordering Information
Part Number
Standard RoHS Compliant* Temperature Range Package
MIC2171BT MIC2171WT –40° to +85°C 5-Pin TO-220
MIC2171BU MIC2171WU –40° to +85°C 5-Pin TO-263
*RoHS compliant with "high-melting solder" exemption.
Pin Configur ation
Tab GND
5IN
4SW
3GND
2FB
1COMP
Tab GND
5
I
N
4SW
3GND
2FB
1COMP
5-Pin TO-220 (T) 5-Pin TO-263 (U)
Pin Description
Pin Number Pin Name Pin Function
1 COMP
Frequency Compensation: Output of transconductance-type error amplifier.
Primary function is for loop stabilization. Can also be used for output voltage
soft-start and current limit tailoring.
2 FB
Feedback: Inverting input of error amplifier. Connect to external resistive divider
to set power supply output voltage.
3 GND
Ground: Connect directly to the input filter capacitor for proper operation (see
applications info).
4 SW
Power Switch Collector: Collector of NPN switch. Connect to external inductor
or input voltage depending on circuit topology.
5 IN Supply Voltage: 3.0V to 40V
Micrel, Inc. MIC2171
May 2007
3 M9999-051107
Absolute Maximum Ratings
Supply Voltage (V
IN
).......................................................40V
Switch Voltage (V
SW
)......................................................65V
Feedback Voltage (transient, 1ms) (V
FB
) .....................±15V
Lead Temperature (soldering, 10 sec.)...................... 300°C
Storage Temperature (T
s
) .........................–65°C to +150°C
ESD Rating
(1)
Operating Ratings
Operating Temperature Range ................... –40°C to +85°C
Junction Temperature (T
J
) ........................ –55°C to +150°C
Thermal Resistance
TO-220-5 (θ
JA
)
(2)
...............................................45°C/W
TO-263-5 (θ
JA
)
(3)
................................................45°C/W
Electrical Characteristics
V
IN
= 5V; T
A
= 25°C, bold values indicate –40°C< T
A
< +85°C, unless noted.
Parameter Condition Min Typ Max Units
Reference Section
Feedback Voltage (V
FB
) V
COMP
= 1.24V 1.220
1.214
1.240 1.264
1.274
V
V
Feedback Voltage
Line Regulation
3V V
IN
40V
V
COMP
= 1.24V
0.6 %/V
Feedback Bias Current (I
FB
) V
FB
= 1.24V 310 750
1100
nA
nA
Error Amplifier Section
Transconductance (g
m
) I
COMP
= ±25µA 3.0
2.4
3.9 6.0
7.0
µA/mV
µA/mV
Voltage Gain (A
V
) 0.9V V
COMP
1.4V 400 800 2000 V/V
Output Current V
COMP
= 1.5V 125
100
175 350
400
µA
µA
Output Swing High Clamp, V
FB
= 1V
Low Clamp, V
FB
= 1.5V
1.8
0.25
2.1
0.35
2.3
0.52
V
V
Compensation Pin Threshold Duty Cycle = 0 0.8
0.6
0.9 1.08
1.25
V
V
Output Switch Section
ON Resistance I
SW
= 2A, V
FB
= 0.8V 0.37 0.50
0.55
Current Limit Duty Cycle = 50%, T
J
25°C
Duty Cycle = 50%, T
J
< 25°C
Duty Cycle = 80%, Note 4
2.5
2.5
2.5
3.6
4.0
3.0
5.0
5.5
5.0
A
A
A
Breakdown Voltage (BV) 3V V
IN
40V
I
SW
= 5mA
65 75 V
Oscillator Section
Frequency (f
O
) 88
85
100 112
115
kHz
kHz
Duty Cycle [δ(max)] 80 90 95 %
Input Supply Voltage Section
Minimum Operating Voltage 2.7 3.0 V
Quiescent Current (I
Q
) 3V V
IN
40V, V
COMP
= 0.6V, I
SW
= 0 7 9 mA
Supply Current Increase (I
IN
) I
SW
= 2A, V
COMP
= 1.5V, during switch on-time 9 20 mA
Notes:
1. Devices are ESD sensitive. Handling precautions recommended.
2. Mounted vertically, no external heat sink, 1/4 inch leads soldered to PC board containing approximately 4 inch squared copper area surrounding
leads.
3. All ground leads soldered to approximately 2 inches squared of horizontal PC board copper area.
4. For duty cycles (δ) between 50% and 95%, minimum guaranteed switch current is I
CL
= 1.66 (2-δ) Amp (Pk).
Micrel, Inc. MIC2171
May 2007
4 M9999-051107
Typical Characteristics
2.3
2.4
2.5
2.6
2.7
2.8
2.9
-100 -50 0 50 100 150
Minimum Operating Voltage (V)
Temperature (°C)
Minimum
Operating Voltage
Switch Current = 2A
0
100
200
300
400
500
600
700
800
-100 -50 0 50 100 150
Feedback Bias Current (nA)
Temperature (°C)
Feedback Bias C u r r e n t
-5
-4
-3
-2
-1
0
1
2
3
4
5
0 10203040
Feedback Voltage Change (mV)
V
IN
Operating (V)
Feedback Voltage
Line Regulation
TJ=-40°C
TJ=25°C
TJ= 125°C
5
6
7
8
9
10
11
12
13
14
15
0 10203040
Supply Current (mA)
V
IN
Operating Voltage (V)
Supply Curren t
ISW=0
D.C.= 90%
D.C.= 50%
D.C.= 0%
0
10
20
30
40
50
01234
Average Supply Current (mA)
Switch Current (A)
Supply Current
δ= 90%
δ= 50%
0
1
2
3
4
5
6
7
8
9
10
-100 -50 0 50 100 150
Supply Current (mA)
Temperature(°C)
Supply Curren t
V
COMP
=0.6V
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0123
Switch ON Voltage (V)
Switch Current (A)
TJ=25°C
TJ=125°C
TJ=–40°C
Switch On-Voltage
60
70
80
90
100
110
120
-50 0 50 100 150
Frequency (kHz)
Temperature(°C)
Oscillator Frequency
0
2
4
6
8
0 20406080100
Switch Current (A)
Duty Cycle (%)
Current Limit
–40°C 25°C
125°C
Error Amplifier Gain
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
-100 -50 0 50 100 150
Transconductance (µA/mV)
Temperature(°C)
0
1000
2000
3000
4000
5000
6000
7000
1 10 100 1000 10000
Transconductance (µS)
Frequency (kHz)
Error Amplifier Gain
210
180
150
120
90
60
30
0
-30
1 10 100 1000 10000
Phase Shift (°)
Frequency (kHz)
Error A mplifier Phas e
Micrel, Inc. MIC2171
May 2007
5 M9999-051107
Functional Diagram
Current
Amp.
Error
Amp.
1.24V
Ref.
Logic
Driver
Anti-Sat. SW
100kHz
Osc.
Reg.
FB
COMP GND
IN D1
Q1
2.3V
Com-
parator
Functional Description
Refer to “Block Diagram MIC2171”.
Internal Power
The MIC2171 operates when V
IN
is 2.6V. An internal
2.3V regulator supplies biasing to all internal circuitry
including a precision 1.24V band gap reference.
PWM Operation
The 100kHz oscillator generates a signal with a duty
cycle of approximately 90%. The current-mode
comparator output is used to reduce the duty cycle when
the current amplifier output voltage exceeds the error
amplifier output voltage. The resulting PWM signal
controls a driver which supplies base current to output
transistor Q1.
Current-Mode Advantages
The MIC2171 operates in current mode rather than
voltage mode. There are three distinct advantages to
this technique. Feedback loop compensation is greatly
simplified because inductor current sensing removes a
pole from the closed loop response. Inherent cycle-by-
cycle current limiting greatly improves the power switch
reliability and provides automatic output current limiting.
Finally, current-mode operation provides automatic input
voltage feed forward which prevents instantaneous input
voltage changes from disturbing the output voltage
setting.
Anti-Saturation
The anti-saturation diode (D1) increases the usable duty
cycle range of the MIC2171 by eliminating the base to
collector stored charge which would delay Q1’s turnoff.
Compensation
Loop stability compensation of the MIC2171 can be
accomplished by connecting an appropriate network
from either COMP to circuit ground (see “Typical
Applications”) or COMP to FB.
The error amplifier output (COMP) is also useful for soft
start and current limiting. Because the error amplifier
output is a transconductance type, the output impedance
is relatively high which means the output voltage can be
easily clamped or adjusted externally.
Micrel, Inc. MIC2171
May 2007
6 M9999-051107
Application Information
Soft Start
A diode-coupled capacitor from COMP to circuit ground
slows the output voltage rise at turn on (Figure 3).
MIC2171
IN
COMP
C2
R1
V
IN
D2
C1
D1
Figure 3. Soft Start
The additional time it takes for the error amplifier to
charge the capacitor corresponds to the time it takes the
output to reach regulation. Diode D1 discharges C1
when V
IN
is removed.
Current Limit
MIC2171
IN
COMP
C2
R3
VIN
Q1
R2
GND
R1
C1
SW
FB VOUT
Note: Input and output
returns not commo
n
ICL 0.6V/R2
Figure 4. Current Limit
The maximum current limit of the MIC2171 can be
reduced by adding a voltage clamp to the COMP output
(Figure 4). This feature can be useful in applications
requiring either a complete shutdown of Q1’s switching
action or a form of current fold-back limiting. This use of
the COMP output does not disable the oscillator,
amplifiers or other circuitry, therefore, the supply current
is never less than approximately 5mA.
Thermal Management
Although the MIC2171 family contains thermal protection
circuitry, for best reliability, avoid prolonged operation
with junction temperatures near the rated maximum.
The junction temperature is determined by first
calculating the power dissipation of the device. For the
MIC2171, the total power dissipation is the sum of the
device operating losses and power switch losses.
The device operating losses are the dc losses
associated with biasing all of the internal functions plus
the losses of the power switch driver circuitry. The dc
losses are calculated from the supply voltage (V
IN
) and
device supply current (I
Q
).The MIC2171 supply current is
almost constant regardless of the supply voltage (see
“Electrical Characteristics”). The driver section losses
(not including the switch) are a function of supply
voltage, power switch current, and duty cycle.
P
(bias+driver)
= (V
IN
I
Q
) + (V
IN(min)
x I
SW
x I
IN
)
where:
P
(bias+driver)
= device operating losses
V
IN(min)
= supply voltage = V
IN
– V
SW
I
Q
= typical quiescent supply current
I
CL
= power switch current limit
I
IN
= typical supply current increase
As a practical example refer to Figure 1.
V
IN
= 5.0V
I
Q
= 0.007A
I
CL
= 2.21A
δ = 66.2% (0.662)
then:
V
IN(min)
= 5.0V – (2.21 x 0.37) = 4.18V
P
(bias+driver)
= (5 x 0.007) + (4.18 x 2.21 x 0.009)
P
(bias+driver)
= 0.1W
Power switch dissipation calculations are greatly
simplified by making two assumptions which are usually
fairly accurate. First, the majority of losses in the power
switch are due to on-losses. To find these losses, assign
a resistance value to the collector/emitter terminals of
the device using the saturation voltage versus collector
current curves (see Typical Performance Character-
istics). Power switch losses are calculated by modeling
the switch as a resistor with the switch duty cycle
modifying the average power dissipation.
P
SW
= (I
SW
)
2
R
SW
δ
where:
δ = duty cycle
FOUT
IN(min)FOUT
VV
VVV
+
−+
=
δ
V
SW
= I
CL
(R
SW
)
V
OUT
= output voltage
V
F
= D1 forward voltage drop at I
OUT
From the Typical performance Characteristics:
R
SW
= 0.37
then:
P
SW
= (2.21)
2
× 0.37 × 0.662
P
SW
= 1.2W
P
(total)
= 1.2 + 0.1
P
(total)
= 1.3W
Micrel, Inc. MIC2171
May 2007
7 M9999-051107
The junction temperature for any semiconductor is
calculated using the following:
T
J
= T
A
+ P
(total)
JA
where:
T
J
= junction temperature
T
A
= ambient temperature (maximum)
P
(total)
= total power dissipation
JA
= junction to ambient thermal resistance
For the practical example:
T
A
= 70°C
JA
= 45°C/W (TO-220)
then:
T
J
= 70 + (1.24 × 45)
T
J
= 126°C
This junction temperature is below the rated maximum of
150°C.
Grounding
Refer to Figure 5. Heavy lines indicate high current
paths.
MIC2171
IN
SW
FB
VC
V
IN
GND
Single point ground
Figure 5. Single Point Ground
A single point ground is strongly recommended for
proper operation.
The signal ground, compensation network ground, and
feed-back network connections are sensitive to minor
voltage variations. The input and output capacitor
grounds and power ground conductors will exhibit
voltage drop when carrying large currents. Keep the
sensitive circuit ground traces separate from the power
ground traces. Small voltage variations applied to the
sensitive circuits can prevent the MIC2171 or any
switching regulator from functioning properly.
Boost Conversion
Refer to Figure 1 for a typical boost conversion
application where a +5V logic supply is available but
+12V at 0.25A is required.
The first step in designing a boost converter is
determining whether inductor L1 will cause the converter
to operate in either continuous or discontinuous mode.
Discontinuous mode is preferred because the feedback
control of the converter is simpler.
When L1 discharges its current completely during the
MIC2171 off-time, it is operating in discontinuous mode.
L1 is operating in continuous mode if it does not
discharge completely before the MIC2171 power switch
is turned on again.
Discontinuous Mode Design
Given the maximum output current, solve equation (1) to
determine whether the device can operate in
discontinuous mode without initiating the internal device
current limit.
(1)
OUT
IN(min)
CL
OUT
V
V
2
I
I
δ
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
≤
(1a)
FOUT
IN(min)FOUT
VV
VVV
+
−+
=
δ
where:
I
CL
= internal switch current limit
I
CL
= 2.5A when δ < 50%
I
CL
= 1.67 (2 – δ) when δ 50%
(Refer to Electrical Characteristics.)
I
OUT
= maximum output current
V
IN(min)
= minimum input voltage = V
IN
– V
SW
δ = duty cycle
V
OUT
= required output voltage
V
F
= D1 forward voltage drop
For the example in Figure 1.
I
OUT
= 0.25A
I
CL
= 1.67 (2–0.662) = 2.24A
V
IN(min)
= 4.18V
δ = 0.662
V
OUT
= 12.0V
V
F
= 0.36V (@ .26A, 70°C)
then:
12
0.6624.178
2
2.235
I
OUT
××
⎟
⎠
⎞
⎜
⎝
⎛
≤
I
OUT
0.258A
This value is greater than the 0.25A output current
requirement, so we can proceed to find the minimum
inductance value of L1 for discontinuous operation at
P
OUT
.
(2)
SWOUT
2
IN
f2P
)(V
L1
δ
≥
Micrel, Inc. MIC2171
May 2007 8
M9999-051107
where:
P
OUT
= 12 × 0.25 = 3W
f
SW
= 1×105Hz (100kHz)
For our practical example:
()
5
2
1013.02
0.6624.178
L1 ×××
×
≥
L1 12.4µH (use 15µH)
Equation (3) solves for L1’s maximum current value.
(3) L1
TV
I
ONIN
L1(peak)
=
where:
T
ON
= / fSW = 6.62×10-6 sec
6
6
L1(peak)
1015
106.624.178
I
−
−
×
××
=
I
L1(peak)
= 1.84A
Use a 15µH inductor with a peak current rating of at
least 2A.
Flyback Conversion
Flyback converter topology may be used in low power
applications where voltage isolation is required or
whenever the input voltage can be less than or greater
than the output voltage. As with the step-up converter
the inductor (transformer primary) current can be
continuous or discontinuous. Discontinuous operation is
recommended.
Figure 2 shows a practical flyback converter design
using the MIC2171.
Switch Operation
During Q1’s on time (Q1 is the internal NPN transistor—
see block diagrams), energy is stored in T1’s primary
inductance. During Q1’s off time, stored energy is
partially discharged into C4 (output filter capacitor).
Careful selection of a low ESR capacitor for C4 may
provide satisfactory output ripple voltage making
additional filter stages unnecessary.
C1 (input capacitor) may be reduced or eliminated if the
MIC2171 is located near a low impedance voltage
source.
Output Diode
The output diode allows T1 to store energy in its primary
inductance (D2 non-conducting) and release energy into
C4 (D2 conducting). The low forward voltage drop of a
Schottky diode minimizes power loss in D2.
Frequency Compensation
A simple frequency compensation network consisting of
R3 and C2 prevents output oscillations.
High impedance output stages (transconductance type)
in the MIC2171 often permit simplified loop-stability
solutions to be connected to circuit ground, although a
more conventional technique of connecting the
components from the error amplifier output to its
inverting input is also possible.
Voltage Clipper
Care must be taken to minimize T1’s leakage
inductance, otherwise it may be necessary to
incorporate the voltage clipper consisting of D1, R4, and
C3 to avoid second breakdown (failure) of the
MIC2171’s internal power switch.
Discontinuous Mode Design
When designing a discontinuous flyback converter, first
determine whether the device can safely handle the
peak primary current demand placed on it by the output
power. Equation (8) finds the maximum duty cycle
required for a given input voltage and output power. If
the duty cycle is greater than 0.8, discontinuous
operation cannot be used.
(8)
()
VSWVI
2P
IN(min)CL
OUT
−
≥
δ
For a practical example let: (see Figure 2)
P
OUT
= 5.0V × 0.5A = 2.5W
V
IN
= 4.0V to 6.0V
I
CL
= 2.5A when δ < 50%
1.67 (2 – δ) when δ 50%
then:
V
IN(min)
= V
IN
– (I
CL
× R
SW
V
IN(min)
= 4 – 0.78V
V
IN(min)
= 3.22V
δ 0.74 (74%), less than 0.8 so discontinuous is
permitted.
A few iterations of equation (8) may be required if the
duty cycle is found to be greater than 50%.
Calculate the maximum transformer turns ratio a, or
N
PRI
/N
SEC
, that will guarantee safe operation of the
MIC2171 power switch.
(9)
SEC
IN(max)CECE
V
VFV
a−
≤
where:
a = transformer maximum turns ratio
V
CE
= power switch collector to emitter maximum
voltage
F
CE
= safety derating factor (0.8 for most
commercial and industrial applications)
V
IN(max)
= maximum input voltage
V
SEC
= transformer secondary voltage (V
OUT
+
V
F
)
Micrel, Inc. MIC2171
May 2007
9 M9999-051107
For the practical example:
V
CE
= 65V max. for the MIC2171
F
CE
= 0.8
V
SEC
= 5.6V
then:
5.6
6.00.865
a−×
≤
a 8.2 (N
PRI
/N
SEC
)
Next, calculate the maximum primary inductance
required to store the needed output energy with a power
switch duty cycle of 55%.
(10)
OUT
2
ON
2
IN(min)SW
PRI
P
TV0.5f
L≥
where:
L
PRI
= maximum primary inductance
f
SW
= device switching frequency (100kHz)
V
IN(min)
= minimum input voltage
T
ON
= power switch on time
then:
2.5
)210(7.4(3.22)1010.5
L
625
PRI
−
×××××
≥
L
PRI
11.4µH
Use a 12µH primary inductance to overcome circuit
inefficiencies.
To complete the design the inductance value of the
secondary is found which will guarantee that the energy
stored in the transformer during the power switch on
time will be completed discharged into the output during
the off-time. This is necessary when operating in
discontinuous-mode.
(11)
OUT
2
OFF
2
SECSW
SEC
P
TV0.5f
L≤
where:
L
SEC
= maximum secondary inductance
T
OFF
= power switch off time
then:
2.5
)10(2.6(5.41)1010.5
L
2625
SEC
−
×××××
≤
L
SEC
7.9µH
Finally, recalculate the transformer turns ratio to insure
that it is less than the value earlier found in equation (9).
(12)
SEC
PRI
L
L
a≤
then:
20.1
7.9
11.4
a=≤
This ratio is less than the ratio calculated in equation (9).
When specifying the transformer it is necessary to know
the primary peak current which must be withstood
without saturating the transformer core.
(13)
PRI
ONIN(min)
PEAK(pri)
L
TV
I=
so:
PRI
6
PEAK(pri)
L
107.63.22
I
−
××
=
I
PEAK(pri)
= 2.1A
Now find the minimum reverse voltage requirement for
the output rectifier. This rectifier must have an average
current rating greater than the maximum output current
of 0.5A.
(14)
aF
a)(VV
V
BR
OUTIN(max)
BR
+
≥
where:
V
BR
= output rectifier maximum peak reverse
voltage rating
a = transformer turns ratio (1.2)
F
BR
= reverse voltage safety derating factor (0.8)
then:
1.20.8
1.2)(5.06.0
V
BR
×
×+
≥
V
BR
12.5V
A 1N5817 will safely handle voltage and current require-
ments in this example.
Forward Conv erters
Micrel’s MIC2171 can be used in several circuit
configurations to generate an output voltage which is
less than the input voltage (buck or step-down topology).
Figure 6 shows the MIC2171 in a voltage step-down
application. Because of the internal architecture of these
devices, more external components are required to
implement a step-down regulator than with other devices
offered by Micrel (refer to the LM257x or MIC457x family
of buck switchers). However, for step-down conversion
requiring a transformer (forward), the MIC2171 is a good
choice.
A 12V to 5V step-down converter using transformer
isolation (forward) is shown in Figure 6. Unlike the
isolated flyback converter which stores energy in the
primary inductance during the controller’s on-time and
releases it to the load during the off-time, the forward
converter transfers energy to the output during the on-
Micrel, Inc. MIC2171
May 2007
10 M9999-051107
time, using the off-time to reset the transformer core. In
the application shown, the transformer core is reset by
the tertiary winding discharging T1’s peak magnetizing
current through D2.
For most forward converters the duty cycle is limited to
50%, allowing the transformer flux to reset with only two
times the input voltage appearing across the power
switch. Although during normal operation this circuit’s
duty cycle is well below 50%, the MIC2172 has a
maximum duty cycle capability of 90%. If 90% was
required during operation (start-up and high load
currents), a complete reset of the transformer during the
off-time would require the voltage across the power
switch to be ten times the input voltage. This would limit
the input voltage to 6V or less for forward converter
applications.
To prevent core saturation, the application given here
uses a duty cycle limiter consisting of Q1, C4 and R3.
Whenever the MIC2171 exceeds a duty cycle of 50%,
T1’s reset winding current turns Q1 on. This action
reduces the duty cycle of the MIC2171 until T1 is able to
reset during each cycle.
D3
1N5819
MIC2171
IN
SW
FB
COMP
C1
22µF
C3
1µF
R2
1k
V
OUT
5V, 1
A
R4
3.74k
1%
R5
1.24k
1%
C5
470µF
L1100µH
V
IN
12V
GND
T1
1:1:1
D4
1N5819
D2
1N5819
* Voltage clipper
†
Duty cycle limiter
D1*
C2*
Q1
†
C4
†
R3
†
R1*
Figure 6. MIC2171 Forward Converter
Micrel, Inc. MIC2171
May 2007
11 M9999-051107
Package Information
5-Pin TO-220 (T)
5-Pin TO-263 (U)
Micrel, Inc. MIC2171
May 2007
12 M9999-051107
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
