VIN
4.5V - 36V
SS/ON1 SS/ON2
SYNC
UV_Delay Vout1
1.3V-0.9VIN
Vout2
1.3V-0.9VIN
LM5642/LM5642X
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
LM5642/LM5642X High Voltage, Dual Synchronous Buck Converter with Oscillator
Synchronization
Check for Samples: LM5642,LM5642X
1FEATURES DESCRIPTION
The LM5642 series consists of two current mode
2 Two Synchronous Buck Regulators synchronous buck regulator controllers operating
180° Out of Phase Operation 180° out of phase with each other at a normal
200 kHz Fixed Nominal Frequency: LM5642 switching frequency of 200kHz for the LM5642 and at
375kHz for the LM5642X.
375 kHz Fixed Nominal Frequency: LM5642X
Synchronizable Switching Frequency from 150 Out of phase operation reduces the input RMS ripple
current, thereby significantly reducing the required
kHz to 250 kHz for the LM5642 and 200 kHz to input capacitance. The switching frequency can be
500 kHz for the LM5642X synchronized to an external clock between 150 kHz
4.5V to 36V Input Range and 250 kHz for the LM5642 and between 200 kHz
50 µA Shutdown Current and 500 kHz for the LM5642X. The two switching
regulator outputs can also be paralleled to operate as
Adjustable Output from 1.3V to 90% of Vin a dual-phase, single output regulator.
0.04% (Typical) Line and Load Regulation
Accuracy The output of each channel can be independently
adjusted from 1.3V to 90% of Vin. An internal 5V rail
Current Mode Control with or without a Sense is also available externally for driving bootstrap
Resistor circuitry.
Independent Enable/Soft-start Pins Allow Current-mode feedback control assures excellent line
Simple Sequential Startup Configuration. and load regulation and wide loop bandwidth for
Configurable for Single Output Parallel excellent response to fast load transients. Current is
Operation. (See Figure 4)sensed across either the Vds of the top FET or
Adjustable Cycle-by-cycle Current Limit across an external current-sense resistor connected
in series with the drain of the top FET.
Input Under-voltage Lockout
Output Over-voltage Latch Protection The LM5642 features analog soft-start circuitry that is
independent of the output load and output
Output Under-voltage Protection with Delay capacitance making the soft-start behavior more
Thermal Shutdown predictable and controllable than traditional soft-start
Self Discharge of Output Capacitors when the circuits.
Regulator is OFF Over-voltage protection is available for both outputs.
TSSOP and HTSSOP (Exposed PAD) Packages A UV-Delay pin is also available to allow delayed shut
off time for the IC during an output under-voltage
APPLICATIONS event.
Embedded Computer Systems Typical Application Circuit
Navigation Systems
Telecom Systems
Set-Top Boxes
WebPAD
Point Of Load Power Architectures
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2003–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
RSNS2
SW2
HDRV2
CBOOT2
VDD2
LDRV2
PGND
VIN
LDRV1
VDD1
CBOOT1
HDRV1
SW1
RSNS1
ON/SS2
FB2
COMP2
ILIM1
COMP1
FB1
SYNC
UVDELAY
VLIN5
SGND
KS1
ON/SS1
ILIM2
KS2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
RSNS2
SW2
HDRV2
CBOOT2
VDD2
LDRV2
PGND
VIN
LDRV1
VDD1
CBOOT1
HDRV1
SW1
RSNS1
ON/SS2
FB2
COMP2
ILIM1
COMP1
FB1
SYNC
UVDELAY
VLIN5
SGND
KS1
ON/SS1
ILIM2
KS2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
DAP
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
Connection Diagram
Figure 1. Top View Figure 2. Top View
PIN DESCRIPTIONS
The positive (+) Kelvin sense for the internal current sense amplifier of Channel 1. Use a separate trace to
KS1 (Pin 1) connect this pin to the current-sense point. It should be connected to VIN as close as possible to the current-
sense resistor. When no current-sense resistor is used, connect as close as possible to the drain node of the
upper MOSFET.
Current limit threshold setting for Channel 1. It sinks a constant current of 9.9 µA, which is converted to a voltage
ILIM1 (Pin 2) across a resistor connected from this pin to VIN. The voltage across the resistor is compared with either the VDS
of the top MOSFET or the voltage across the external current sense resistor to determine if an over-current
condition has occurred in Channel 1.
Compensation pin for Channel 1. This is the output of the internal transconductance error amplifier. The loop
COMP1 (Pin 3) compensation network should be connected between this pin and the signal ground, SGND (Pin 8).
Feedback input for channel 1. Connect to VOUT through a voltage divider to set the Channel 1 output voltage.
FB1 (Pin 4)
The switching frequency of the LM5642 can be synchronized to an external clock.
SYNC (Pin 5) SYNC = LOW: Free running at 200 kHz for LM5642, and at 375kHz for LM5642X. Channels are 180° out of
phase.
SYNC = HIGH: Waiting for external clock
SYNC = Falling Edge: Channel 1 HDRV pin goes high. Channel 2 HDRV pin goes high after 2.5 µs delay. The
maximum SYNC pulse width must be greater than 100 ns.
For SYNC = Low operation, connect this pin to signal ground through a 220 kresistor.
A capacitor from this pin to ground sets the delay time for UVP. The capacitor is charged from a 5 µA current
UV_DELAY (Pin 6) source. When UV_DELAY charges to 2.3V (typical), the system immediately latches off. Connecting this pin to
ground will disable the output under-voltage protection.
The output of an internal 5V LDO regulator derived from VIN. It supplies the internal bias for the chip and powers
VLIN5 (Pin 7) the bootstrap circuitry for gate drive. Bypass this pin to signal ground with a minimum of 4.7 µF ceramic capacitor.
The ground connection for the signal-level circuitry. It should be connected to the ground rail of the system.
SGND (Pin 8)
Channel 1 enable pin. This pin is internally pulled up to one diode drop above VLIN5. Pulling this pin below 1.2V
ON/SS1 (Pin 9) (open-collector type) turns off Channel 1. If both ON/SS1 and ON/SS2 pins are pulled below 1.2V, the whole chip
goes into shut down mode. Adding a capacitor to this pin provides a soft-start feature that minimizes inrush
current and output voltage overshoot.
Channel 2 enable pin. See the description for Pin 9, ON/SS1. May be connected to ON/SS1 for simultaneous
ON/SS2 (Pin 10) startup or for parallel operation.
Feedback input for channel 2. Connect to VOUT through a voltage divider to set the Channel 2 output voltage.
FB2 (Pin 11)
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PIN DESCRIPTIONS (continued)
Compensation pin for Channel 2. This is the output of the internal transconductance error amplifier. The loop
COMP2 (Pin 12) compensation network should be connected between this pin and the signal ground SGND (Pin 8).
Current limit threshold setting for Channel 2. See ILIM1 (Pin 2).
ILIM2 (Pin 13)
The positive (+) Kelvin sense for the internal current sense amplifier of Channel 2. See KS1 (Pin 1).
KS2 (Pin 14)
The negative (-) Kelvin sense for the internal current sense amplifier of Channel 2. Connect this pin to the low side
RSNS2 (Pin 15) of the current sense resistor that is placed between VIN and the drain of the top MOSFET. When the Rds of the
top MOSFET is used for current sensing, connect this pin to the source of the top MOSFET. Always use a
separate trace to form a Kelvin connection to this pin.
Switch-node connection for Channel 2, which is connected to the source of the top MOSFET of Channel 2. It
SW2 (Pin 16) serves as the negative supply rail for the top-side gate driver, HDRV2.
Top-side gate-drive output for Channel 2. HDRV is a floating drive output that rides on the corresponding
HDRV2 (Pin 17) switching-node voltage.
Bootstrap capacitor connection. It serves as the positive supply rail for the Channel 2 top-side gate drive. Connect
CBOOT2 (Pin 18) this pin to VDD2 (Pin 19) through a diode, and connect the low side of the bootstrap capacitor to SW2 (Pin16).
The supply rail for the Channel 2 low-side gate drive. Connected to VLIN5 (Pin 7) through a 4.7resistor and
VDD2 (Pin 19) bypassed to power ground with a ceramic capacitor of at least 1µF. Tie this pin to VDD1 (Pin 24).
Low-side gate-drive output for Channel 2.
LDRV2 (Pin 20)
The power ground connection for both channels. Connect to the ground rail of the system.
PGND (Pin 21)
The power input pin for the chip. Connect to the positive (+) input rail of the system. This pin must be connected
VIN (Pin 22) to the same voltage rail as the top FET drain (or the current sense resistor when used).
Low-side gate-drive output for Channel 1.
LDRV1 (Pin 23)
The supply rail for Channel 1 low-side gate drive. Tie this pin to VDD2 (Pin 19).
VDD1 (Pin 24)
Bootstrap capacitor connection. This pin serves as the positive supply rail for the Channel 1 top-side gate drive.
CBOOT1 (Pin 25) See CBOOT2 (Pin 18).
Top-side gate-drive output for Channel 1. See HDRV2 (Pin 17).
HDRV1 (Pin 26)
Switch-node connection for Channel 1. See SW2 (Pin16).
SW1 (Pin 27)
The negative (-) Kelvin sense for the internal current sense amplifier of Channel 1. See RSNS2 (Pin 15).
RSNS1 (Pin 28)
The power ground connection for both channels. Connect to the ground rail of the system. Use of multiple vias to
PGND (DAP) internal ground plane or GND layer helps to dissipate heat generated by output power.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS(1)(2)
Voltages from the indicated pins to SGND/PGND:
VIN, ILIM1, ILIM2, KS1, KS2 0.3V to 38V
SW1, SW2, RSNS1, RSNS2 0.3 to (VIN + 0.3)V
FB1, FB2, VDD1, VDD2 0.3V to 6V
SYNC, COMP1, COMP2, UV Delay 0.3V to (VLIN5 +0.3)V
ON/SS1, ON/SS2 (3) 0.3V to (VLIN5 +0.6)V
CBOOT1, CBOOT2 43V
CBOOT1 to SW1, CBOOT2 to SW2 0.3V to 7V
LDRV1, LDRV2 0.3V to (VDD+0.3)V
HDRV1 to SW1, HDRV2 to SW2 0.3V
HDRV1 to CBOOT1, HDRV2 to CBOOT2 +0.3V
Power Dissipation (TA= 25°C)(4)
TSSOP 1.1W
HTSSOP 3.4W
Ambient Storage Temp. Range 65°C to +150°C
Soldering Dwell Time, Temp.(5) Wave 4 sec, 260°C
Infrared 10sec, 240°C
Vapor Phase 75sec, 219°C
ESD Rating (6) 2kV
(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance
characteristics may degrade when the device is not operated under the listed test conditions.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) ON/SS1 and ON/SS2 are internally pulled up to one diode drop above VLIN5. Do not apply an external pull-up voltage to these pins. It
may cause damage to the IC.
(4) The maximum allowable power dissipation is calculated by using PDMAX = (TJMAX - TA)/θJA, where TJMAX is the maximum junction
temperature, TAis the ambient temperature and θJA is the junction-to-ambient thermal resistance of the specified package. The power
dissipation ratings results from using 125°C, 25°C, and 90.6°C/W for TJMAX, TA, and θJA respectively. A θJA of 90.6°C/W represents the
worst-case condition of no heat sinking of the 28-pin TSSOP. The HTSSOP package has a θJA of 29°C/W. The HTSSOP package
thermal ratings results from the IC being mounted on a 4 layer JEDEC standard board using the same temperature conditions as the
TSSOP package above. A thermal shutdown will occur if the temperature exceeds the maximum junction temperature of the device.
(5) See http://www.ti.com for other methods of soldering plastic small-outline packages.
(6) For testing purposes, ESD was applied using the human-body model, a 100pF capacitor discharged through a 1.5 kresistor.
OPERATING RATINGS (1)
VIN (VLIN5 tied to VIN) 4.5V to 5.5V
VIN (VIN and VLIN5 separate) 5.5V to 36V
Junction Temperature 40°C to +125°C
(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance
characteristics may degrade when the device is not operated under the listed test conditions.
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SNVS219K JUNE 2003REVISED APRIL 2013
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol Parameter Conditions Min Typ Max Units
System
ΔVOUT/VOUT Load Regulation VIN = 28V, Vcompx = 0.5V to 1.5V 0.04 %
Line Regulation 5.5V VIN 36V, Vcompx =1.25V 0.04 %
VFB1_FB2 Feedback Voltage 5.5V VIN 36V 1.2154 1.2364 1.2574 V
-20°C to 85°C 1.2179 1.2364 1.2549
IVIN Input Supply Current VON_SSx > 2V 1.1 2.0 mA
5.5V VIN 36V
Shutdown (3) 50 110 µA
VON_SS1 = VON_SS2= 0V
VLIN5 VLIN5 Output Voltage IVLIN5 = 0 to 25mA, 4.70 55.30 V
5.5V VIN 36V
VCLos Current Limit Comparator VIN = 6V ±2 ±7.0 mV
Offset (VILIMX VRSNSX)
ICL Current Limit Sink Current 8.4 9.9 11.4 µA
Iss_SC1, Soft-Start Source Current VON_ss1 = VON_ss2 = 1.5V (on) 0.5 2.4 5.0 µA
Iss_SC2
Iss_SK1, Soft-Start Sink Current VON_ss1 = VON_ss2 = 1.5V 25.5 10 µA
Iss_SK2
VON_SS1, Soft-Start On Threshold 0.7 1.12 1.4 V
VON_SS2
VSSTO Soft-Start Timeout (4) 3.4 V
Threshold
Isc_uvdelay UV_DELAY Source Current UV-DELAY = 2V 259µA
Isk_uvdelay UV_DELAY Sink Current UV-DELAY = 0.4V 0.2 0.48 1.2 mA
VUVDelay UV_DELAY Threshold 2.3 V
Voltage
VUVP FB1, FB2, Under Voltage As a percentage of nominal output voltage 75 80.7 86 %
Protection Latch Threshold (falling edge)
Hysteresis 3.7 %
VOVP VOUT Overvoltage As a percentage measured at VFB1, VFB2 107 114 122 %
Shutdown Latch Threshold
Swx_R SW1, SW2 ON-Resistance VSW1 = VSW2 = 0.4V 420 487 560
(1) A typical is the center of characterization data measured with low duty cycle pulse tsting at TA= 25°C. Typicals are not ensured.
(2) All limits are specified. All electrical characteristics having room-temperature limits are tested during production with TA= TJ= 25°C. All
hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical
process control.
(3) Both switching controllers are off. The linear regulator VLIN5 remains on.
(4) When SS1 and SS2 pins are charged above this voltage and either of the output voltages at Vout1 or Vout2 is still below the regulation
limit, the under voltage protection feature is initialized.
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ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol Parameter Conditions Min Typ Max Units
Gate Drive
ICBOOT CBOOTx Leakage Current VCBOOT1 = VCBOOT2 = 7V 10 nA
ISC_DRV HDRVx and LDRVx Source VCBOOT1 = VCBOOT2 = 5V, VSWx=0V, 0.5 A
Current HDRVx=LDRVx=2.5V
Isk_HDRV HDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, HDRVX 0.8 A
= 2.5V
Isk_LDRV LDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, LDRVX 1.1 A
= 2.5V
RHDRV HDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1
Resistance VSW1 = VSW2 = 0V
HDRV1 & 2 Sink On- 1.5
Resistance
RLDRV LDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1
Resistance VSW1 = VSW2 = 0V
VDD1 = VDD1 = 5V
LDRV1 & 2 Sink On- 1.1
Resistance
Oscillator and Sync Controls
5.5 VIN 36V, LM5642 166 200 226
Fosc Oscillator Frequency kHz
5.5 VIN 36V, LM5642X 311 375 424
Don_max Maximum On-Duty Cycle VFB1 = VFB2 = 1V, Measured at pins 96 98.9 %
HDRV1 and HDRV2
Ton_min Minimum On-Time 166 ns
SSOT_delta HDRV1 and HDRV2 Delta ON/SS1 = ON/SS2 = 2V 20 250 ns
On Time
VHS SYNC Pin Min High Input 21.52 V
VLS SYNC Pin Max Low Input 1.44 0.8 V
Error Amplifier
IFB1, IFB2 Feedback Input Bias VFB1_FIX = 1.5V, VFB2_FIX = 1.5V 80 ±200 nA
Current
Icomp1_SC, COMP Output Source VFB1_FIX = VFB2_FIX = 1V, 6127
Icomp2_SC Current VCOMP1 = VCOMP2 = 1V µA
-20°C to 85°C 18
Icomp1_SK, COMP Output Sink Current VFB1_FIX = VFB2_FIX = 1.5V and 6118
Icomp2_SK VCOMP1 = VCOMP2 = 0.5V µA
-20°C to 85°C 18
gm1, gm2 Transconductance 720 µmho
GISNS1, Current Sense Amplifier VCOMPx = 1.25V 4.2 5.2 7.5
GISNS2 (1&2) Gain
Voltage References and Linear Voltage Regulators
UVLO VLIN5 Under-voltage ON/SS1, ON/SS2 transition
Lockout from low to high 3.6 4.0 4.4 V
Threshold Rising
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4
2
SGND
8
COMP1
3
ON/SS1
9
COMP2
12
28
1
26
27
VDD1
24
23
21
VIN
22
VLIN5
7
SYNC
5
VDD2
19
UV_DELAY
6
11
13
15
14
18
17
16
20
IC1
LM5642
ON/SS2
10
+
25
C1
C34
R28
C11
C12
R27
R23 R24
C27 C26
C20
C19
C2
R1
Q1
Q2
L1
R10
R11
C3
R2
R7
C4R6
C6
C7
D3A VDD
C9
1 PF
100 nF
220 k:
10 nF
10 nF
4.7 :
1 PF4.7 PF
8.2 nF
15 nF
8.45
k:
13.7
k:
Si4840DY
Si4850EY
10 PF
50V
2.8Arms
VIN
10 m:
10 nF 100 pF
12 k:
100:
100 pF
100:
100 nF
BAS40-06
4.2 PH
7 m: 330 PF
6.3V
10 m:
2.26
k:
4.99 k:
+
C13
R13
Q4
Q5
L2
R19
R20
C14
R14
R15
R16
C16
D3B VDD
C23
Si4840DY
Si4850EY
10 PF
50V
2.8Arms
VIN
10 m:
10 nF 100 pF
6.8 k:
100:
100 pF
100:
100 nF
BAS40-06
10 P+
12 m: 330 PF
6.3V
10 m:
8.25
k:
4.99 k:
C16
C25
S1
S2
VDD
SYNC
Vin = 24V+10%
Vo1 = 1.8V, 7A
Vo2 = 3.3V, 4A
FB1
ILIM1
RSNS1
KS1
CBOOT1
HDRV1
SW1
LDRV1
PGND
FB2
ILIM2
RSNS2
KS2
SW2
HDRV2
CBOOT2
LDRV2
LM5642, LM5642X
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SNVS219K JUNE 2003REVISED APRIL 2013
Figure 3. Typical 2 Channel Application Circuit
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4
2
8
3
9
12
28
1
26
27
24
23
21
22
7
5
19
6
11
13
15
14
18
17
16
20
IC1
LM5642
10
+
25
C1
C34
R28
C11
R27
R23
C27 C26 C19
C2
R1
Q1
Q2
L1
R10
R11
C3
R2
R7
C4R6
C6
C7
D3A VDD
C9
1 PF
100
nF
220 k:
22 nF
4.7:
1 PF 4.7 PF
11.5 k:
Si4470DY x 2
Si4850EY
10 PF
50V
2.8Arms
VIN
10 m:
10 nF 100 pF
16.9 k:
100:
100 pF
100:
100 nF
BAS40-06
2.7 PH
4.5 m: 1000 PF
16V
22 m:
2.26
k:
4.99 k:
+
C13
R13
Q4
Q5
L2
C14
R14
R15
R16
C16
D3B VDD
C23
Si4470DY x 2
Si4850EY
10 PF
50V
2.8Arms
VIN
10 m:
10 nF 100 pF
16.9 k:
100:
100 pF
100:
100 nF
BAS40-06
2.7 PH
4.5 m: 1000 PF
16V
22 m:
C16
C25
S1
VDD
SYNC Vo = 1.8V, 20A
Q6
Q3
27 nF
C10
1 PF
C24
1 PF
Vin = 30V
r
10%
SGND
COMP1
ON/SS1
COMP2
VDD1
VIN
VLIN5
SYNC
VDD2
UV_DELAY
ON/SS2
FB1
ILIM1
RSNS1
KS1
CBOOT1
HDRV1
SW1
LDRV1
PGND
ILIM2
RSNS2
KS2
SW2
HDRV2
CBOOT2
LDRV2
FB2
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
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Figure 4. Typical Single Channel Application Circuit
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-
Vref
+
-
ILIMx
KSx
10 PA
RSNSx
+
VDDx
LDRVx
PGNDx
Input Power
Supply
Shoot through
protection
sequencer
Q
R
S
Shifter
and latch
+
-
+
0.50V
+
-
Corrective
ramp
SGND
OSC
200 kHz LM5642
or
375 kHz LM5642X
02.5 Ps
delay
To Ch2
Q
Q
R
S
Reset by
POR or SD
UVP
OVP
ON/SSx
ON/OFF
and
S/S
control
PWM comp
5V LDO
(Allways ON)
fault FAULT
TSD
UVLO
OVP
UVPG1
comparator
UVP
COMPx
CHx
output
error amp
BG
Ch1 and Ch2 are identical
HDRVx
CHx
Output
S/S level Cycle
Skip
comp
SWx
Q
Q
R
S
UV_DELAY
+
-
+
-
ILIM
Comp
ISENSE
amp
+
-CBOOTx
-
+
BG
reference
BG
IREF Current
bias
Voltage
and
Current
generator
From another Ch.
From
another
CH.
Bias
Generator
VIN
Normal:
ON
SS:
ON
FBx
2 PA
VLIN5
SD Disable
Active
discharge
Rdson =
500:
UV
PWM logic
control
SYNC
Q
7 PA
5 PA
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
BLOCK DIAGRAM
Figure 5. Block Diagram
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Product Folder Links: LM5642 LM5642X
100Ps/DIV
Io2, 2A/DIV
Vo2, 100mV/DIV
100Ps/DIV
Io1, 2A/DIV
Vo1, 100mV/DIV
20ms/DIV
UV DELAY, 1V/div
Vo1, 1V/div
Vo2, 1V/div
Io1, 5A/div
100ms/DIV
ON/SS1 and 2, 5V/div
Vo1, 1V/div
Vo2, 1V/div
4 ms/DIV
ON/SS1 and 2,
5V/div, VIN = 24V
Vo1, 2V/div
Vo2, 2V/div
Vo1, 2V/div
Vo2, 2V/div
5V/div, VIN = 36V
ON/SS1 and 2,
20ms/DIV
UV DELAY, 2V/div
ON/SS1, 2V/div
Vo1,
1V/div
Io1, 5A/div
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS
Softstart Waveforms (No-Load Both Channels) UVP Startup Waveform (VIN = 24V)
Figure 6. Figure 7.
Over-Current and UVP Shutdown (VIN = 24V, Io2 = 0A) Shutdown Waveforms (VIN = 24V, No-Load)
Figure 8. Figure 9.
Ch.1 Load Transient Response (VIN = 24V, Vo1 = 1.8V) Ch.2 Load Transient Response (VIN = 24V, Vo2 = 3.3V)
Figure 10. Figure 11.
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5.5 8 12 16 20 24 28 32 36
VIN (V)
VLIN5 (V)
5.095
5.09
5.085
5.08
5.075
5.07
5.065
5.5 8 12 16 20 24 28 32 36
VIN
IQ (PA)
53.5
49.5
50.5
51.5
52.5
50
51
52
53
-40 -20 0 25 50 75 100
TEMPERATURE (oC)
30
35
40
45
50
55
IQ (PA)
125
100Ps/DIV
Io2, 2A/DIV
Vo2, 100mV/DIV
100Ps/DIV
Io1, 2A/DIV
Vo1, 100mV/DIV
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Ch. 2 Load Transient Response (VIN = 36V, Vo2 = 3.3V) Ch.1 Load Transient Response (VIN = 36V, Vo1 = 1.8V)
Figure 12. Figure 13.
Input Supply Current vs Temperature Input Supply Current vs VIN
(Shutdown Mode VIN = 28V) Shutdown Mode (25°C)
Figure 14. Figure 15.
VLIN5 vs Temperature VLIN5 vs VIN (25°C)
Figure 16. Figure 17.
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0 1 2 3 4 5 6 7
LOAD CURRENT (A)
EFFICIENCY (%)
VIN = 24V
VIN = 36V
30
40
50
60
70
80
90
100
-40 -20 0 25 50 75 100 125
1.232
1.2325
1.233
1.2335
1.234
1.2345
1.235
1.2355
1.236
1.2365
1.237
VREF (V)
TEMPERATURE (oC)
-40 -20 0 25 50 75 100 125
184
186
188
190
192
194
196
198
200
202
204
FREQUENCY (kHz)
TEMPERATURE (oC)
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Operating Frequency vs Temperature
FB Reference Voltage vs Temperature (VIN = 28V)
Figure 18. Figure 19.
Error Amplifier Tranconductance Gain
vs Efficiency vs Load Current Using Resistor Sense
Temperature Ch.1 = 1.8V, Ch.2 = Off
Figure 20. Figure 21.
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0 1 2 3 4 5
EFFICIENCY (%)
LOAD CURRENT (A)
VIN = 24V
VIN = 36V
50
60
70
80
90
100
0 1 2 3 4 5
EFFICIENCY (%)
LOAD CURRENT (A)
VIN = 24V
VIN = 36V
50
60
70
80
90
100
0 1 2 3 4 5 6 7
LOAD CURRENT (A)
EFFICIENCY (%)
VIN = 24V
VIN = 36V
30
40
50
60
70
80
90
100
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Efficiency vs Load Current Efficiency vs Load Current Using Vds Sense
Ch.2 = 3.3V, Ch.1 = Off Ch.2 = 1.8V, Ch.2 = Off
Figure 22. Figure 23.
Efficiency vs Load Current Using Vds Sense
Ch.2 = 3.3V, Ch.1 = Off
Figure 24.
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+
-
2PA
7PA
+
-
Q
Q
R
S
S>R
1.2V/
1.05V ON/OFF
comparator
+
-S/S level
S/S buffer
ON: 2.4PA source
Fault: 5.5PA sink
ON/SSx ONx
fault
disable
Vss = 1.5 Vo
Vin + 1
¹
·
©
§
Css = Iss x tss
Vss
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
OPERATING DESCRIPTIONS
SOFT START
The ON/SS1 pin has dual functionality as both channel enable and soft start control. Referring to the soft start
block diagram is shown in Figure 25, the LM5642 will remain in shutdown mode while both soft start pins are
grounded.
In a normal application (with a soft start capacitor connected between the ON/SS1 pin and SGND) soft start
functions as follows: As the input voltage rises (note, Iss starts to flow when VIN 2.2V), the internal 5V LDO
starts up, and an internal 2.4 µA current charges the soft start capacitor. During soft start, the error amplifier
output voltage at the COMPx pin is clamped at 0.55V and the duty cycle is controlled only by the soft start
voltage. As the SSx pin voltage ramps up, the duty cycle increases proportional to the soft start ramp, causing
the output voltage to ramp up. The rate at which the duty cycle increases depends on the capacitance of the soft
start capacitor. The higher the capacitance, the slower the output voltage ramps up. When the corresponding
output voltage exceeds 98% (typical) of the set target voltage, the regulator switches from soft start to normal
operating mode. At this time, the 0.55V clamp at the output of the error amplifier releases and peak current
feedback control takes over. Once in peak current feedback control mode, the output voltage of the error
amplifier will travel within a 0.5V and 2V window to achieve PWM control. See Figure 26.
The amount of capacitance needed for a desired soft-start time can be approximated in the following equation:
where
Iss = 2.4 µA for one channel and 4.8µA if the channels are paralleled
tss is the desired soft-start time (1)
Finally,
(2)
During soft start, over-voltage protection and current limit remain in effect. The under voltage protection feature is
activated when the ON/SS pin exceeds the timeout threshold (3.4V typical). If the ON/SSx capacitor is too small,
the duty cycle may increase too rapidly, causing the device to latch off due to output voltage overshoot above the
OVP threshold. This becomes more likely in applications with low output voltage, high input voltage and light
load. A capacitance of 10 nF is recommended at each soft start pin to provide a smooth monotonic output ramp.
Figure 25. Soft-Start and ON/OFF
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OVP 1/2
1.13BG
in: 0.84BG
out:0.80BG
FBx
Q
Q
S
R
shutdown
HDRV: off
LDRV:off
UVPx
UVP 5u
AUV_DELAY
from other CH.
ONx
SS Timeout
Q
Q
S
R
from other CH. shutdown
latch OVP
HDRV: off
LDRV:on
OVP
latch UVP
fault
UVLO
TSD
+
-
+
-
OVPx
SD
power on
reset
+
-
+
-SS:0.55V
OP:2V
0.45V
high clamp
low clamp
COMPx
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
Figure 26. Voltage Clamp at COMPx Pin
Figure 27. OVP and UVP
OVER VOLTAGE PROTECTION (OVP)
If the output voltage on either channel rises above 113% of nominal, over voltage protection activates. Both
channels will latch off. When the OVP latch is set, the high side FET driver, HDRVx, is immediately turned off
and the low side FET driver, LDRVx, is turned on to discharge the output capacitor through the inductor. To reset
the OVP latch, either the input voltage must be cycled, or both channels must be switched off (both ON/SS pins
pulled low).
UNDER VOLTAGE PROTECTION (UVP) AND UV DELAY
If the output voltage on either channel falls below 80% of nominal, under voltage protection activates. As shown
in Figure 27, an under-voltage event will shut off the UV_DELAY MOSFET, which will allow the UV_DELAY
capacitor to charge with 5µA (typical). If the UV_DELAY pin voltage reaches the 2.3V threshold both channels
will latch off. UV_DELAY will then be disabled and the UV_DELAY pin will return to 0V. During UVP, both the
high side and low side FET drivers will be turned off. If no capacitor is connected to the UV_DELAY pin, the UVP
latch will be activated immediately. To reset the UVP latch, either the input voltage must be cycled, or both
ON/SS pins must be pulled low. The UVP function can be disabled by connecting the UV_DELAY pin to ground.
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THERMAL SHUTDOWN
The LM5642 IC will enter thermal shutdown if the die temperature exceeds 160°C. The top and bottom FETs of
both channels will be turned off immediately. In addition, both soft start capacitors will begin to discharge through
separate 5.5 µA current sinks. The voltage on both capacitors will settle to approximately 1.1V, where it will
remain until the thermal shutdown condition has cleared. The IC will return to normal operating mode when the
die temperature has fallen to below 146°C. At this point the two soft start capacitors will begin to charge with
their normal 2.4 µA current sources. This allows a controlled return to normal operation, similar to the soft start
during turn-on. If the thermal shutdown condition clears before the voltage on the soft start capacitors has fallen
to 1.1V, the capacitors will first be discharged to 1.1V, and then immediately begin charging back up.
OUTPUT CAPACITOR DISCHARGE
Each channel has an embedded 480MOSFET with the drain connected to the SWx pin. This MOSFET will
discharge the output capacitor of its channel if its channel is off, or the IC enters a fault state caused by one of
the following conditions:
1. UVP
2. UVLO
If an output over voltage event occurs, the HDRVx will be turned off and LDRVx will be turned on immediately to
discharge the output capacitors of both channels through the inductors.
BOOTSTRAP DIODE SELECTION
The bootstrap diode and capacitor form a supply that floats above the switch node voltage. VLIN5 powers this
supply, creating approximately 5V (minus the diode drop) which is used to power the high side FET drivers and
driver logic. When selecting a bootstrap diode, Schottky diodes are preferred due to their low forward voltage
drop, but care must be taken for circuits that operate at high ambient temperature. The reverse leakage of some
Schottky diodes can increase by more than 1000x at high temperature, and this leakage path can deplete the
charge on the bootstrap capacitor, starving the driver and logic. Standard PN junction diodes and fast rectifier
diodes can also be used, and these types maintain tighter control over reverse leakage current across
temperature.
SWITCHING NOISE REDUCTION
Power MOSFETs are very fast switching devices. In synchronous rectifier converters, the rapid increase of drain
current in the top FET coupled with parasitic inductance will generate unwanted Ldi/dt noise spikes at the source
node of the FET (SWx node) and also at the VIN node. The magnitude of this noise will increase as the output
current increases. This parasitic spike noise may produce excessive electromagnetic interference (EMI), and can
also cause problems in device performance. Therefore, it must be suppressed using one of the following
methods.
When using resistor based current sensing, it is strongly recommended to add R-C filters to the current sense
amplifier inputs as shown in Figure 29. This will reduce the susceptibility to switching noise, especially during
heavy load transients and short on time conditions. The filter components should be connected as close as
possible to the IC.
As shown in Figure 28, adding a resistor in series with the HDRVx pin will slow down the gate drive, thus slowing
the rise and fall time of the top FET, yielding a longer drain current transition time.
Usually a 3.3to 4.7resistor is sufficient to suppress the noise. Top FET switching losses will increase with
higher resistance values.
Small resistors (1-5 ohms) can also be placed in series with the CBOOTx pin to effectively reduce switch node
ringing. A CBOOT resistor will slow the rise time of the FET, whereas a resistor at HDRV will increase both rise
and fall times.
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Rsw
CBOOTx
4R7
HDRVx
SWx 0.1 PF
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SNVS219K JUNE 2003REVISED APRIL 2013
Figure 28. HDRV Series Resistor
CURRENT SENSING AND LIMITING
As shown in Figure 29, the KSx and RSNSx pins are the inputs of the current sense amplifier. Current sensing is
accomplished either by sensing the Vds of the top FET or by sensing the voltage across a current sense resistor
connected from VIN to the drain of the top FET. The advantages of sensing current across the top FET are
reduced parts count, cost and power loss.
The RDS-ON of the top FET is not as stable over temperature and voltage as a sense resistor, hence great care
must be used in layout for VDS sensing circuits. At input voltages above 30V, the maximum recommended output
current is 5A per channel.
Keeping the differential current-sense voltage below 200mV ensures linear operation of the current sense
amplifier. Therefore, the RDS-ON of the top FET or the current sense resistor must be small enough so that the
current sense voltage does not exceed 200 mV when the top FET is on. There is a leading edge blanking circuit
that forces the top FET on for at least 166ns. Beyond this minimum on time, the output of the PWM comparator
is used to turn off the top FET. Additionally, a minimum voltage of at least 50 mV across Rsns is recommended
to ensure a high SNR at the current sense amplifier.
Assuming a maximum of 200 mV across Rsns, the current sense resistor can be calculated as follows:
where
Imax is the maximum expected load current, including overload multiplier (ie: 120%)
Irip is the inductor ripple current (see Equation 17) (3)
The above equation gives the maximum allowable value for Rsns. Conduction losses will increase with larger
Rsns, thus lowering efficiency.
The peak current limit is set by an external resistor connected between the ILIMx pin and the KSx pin. An
internal 10 µA current sink on the ILIMx pin produces a voltage across the resistor to set the current limit
threshold which is then compared to the current sense voltage. A 10 nF capacitor across this resistor is required
to filter unwanted noise that could improperly trip the current limit comparator.
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+
-
+
-
10 PA
comp
ISENSE
amp
POWER
SUPPLY
100
100
100 pF
100 pF
LIMx
KSx
RSNSx
10 nF
20m
13k
LIMx
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
Figure 29. Current Sense and Current Limit
Current limit is activated when the inductor current is high enough to cause the voltage at the RSNSx pin to be
lower than that of the ILIMx pin. This toggles the Ilim comparator, thus turning off the top FET immediately. The
comparator is disabled when the top FET is turned off and during the leading edge blanking time. The equation
for current limit resistor, Rlim, is as follows:
where
Ilim is the load current at which the current limit comparator will be tripped (4)
When sensing current across the top FET, replace Rsns with the RDS-ON of the FET. This calculated Rlim value
specifies that the minimum current limit will not be less than Imax. It is recommended that a 1% tolerance resistor
be used.
When sensing across the top FET (VDS sensing), RDS-ON will show more variation than a current-sense resistor,
largely due to temperature variation. RDS-ON will increase proportional to temperature according to a specific
temperature coefficient. Refer to the FET manufacturer's datasheet to determine the range of RDS-ON values over
operating temperature or see the Component Selection section (Equation 27) for a calculation of maximum RDS-
ON. This will prevent RDS-ON variations from prematurely tripping the current limit comparator as the operating
temperature increases.
To ensure accurate current sensing using VDS sensing, special attention in board layout is required. The KSx and
RSNSx pins require separate traces to form a Kelvin connection at the corresponding current sense nodes. In
addition, the filter components R14, R16, C14, C15 should be removed.
INPUT UNDER VOLTAGE LOCKOUT (UVLO)
The input under-voltage lock out threshold, which is sensed via the VLIN5 internal LDO output, is 4.0V (typical).
Below this threshold, both HDRVx and LDRVx will be turned off and the internal 480MOSFETs will be turned
on to discharge the output capacitors through the SWx pins. When the input voltage is below the UVLO
threshold, the ON/SS pins will sink 5mA to discharge the soft start capacitors and turn off both channels. As the
input voltage increases again above 4.0V, UVLO will be de-activated, and the device will restart through a normal
soft start phase. If the voltage at VLIN5 remains below 4.5V, but above the 4.0V UVLO threshold, the device
cannot be ensured to operate within specification.
If the input voltage is between 4.0V and 5.2V, the VLIN5 pin will not regulate, but will follow approximately 200
mV below the input voltage.
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DUAL-PHASE PARALLEL OPERATION
In applications with high output current demand, the two switching channels can be configured to operate as a
two phase converter to provide a single output voltage with current sharing between the two switching channels.
This approach greatly reduces the stress and heat on the output stage components while lowering input ripple
current. The inductor ripple currents also cancel to a varying degree which results in lowered output ripple
voltage. Figure 4 shows an example of a typical two-phase circuit. Because precision current sense is the
primary design criteria to ensure accurate current sharing between the two channels, both channels must use
external sense resistors for current sensing. To minimize the error between the error amplifiers of the two
channels, tie the feedback pins FB1 and FB2 together and connect to a single voltage divider for output voltage
sensing. Also, tie the COMP1 and COMP2 together and connect to the compensation network. ON/SS1 and
ON/SS2 must be tied together to enable and disable both channels simultaneously.
EXTERNAL FREQUENCY SYNC
The LM5642 series has the ability to synchronize to external sources in order to set the switching frequency. This
allows the LM5642 to use frequencies from 150 kHz to 250 kHz and the LM5642X to use frequencies from 200
kHz to 500 kHz. Lowering the switching frequency allows a smaller minimum duty cycle, DMIN, and hence a
greater range between input and output voltage. Increasing switching frequency allows the use of smaller output
inductors and output capacitors (see Component Selection). In general, synchronizing all the switching
frequencies in multi-converter systems makes filtering of the switching noise easier.
The sync input can be from a system clock, from another switching converter in the system, or from any other
periodic signal with a logic low-level less than 1.4V and a logic high level greater than 2V. Both CMOS and TTL
level inputs are acceptable.
The LM5642 series uses a fixed delay between Channel 1 and Channel 2. The nominal switching frequency of
200kHz for the LM5642 corresponds to a switching period of 5µs. Channel 2 always turns its high-side switch on
2.5µs after Channel 1 Figure 30 (a). When the converter is synchronized to a frequency other than 200kHz, the
switching period is reduced or increased, while the fixed delay between Channel 1 and Channel 2 remains
constant. The phase difference between channels is therefore no longer 180°. At the extremes of the sync range,
the phase difference drops to 135° Figure 30 (b) and Figure 30 (c). The result of this lower phase difference is a
reduction in the maximum duty cycle of one channel that will not overlap the duty cycle of the other. As shown in
Input Capacitor Selection section, when the duty cycle D1 for Channel 1 overlaps the duty cycle D2 for Channel
2, the input rms current increases, requiring more input capacitors or input capacitors with higher ripple current
ratings. The new, reduced maximum duty cycle can be calculated by multiplying the sync frequency (in Hz) by
2.5x10-6 (the fixed delay in seconds). The same logic applies to the LM5642X. However the LM5642X has a
nominal switching frequency of 375kHz which corresponds to a period of 2.67µs. Therefore channel 2 of the
LM5642X always begins it's period after 1.33µs.
DMAX = FSYNC*2.5x10-6 (5)
At a sync frequency of 150 kHz, for example, the maximum duty cycle for Channel 1 that will not overlap
Channel 2 would be 37.5%. At 250 kHz, it is the duty cycle for Channel 2 that is reduced to a DMAX of 37.5%.
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
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FSW = 200 kHz
D1
D2
5 Ps
2.5 Ps
5 Ps
FSW = 150 kHz
D1
D2
6.67 Ps
6.67 Ps
2.5 Ps
FSW = 250 kHz
D1
D2
4 Ps
4 Ps
2.5 Ps
(a)
(b)
(c)
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
Figure 30. Period Fixed Delay Example
Component Selection
OUTPUT VOLTAGE SETTING
The output voltage for each channel is set by the ratio of a voltage divider as shown in Figure 31. The resistor
values can be determined by the following equation:
where
Vfb = 1.238V (6)
Although increasing the value of R1 and R2 will increase efficiency, this will also decrease accuracy. Therefore, a
maximum value is recommended for R2 in order to keep the output within .3% of Vnom. This maximum R2 value
should be calculated first with the following equation:
where
200nA is the maximum current drawn by FBx pin (7)
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48 12 16 20 24 28 32 36
VIN
VOUT
0
5
10
15
20
25
30
35
Operating Region
Vout
R2
FBx
GND R1
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
Figure 31. Output Voltage Setting
Example: Vnom = 5V, Vfb = 1.2364V, Ifbmax = 200nA.
(8)
Choose 60K
(9)
The Cycle Skip and Dropout modes of the LM5642 series regulate the minimum and maximum output
voltage/duty cycle that the converter can deliver. Both modes check the voltage at the COMP pin. Minimum
output voltage is determined by the Cycle Skip Comparator. This circuitry skips the high side FET ON pulse
when the COMP pin voltage is below 0.5V at the beginning of a cycle. The converter will continue to skip every
other pulse until the duty cycle (and COMP pin voltage) rise above 0.5V, effectively halving the switching
frequency.
Maximum output voltage is determined by the Dropout circuitry, which skips the low side FET ON pulse
whenever the COMP pin voltage exceeds the ramp voltage derived from the current sense. Up to three low side
pulses may be skipped in a row before a minimum on-time pulse must be applied to the low side FET.
Figure 32 shows the range of ouput voltage (for Io = 3A) with respect to input voltage that will keep the converter
from entering either Skip Cycle or Dropout mode.
For input voltages below 5.5V, VLIN5 must be connected to Vin through a small resistor (approximately 4.7
ohm). This will ensure that VLIN5 does not fall below the UVLO threshold.
Figure 32. Output Voltage Range
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Output Capacitor Selection
In applications that exhibit large, fast load current swings, the slew rate of such a load current transient will likely
be beyond the response speed of the regulator. Therefore, to meet voltage transient requirements during worst-
case load transients, special consideration should be given to output capacitor selection. The total combined
ESR of the output capacitors must be lower than a certain value, while the total capacitance must be greater
than a certain value. Also, in applications where the specification of output voltage regulation is tight and ripple
voltage must be low, starting from the required output voltage ripple will often result in fewer design iterations.
ALLOWED TRANSIENT VOLTAGE EXCURSION
The allowed output voltage excursion during a load transient (ΔVc_s) is:
where
±δ% is the output voltage regulation window
±ε% is the output voltage initial accuracy (10)
Example: Vnom = 5V, δ% = 7%, ε% = 3.4%, Vrip = 40mV peak to peak.
(11)
MAXIMUM ESR CALCULATION
Unless the rise and fall times of a load transient are slower than the response speed of the control loop, if the
total combined ESR (Re) is too high, the load transient requirement will not be met, no matter how large the
capacitance.
The maximum allowed total combined ESR is:
(12)
Since the ripple voltage is included in the calculation of ΔVc_s, the inductor ripple current should not be included
in the worst-case load current excursion. Simply use the worst-case load current excursion for ΔIc_s.
Example: ΔVc_s = 160 mV, ΔIc_s = 3A. Then Re_max = 53.3 m.
Maximum ESR criterion can be used when the associated capacitance is high enough, otherwise more
capacitors than the number determined by this criterion should be used in parallel.
MINIMUM CAPACITANCE CALCULATION
In a switch mode power supply, the minimum output capacitance is typically dictated by the load transient
requirement. If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed
value even if the maximum ESR requirement is met. The worst-case load transient is an unloading transient that
happens when the input voltage is the highest and when the current switching cycle has just finished. The
corresponding minimum capacitance is calculated as follows:
(13)
Notice it is already assumed the total ESR, Re, is no greater than Re_max, otherwise the term under the square
root will be a negative value. Also, it is assumed that L has already been selected, therefore the minimum L
value should be calculated before Cmin and after Re (see Inductor Selection below). Example: Re = 20 m,
Vnom = 5V, ΔVc_s = 160 mV, ΔIc_s = 3A, L = 8 µH
(14)
Generally speaking, Cmin decreases with decreasing Re, ΔIc_s, and L, but with increasing Vnom and ΔVc_s.
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L = 36 - 3.3
200kHz x 1.2A x3.3
36 = 12.5PH
1.2A = 36 - 3.3
L x 200kHz x3.3
36
Irip = 36 - 3.3
200kHz x 5x10-6
3.3
36
x= 3A
Lmin = 36 - 3.3
200kHz x 36
3.3 x 0.02
.060
x= 5PH
LM5642, LM5642X
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SNVS219K JUNE 2003REVISED APRIL 2013
Inductor Selection
The size of the output inductor can be determined from the desired output ripple voltage, Vrip, and the
impedance of the output capacitors at the switching frequency. The equation to determine the minimum
inductance value is as follows:
(15)
In the above equation, Re is used in place of the impedance of the output capacitors. This is because in most
cases, the impedance of the output capacitors at the switching frequency is very close to Re. In the case of
ceramic capacitors, replace Re with the true impedance at the switching frequency.
Example: Vin = 36V, Vo = 3.3V, VRIP = 60 mV, Re = 20 m, F = 200 kHz.
(16)
The actual selection process usually involves several iterations of all of the above steps, from ripple voltage
selection, to capacitor selection, to inductance calculations. Both the highest and the lowest input and output
voltages and load transient requirements should be considered. If an inductance value larger than Lmin is
selected, make sure that the Cmin requirement is not violated.
Priority should be given to parameters that are not flexible or more costly. For example, if there are very few
types of capacitors to choose from, it may be a good idea to adjust the inductance value so that a requirement of
3.2 capacitors can be reduced to 3 capacitors.
Since inductor ripple current is often the criterion for selecting an output inductor, it is a good idea to double-
check this value. The equation is:
(17)
Also important is the ripple content, which is defined by Irip /Inom. Generally speaking, a ripple content of less
than 50% is ok. Larger ripple content will cause too much power loss in the inductor.
Example: Vin = 36V, Vo = 3.3V, F = 200 kHz, L = 5 µH, 3A max IOUT
(18)
3A is 100% ripple which is too high.
In this case, the inductor should be reselected on the basis of ripple current.
Example: 40% ripple, 40% 3A = 1.2A
(19)
(20)
When choosing the inductor, the saturation current should be higher than the maximum peak inductor current
and the RMS current rating should be higher than the maximum load current.
Input Capacitor Selection
The fact that the two switching channels of the LM5642 are 180° out of phase will reduce the RMS value of the
ripple current seen by the input capacitors. This will help extend input capacitor life span and result in a more
efficient system. Input capacitors must be selected that can handle both the maximum ripple RMS current at
highest ambient temperature as well as the maximum input voltage. In applications in which output voltages are
less than half of the input voltage, the corresponding duty cycles will be less than 50%. This means there will be
no overlap between the two channels' input current pulses.
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 23
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LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
The equation for calculating the maximum total input ripple RMS current for duty cycles under 50% is:
where
I1 is maximum load current of Channel 1
I2 is the maximum load current of Channel 2
D1 is the duty cycle of Channel 1
D2 is the duty cycle of Channel 2 (21)
Example: Imax_1 = 3.6A, Imax_2 = 3.6A, D1 = 0.42, and D2 = 0.275
(22)
Choose input capacitors that can handle 1.66A ripple RMS current at highest ambient temperature. In
applications where output voltages are greater than half the input voltage, the corresponding duty cycles will be
greater than 50%, and there will be overlapping input current pulses. Input ripple current will be highest under
these circumstances. The input RMS current in this case is given by:
(23)
Where, again, I1 and I2 are the maximum load currents of channel 1 and 2, and D1 and D2 are the duty cycles.
This equation should be used when both duty cycles are expected to be higher than 50%.
If the LM5642 is being used with an external clock frequency other than 200kHz, or 375 kHz for the LM5642X,
the preceding equations for input rms current can still be used. The selection of the first equation or the second
changes because overlap can now occur at duty cycles that are less than 50%. From the EXTERNAL
FREQUENCY SYNC section, the maximum duty cycle that ensures no overlap between duty cycles (and hence
input current pulses) is:
DMAX = FSYNC*2.5 x 10-6 (24)
There are now three distinct possibilities which must be considered when selecting the equation for input rms
current. The following applies for the LM5642, and also the LM5642X by replacing 200 kHz with 375 kHz:
1. Both duty cycles D1and D2are less than DMAX. In this case, the first, simple equation can always be used.
2. One duty cycle is greater than DMAX and the other duty cycle is less than DMAX. In this case, the system
designer can take advantage of the fact that the sync feature reduces DMAX for one channel, but lengthens it
for the other channel. For FSYNC < 200kHz, D1is reduced to DMAX while D2actually increases to (1-DMAX).
For FSYNC > 200kHz, D2is reduced to DMAX while D1increases to (1-DMAX). By using the channel reduced to
DMAX for the lower duty cycle, and the channel that has been increased for the higher duty cycle, the first,
simple rms input current equation can be used.
3. Both duty cycles are greater than DMAX. This case is identical to a system at 200 kHz where either duty cycle
is 50% or greater. Some overlap of duty cycles is specified, and hence the second, more complicated rms
input current equation must be used.
Input capacitors must meet the minimum requirements of voltage and ripple current capacity. The size of the
capacitor should then be selected based on hold up time requirements. Bench testing for individual applications
is still the best way to determine a reliable input capacitor value. Input capacitors should always be placed as
close as possible to the current sense resistor or the drain of the top FET. When high ESR capacitors such as
tantalum are used, a 1µF ceramic capacitor should be added as closely as possible to the high-side FET drain
and low-side FET source.
24 Submit Documentation Feedback Copyright © 2003–2013, Texas Instruments Incorporated
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SNVS219K JUNE 2003REVISED APRIL 2013
MOSFET Selection
BOTTOM FET SELECTION
During normal operation, the bottom FET is switching on and off at almost zero voltage. Therefore, only
conduction losses are present in the bottom FET. The most important parameter when selecting the bottom FET
is the on-resistance (RDS-ON). The lower the on-resistance, the lower the power loss. The bottom FET power loss
peaks at maximum input voltage and load current. The equation for the maximum allowed on-resistance at room
temperature for a given FET package, is:
where
Tj_max is the maximum allowed junction temperature in the FET
Ta_max is the maximum ambient temperature
Rθja is the junction-to-ambient thermal resistance of the FET
TC is the temperature coefficient of the on-resistance which is typically in the range of 4000ppm/°C (25)
If the calculated RDS-ON (MAX) is smaller than the lowest value available, multiple FETs can be used in parallel.
This effectively reduces the Imax term in the above equation, thus reducing RDS-ON. When using two FETs in
parallel, multiply the calculated RDS-ON (MAX) by 4 to obtain the RDS-ON (MAX) for each FET. In the case of three
FETs, multiply by 9.
(26)
If the selected FET has an Rds value higher than 35.3, then two FETs with an RDS-ON less than 141 m(4 x
35.3 m) can be used in parallel. In this case, the temperature rise on each FET will not go to Tj_max because
each FET is now dissipating only half of the total power.
TOP FET SELECTION
The top FET has two types of losses: switching loss and conduction loss. The switching losses mainly consist of
crossover loss and losses related to the low-side FET body diode reverse recovery. Since it is rather difficult to
estimate the switching loss, a general starting point is to allot 60% of the top FET thermal capacity to switching
losses. The best way to precisely determine switching losses is through bench testing. The equation for
calculating the on resistance of the top FET is thus:
(27)
Example: Tj_max = 100°C, Ta_max = 60°C, Rqja = 60°C/W, Vin_min = 5.5V, Vnom = 5V, and Iload_max = 3.6A.
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM5642 LM5642X
-20dB/dec
(fp1 is at zero frequency)
-20dB/dec
FREQUENCY
GAIN (dB)
B
fz1 fz2
fp2
10 100 1
k10
k100
k1M
FREQUENCY
(Hz)
-60
-40
-20
0
20
GAIN
(dB)
0
-45
-90
-135
-180
PHASE (°)
Asymptoti
c
Gain
Phas
e
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
(28)
When using FETs in parallel, the same guidelines apply to the top FET as apply to the bottom FET.
Loop Compensation
The general purpose of loop compensation is to meet static and dynamic performance requirements while
maintaining stability. Loop gain is what is usually checked to determine small-signal performance. Loop gain is
equal to the product of control-output transfer function and the feedback transfer function (the compensation
network transfer function). Generally speaking it is desirable to have a loop gain slope that is roughly -20dB
/decade from a very low frequency to well beyond the crossover frequency. The crossover frequency should not
exceed one-fifth of the switching frequency. The higher the bandwidth, the faster the load transient response
speed will be. However, if the duty cycle saturates during a load transient, further increasing the small signal
bandwidth will not help. Since the control-output transfer function usually has very limited low frequency gain, it is
a good idea to place a pole in the compensation at zero frequency, so that the low frequency gain will be
relatively large. A large DC gain means high DC regulation accuracy (i.e. DC voltage changes little with load or
line variations). The rest of the compensation scheme depends highly on the shape of the control-output plot.
Figure 33. Control-Output Transfer Function
As shown in Figure 33, the control-output transfer function consists of one pole (fp), one zero (fz), and a double
pole at fn (half the switching frequency). The following can be done to create a -20dB /decade roll-off of the loop
gain: Place the first pole at 0Hz, the first zero at fp, the second pole at fz, and the second zero at fn. The
resulting feedback transfer function is shown in Figure 34.
Figure 34. Feedback Transfer Function
26 Submit Documentation Feedback Copyright © 2003–2013, Texas Instruments Incorporated
Product Folder Links: LM5642 LM5642X
fP = 1
2SRO CO+1 - D - .5
2SfLCO
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
The control-output corner frequencies, and thus the desired compensation corner frequencies, can be
determined approximately by the following equations:
(29)
(30)
Since fp is determined by the output network, it will shift with loading (Ro). It is best to use a minimum Iout value
of approximately 100mA when determining the maximum Ro value.
Example: Re = 20 m, Co = 100 uF, Romax = 5V/100 mA = 50:
(31)
(32)
First determine the minimum frequency (fpmin) of the pole across the expected load range, then place the first
compensation zero at or below that value. Once fpmin is determined, Rc1 should be calculated using:
where
B is the desired gain in V/V at fp (fz1)
gm is the transconductance of the error amplifier
R1 and R2 are the feedback resistors (33)
A gain value around 10dB (3.3v/v) is generally a good starting point.
Example: B = 3.3v/v, gm = 650m, R1 = 20 kK, R2 = 60.4 k:
(34)
Bandwidth will vary proportional to the value of Rc1. Next, Cc1 can be determined with the following equation:
(35)
Example: fpmin = 995 Hz, Rc1 = 20 k:
(36)
The compensation network (Figure 35) will also introduce a low frequency pole which will be close to 0 Hz.
A second pole should also be placed at fz. This pole can be created with a single capacitor Cc2 and a shorted
Rc2 (see Figure 35). The minimum value for this capacitor can be calculated by:
(37)
Cc2 may not be necessary, however it does create a more stable control loop. This is especially important with
high load currents and in current sharing mode.
Example: fz = 80 kHz, Rc1 = 20 k:
(38)
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 27
Product Folder Links: LM5642 LM5642X
Vo
Vc
R2
R1
gm
compensation
network
CC1 CC2
RC1 RC2
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
A second zero can also be added with a resistor in series with Cc2. If used, this zero should be placed at fn,
where the control to output gain rolls off at -40dB/dec. Generally, fn will be well below the 0dB level and thus will
have little effect on stability. Rc2 can be calculated with the following equation:
(39)
Figure 35. Compensation Network
PCB Layout Considerations
To produce an optimal power solution with the LM5642 series, good layout and design of the PCB are as
important as the component selection. The following are several guidelines to aid in creating a good layout.
KELVIN TRACES FOR SENSE LINES
When using the current sense resistor to sense the load current connect the KS pin using a separate trace to
VIN, as close as possible to the current-sense resistor. The RSNS pin should be connected using a separate
trace to the low-side of the current sense resistor. The traces should be run parallel to each other to give
common mode rejection. Although it can be difficult in a compact design, these traces should stay away from the
output inductor and switch node if possible, to avoid coupling stray flux fields. When a current-sense resistor is
not used the KS pin should be connected as close as possible to the drain node of the upper MOSFET and the
RSNS pin should be connected as close as possible to the source of the upper MOSFET using Kelvin traces. To
further help minimize noise pickup on the sense lines is to use RC filtering on the KS and RSNS pins.
SEPARATE PGND AND SGND
Good layout techniques include a dedicated ground plane, usually on an internal layer. Signal level components
like the compensation and feedback resistors should be connected to a section of this internal SGND plane. The
SGND section of the plane should be connected to the power ground at only one point. The best place to
connect the SGND and PGND is right at the PGND pin..
MINIMIZE THE SWITCH NODE
The plane that connects the power FETs and output inductor together radiates more EMI as it gets larger. Use
just enough copper to give low impedance to the switching currents, preferably in the form of a wide, but short,
trace run.
LOW IMPEDANCE POWER PATH
The power path includes the input capacitors, power FETs, output inductor, and output capacitors. Keep these
components on the same side of the PCB and connect them with thick traces or copper planes (shapes) on the
same layer. Vias add resistance and inductance to the power path, and have relatively high impedance
connections to the internal planes. If high switching currents must be routed through vias and/or internal planes,
use multiple vias in parallel to reduce their resistance and inductance. The power components should be kept
close together. The longer the paths that connect them, the more they act as antennas, radiating unwanted EMI.
Please see AN-1229 (literature number SNVA054) for further PCB layout considerations.
28 Submit Documentation Feedback Copyright © 2003–2013, Texas Instruments Incorporated
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SNVS219K JUNE 2003REVISED APRIL 2013
Table 1. Bill Of Materials for Figure 3 24V to 1.8, 3.3V LM5642
ID Part Number Type Size Parameters Qty Vendor
U1 LM5642 Dual TSSOP-28 1 TI
Synchronous
Controller
Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay
Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay
D3 BAS40-06 Schottky Diode SOT-23 40V 1 Vishay
L1 RLF12560T-4R2N100 Inductor 12.5x12.5x 6mm 4.2µH, 7m10A 1 TDK
L2 RLF12545T-100M5R1 Inductor 12.5x12.5x 4.5mm 10µH, 12m5.1A 1 TDK
C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK
C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 3 Vishay
C15
C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK
C6, C16 C5750X5R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK
C9, C23 6TPD330M Capacitor 7.3x4.3x 3.8mm 330µF, 6.3V, 10m2 Sanyo
C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay
C13
C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay
C19 VJ1206Y822KXXAT Capacitor 1206 8.2nF 10% 1 Vishay
C20 VJ1206Y153KXXAT Capacitor 1206 15nF 10% 1 Vishay
C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK
R1 CRCW1206123J Resistor 1206 12k5% 1 Vishay
R2, R6, R14, CRCW1206100J Resistor 1206 1005% 1 Vishay
R16
R13 CRCW1206682J Resistor 1206 6.8k12% 1 Vishay
R7, R15 WSL-2512 .010 1% Resistor 2512 10m1W 2 Vishay
R8, R9, R12, CRCW1206000Z Resistor 1206 08 Vishay
R17, R18, R21,
R31, R32
R10 CRCW12062261F Resistor 1206 2.26k1% 1 Vishay
R23 CRCW12068451F Resistor 1206 8.45k1% 1 Vishay
R24 CRCW12061372F Resistor 1206 13.7k1% 1 Vishay
R11, R20 CRCW12064991F Resistor 1206 4.99k1% 2 Vishay
R19 CRCW12068251F Resistor 1206 8.25k1% 1 Vishay
R27 CRCW12064R7J Resistor 1206 4.75% 1 Vishay
R28 CRCW1206224J Resistor 1206 220k5% 1 Vishay
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 29
Product Folder Links: LM5642 LM5642X
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
Table 2. Bill of Materials for Figure 4 30V to 1.8V, 20A LM5642
ID Part Number Type Size Parameters Qty Vendor
U1 LM5642 Dual TSSOP-28 1 TI
Synchronou
s Controller
Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay
Q2, Q3, Q5, Q6 Si4470DY N-MOSFET SO-8 60V 4 Vishay
D3 BAS40-06 Schottky SOT-23 40V 1 Vishay
Diode
L1,L2 RLF12560T-2R7N110 Inductor 12.5x12.5x 6mm 2.7µH,4.5m11.5A 2 TDK
C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK
C10, C24, C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 3 TDK
C6, C16, C28, C5750X5R1H106M Capacitor 2220 10µF 50V, 2.8A 4 TDK
C30
C9, C23 16MV1000WX Capacitor 10mm D20mm H 1000µF, 16V, 22m2 Sanyo
C2, C13 VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 2 Vishay
C11 VJ1206Y223KXXAT Capacitor 1206 22nF, 25V 1 Vishay
C7,C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay
C19 VJ1206Y273KXXAT Capacitor 1206 27nF 10% 1 Vishay
C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK
R1, R13 CRCW1206123J Resistor 1206 16.9k1% 1 Vishay
R2, R6, R14, CRCW1206100J Resistor 1206 1005% 1 Vishay
R16
R7, R15 WSL-2512 .010 1% Resistor 2512 10m1W 2 Vishay
R8, R9, R12, CRCW1206000Z Resistor 1206 08 Vishay
R17, R18, R21,
R31, R32
R10 CRCW12062261F Resistor 1206 2.26k1% 1 Vishay
R11 CRCW12064991F Resistor 1206 4.99k1% 1 Vishay
R23 CRCW12061152F Resistor 1206 11.5k1% 1 Vishay
R27 CRCW12064R7J Resistor 1206 4.75% 1 Vishay
R28 CRCW1206224J Resistor 1206 220k5% 1 Vishay
30 Submit Documentation Feedback Copyright © 2003–2013, Texas Instruments Incorporated
Product Folder Links: LM5642 LM5642X
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
Table 3. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=1.5V,1.8V, 5A LM5642X
ID Part Number Type Size Parameters Qty Vendor
U1 LM5642X Dual TSSOP-28 1 TI
Synchronous
Controller
Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay
Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay
D3 BA54A Schottky Diode SOT-23 30V 1 Vishay
L1, L2 RLF12545T-4R2N100 Inductor 12.5x12.5x 4.5mm 4.2µH, 7m6.5A 2 TDK
C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK
C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 4 Vishay
C15
C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK
C6, C28 C5750X7R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK
C9, C23 C4532X7R0J107M Capacitor 1812 100µF, 6.3V, 1m2 TDK
C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay
C13
C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay
C18, C20 VJ1206Y473KXXAT Capacitor 1206 47nF 10% 2 Vishay
C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK
R1, R13 CRCW12061912F Resistor 1206 19.1k1% 2 Vishay
R2, R6, R14, CRCW1206100J Resistor 1206 1005% 1 Vishay
R16
R7, R15 WSL-1206 .020 1% Resistor 1206 20m1W 2 Vishay
R8, R9, R12, CRCW1206000Z Resistor 1206 08 Vishay
R17, R18, R21,
R31, R32
R10, R19 CRCW12061001F Resistor 1206 1k1% 2 Vishay
R11 CRCW12062611F Resistor 1206 2.61k1% 1 Vishay
R20 CRCW12062321F Resistor 1206 2.32k1% 1 Vishay
R22, R24 CRCW12063011F Resistor 1206 3.01k1% 2 Vishay
R27 CRCW12064R7J Resistor 1206 4.75% 1 Vishay
R28 CRCW1206224J Resistor 1206 220k5% 1 Vishay
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 31
Product Folder Links: LM5642 LM5642X
LM5642, LM5642X
SNVS219K JUNE 2003REVISED APRIL 2013
www.ti.com
Table 4. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=3.3V,5V, 5A LM5642X
ID Part Number Type Size Parameters Qty Vendor
U1 LM5642X Dual TSSOP-28 1 TI
Synchronous
Controller
Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay
Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay
D3 BA54A Schottky Diode SOT-23 30V 1 Vishay
L1, L2 RLF12545T-5R6N6R1 Inductor 12.5x12.5x 4.5mm 5.6µH, 9m6.1A 2 TDK
C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK
C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 4 Vishay
C15
C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK
C6, C28 C5750X7R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK
C9, C23 C4532X7R0J107M Capacitor 1812 100µF, 6.3V, 1m2 TDK
C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay
C13
C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay
C18, C20 VJ1206Y393KXXAT Capacitor 1206 39nF 10% 2 Vishay
C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK
R1, R13 CRCW12061912F Resistor 1206 19.1k1% 2 Vishay
R2, R6, R14, CRCW1206100J Resistor 1206 1005% 1 Vishay
R16
R7, R15 WSL-1206 .020 1% Resistor 1206 20m1W 2 Vishay
R8, R9, R12, CRCW1206000Z Resistor 1206 08 Vishay
R17, R18, R21,
R31, R32
R10, R19 CRCW12061002F Resistor 1206 10k1% 2 Vishay
R11 CRCW12066191F Resistor 1206 6.19k1% 1 Vishay
R20 CRCW12063321F Resistor 1206 3.32k1% 1 Vishay
R22, R24 CRCW12063831F Resistor 1206 3.83k1% 2 Vishay
R27 CRCW12064R7J Resistor 1206 4.75% 1 Vishay
R28 CRCW1206224J Resistor 1206 220k5% 1 Vishay
32 Submit Documentation Feedback Copyright © 2003–2013, Texas Instruments Incorporated
Product Folder Links: LM5642 LM5642X
LM5642, LM5642X
www.ti.com
SNVS219K JUNE 2003REVISED APRIL 2013
REVISION HISTORY
Changes from Revision J (April 2013) to Revision K Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 32
Copyright © 2003–2013, Texas Instruments Incorporated Submit Documentation Feedback 33
Product Folder Links: LM5642 LM5642X
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM5642MH/NOPB ACTIVE HTSSOP PWP 28 48 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM LM5642
MH
LM5642MHX/NOPB ACTIVE HTSSOP PWP 28 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM LM5642
MH
LM5642MTC NRND TSSOP PW 28 48 TBD Call TI Call TI -40 to 125 LM5642
MTC
LM5642MTC/NOPB ACTIVE TSSOP PW 28 48 Green (RoHS
& no Sb/Br) SN Level-3-260C-168 HR -40 to 125 LM5642
MTC
LM5642MTCX NRND TSSOP PW 28 2500 TBD Call TI Call TI -40 to 125 LM5642
MTC
LM5642MTCX/NOPB ACTIVE TSSOP PW 28 2500 Green (RoHS
& no Sb/Br) SN Level-3-260C-168 HR -40 to 125 LM5642
MTC
LM5642XMH/NOPB ACTIVE HTSSOP PWP 28 48 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM5642
XMH
LM5642XMHX/NOPB ACTIVE HTSSOP PWP 28 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM5642
XMH
LM5642XMT/NOPB ACTIVE TSSOP PW 28 48 Green (RoHS
& no Sb/Br) SN Level-3-260C-168 HR -40 to 125 LM5642
XMT
LM5642XMTX/NOPB ACTIVE TSSOP PW 28 2500 Green (RoHS
& no Sb/Br) SN Level-3-260C-168 HR -40 to 125 LM5642
XMT
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 2
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM5642MHX/NOPB HTSSOP PWP 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
LM5642MTCX TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
LM5642MTCX/NOPB TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
LM5642XMHX/NOPB HTSSOP PWP 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
LM5642XMTX/NOPB TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 29-Sep-2019
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM5642MHX/NOPB HTSSOP PWP 28 2500 367.0 367.0 35.0
LM5642MTCX TSSOP PW 28 2500 367.0 367.0 38.0
LM5642MTCX/NOPB TSSOP PW 28 2500 367.0 367.0 38.0
LM5642XMHX/NOPB HTSSOP PWP 28 2500 367.0 367.0 35.0
LM5642XMTX/NOPB TSSOP PW 28 2500 367.0 367.0 38.0
PACKAGE MATERIALS INFORMATION
www.ti.com 29-Sep-2019
Pack Materials-Page 2
www.ti.com
PACKAGE OUTLINE
C
TYP
6.6
6.2
1.1 MAX
26X 0.65
28X 0.30
0.19
2X
8.45
TYP
0.20
0.09
0 - 8
0.10
0.02
5.65
5.25
3.15
2.75
(1)
0.25
GAGE PLANE
0.7
0.5
A
NOTE 3
9.8
9.6
B
NOTE 4
4.5
4.3
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm, per side.
4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm, per side.
5. Reference JEDEC registration MO-153, variation AET.
PowerPAD is a trademark of Texas Instruments.
TM
128
0.1 C A B
15
14
PIN 1 ID
AREA
SEATING PLANE
0.1 C
SEE DETAIL A
DETAIL A
TYPICAL
SCALE 1.800
THERMAL
PAD
www.ti.com
EXAMPLE BOARD LAYOUT
0.05 MAX
ALL AROUND
0.05 MIN
ALL AROUND
28X (1.3)
(6.1)
28X (0.45)
(0.9) TYP
28X (1.5)
28X (0.45)
26X
(0.65)
(3)
(3.4)
NOTE 9
(5.5)
SOLDER
MASK
OPENING
(9.7)
(1.3)
(1.3) TYP
(5.8)
( ) TYP
VIA
0.2
(0.65) TYP
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
SOLDER MASK
DEFINED PAD
LAND PATTERN EXAMPLE
SCALE:6X
HV / ISOLATION OPTION
0.9 CLEARANCE CREEPAGE
OTHER DIMENSIONS IDENTICAL TO IPC-7351
TM
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
METAL
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
SYMM
SYMM
SEE DETAILS
1
14 15
28
METAL COVERED
BY SOLDER MASK
SOLDER
MASK
OPENING
IPC-7351 NOMINAL
0.65 CLEARANCE CREEPAGE
www.ti.com
EXAMPLE STENCIL DESIGN
28X (1.3)
28X (0.45)
(6.1)
28X (1.5)
28X (0.45)
26X (0.65)
(3)
(5.5)
BASED ON
0.127 THICK
STENCIL
(5.8)
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
2.66 X 4.770.178
2.88 X 5.160.152
3.0 X 5.5 (SHOWN)0.127
3.55 X 6.370.1
SOLDER STENCIL
OPENING
STENCIL
THICKNESS
SEE TABLE FOR
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
TM
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE AREA
SCALE:6X
HV / ISOLATION OPTION
0.9 CLEARANCE CREEPAGE
OTHER DIMENSIONS IDENTICAL TO IPC-7351
SYMM
SYMM
1
14 15
28
BASED ON
0.127 THICK
STENCIL BY SOLDER MASK
METAL COVERED
IPC-7351 NOMINAL
0.65 CLEARANCE CREEPAGE
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