1
LTC1622
Low Input Voltage
Current Mode Step-Down
DC/DC Controller
The LTC
®
1622 is a constant frequency current mode step-
down DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltage feature that shuts the LTC1622 down when
the input voltage falls below 2V.
The LTC1622 boasts a ±1.9% output voltage accuracy and
consumes only 350µA of quiescent current. For applica-
tions where efficiency is a prime consideration and the
load current varies from light to heavy, the LTC1622 can
be configured for Burst Mode
TM
operation. Burst Mode
operation enhances low current efficiency and extends
battery run time. Burst Mode operation is inhibited during
synchronization or when the SYNC/MODE pin is pulled low
to reduce noise and possible RF interference.
High constant operating frequency of 550kHz allows the
use of a small inductor. The device can also be synchro-
nized up to 750kHz for special applications. The high
frequency operation and the available 8-lead MSOP pack-
age create a high performance solution in an extremely
small amount of PCB area.
To further maximize the life of the battery source, the
P-channel MOSFET is turned on continuously in dropout
(100% duty cycle). In shutdown, the device draws a mere
15µA.
■
1- or 2-Cell Li-Ion Powered Applications
■
Cellular Telephones
■
Wireless Modems
■
Portable Computers
■
Distributed 3.3V, 2.5V or 1.8V Power Systems
■
Scanners
■
Battery-Powered Equipment
■
High Efficiency
■
Constant Frequency 550kHz Operation
■
V
IN
Range: 2V to 10V
■
Multiampere Output Currents
■
OPTI-LOOP
TM
Compensation Minimizes C
OUT
■
Selectable, Burst Mode
Operation
■
Low Dropout Operation: 100% Duty Cycle
■
Synchronizable up to 750kHz
■
Current Mode Operation for Excellent Line and Load
Transient Response
■
Low Quiescent Current: 350µA
■
Shutdown Mode Draws Only 15µA Supply Current
■
±1.9% Reference Accuracy
■
Available in 8-Lead MSOP
Burst Mode and OPTI-LOOP are a trademarks of Linear Technology Corporation.
FEATURES
DESCRIPTIO
U
APPLICATIO S
U
TYPICAL APPLICATIO
U
, LTC and LT are registered trademarks of Linear Technology Corporation.
Figure 1. High Efficiency Step-Down Converter
2
4
1
8
7
5
6
3
SYNC/MODE
RUN/SS
L1
4.7µH
R2
0.03Ω
D1
IR10BQ015
Si3443DV
R3
159k
V
OUT
2.5V
1.5A
R1
10k
C3
220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: INTERNATIONAL RECTIFIER IR10BQ015
V
IN
2.5V TO 8.5V
R4
75k
1622 F01a
470pF
C1
10µF
10V
LTC1622
V
IN
V
FB
C2
47µF
6V
+
SENSE
–
PDRV
I
TH
GND
L1: MURATA LQN6C-4R7
R2: DALE WSL-1206 0-03Ω
LOAD CURRENT (mA)
EFFICIENCY (%)
100
90
80
70
60
50
40 1 100 1000 5000
1622 F01b
10
V
OUT
= 2.5V
R
SENSE
= 0.03Ω
V
IN
= 3.3V
V
IN
= 6V
V
IN
= 8.4V
V
IN
= 4.2V
Efficiency vs Load Current with
Burst Mode Operation Enabled
2
LTC1622
ABSOLUTE MAXIMUM RATINGS
W
WW
U
Input Supply Voltage (V
IN
).........................–0.3V to 10V
RUN/SS Voltage .......................................–0.3V to 2.4V
SYNC/MODE Voltage ................................. –0.3V to V
IN
SENSE
–
Voltage .......................................... 2.4V to V
IN
PDRV Peak Output Current (<10µs) ......................... 1A
Storage Ambient Temperature Range ... –65°C to 150°C
Operating Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ........................................... –45°C to 85°C
Junction Temperature (Note 2).............................125°C
Lead Temperature (Soldering, 10 sec)..................300°C
PACKAGE/ORDER INFORMATION
W
UU
T
JMAX
= 125°C, θ
JA
= 150°C/ W
S8 PART MARKING
ORDER PART
NUMBER
LTC1622CS8
LTC1622IS8
1622
1622I
ORDER PART
NUMBER
Consult factory for Military grade parts.
1
2
3
4
SENSE
–
I
TH
V
FB
RUN/SS
8
7
6
5
V
IN
PDRV
GND
SYNC/MODE
TOP VIEW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
T
JMAX
= 125°C, θ
JA
= 250°C/W
LTC1622CMS8
MS8 PART MARKING
LTDB
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
I
VFB
Feedback Current (Note 3) V
FB
= 0.8V 10 70 nA
V
FB
Regulated Feedback Voltage (Note 3) Commercial Grade ●0.785 0.8 0.815 V
(Note 3) Industrial Grade ●0.780 0.8 0.820 V
V
OVL
Output Overvoltage Lockout Referenced to Nominal V
OUT
41625 %
∆V
OSENSE
Reference Voltage Line Regulation V
IN
= 4.2V to 8.5V (Note 3) 0.04 0.08 %/V
V
LOADREG
Output Voltage Load Regulation Measured in Servo Loop; V
ITH
= 0.2V to 0.625V 0.3 0.5 %
Measured in Servo Loop; V
ITH
= 0.9V to 0.625V –0.3 –0.5 %
I
S
Input DC Supply Current (Note 4)
Burst Mode Inhibited V
IN
= 2.3V 450 µA
Sleep Mode V
ITH
= 0V, V
SYNC/MODE
= 2.4V 350 400 µA
Shutdown V
RUN/SS
= 0V 15 30 µA
Shutdown V
RUN/SS
= 0V, V
IN
= V
UVLO
– 0.1V 4 10 µA
V
RUN/SS
RUN/SS Threshold Commercial Grade ●0.4 0.7 0.9 V
Industrial Grade ●0.3 0.7 1.0 V
I
RUN/SS
Soft-Start Current Source V
RUN/SS
= 0V 1 2.5 5 µA
f
OSC
Oscillator Frequency V
FB
= 0.8V 475 550 625 kHz
V
FB
= 0V 75 110 140 kHz
V
SYNC/MODE
SYNC/MODE Threshold V
SYNC/MODE
Ramping Down 1 1.5 V
V
UVLO
Undervoltage Lockout V
IN
Ramping Down ●1.55 1.92 2.3 V
V
IN
Ramping Up 1.97 2.36 V
TOP VIEW
S8 PACKAGE
8-LEAD PLASTIC SO
1
2
3
4
8
7
6
5
SENSE
–
I
TH
V
FB
RUN/SS
V
IN
PDRV
GND
SYNC/MODE
(Note 1)
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V
3
LTC1622
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
PDRV t
r
Gate Drive Rise Time C
LOAD
= 3000pF 80 140 ns
PDRV t
f
Gate Drive Fall Time C
LOAD
= 3000pF 100 140 ns
∆V
SENSE(MAX)
Maximum Current Sense Voltage ●80 110 140 mV
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: T
J
is calculated from the ambient temperature T
A
and power
dissipation P
D
according to the following formula:
LTC1622CS8; T
J
= T
A
+ (P
D
• 150°C/W),
LTC1622CMS8; T
J
= T
A
+ (P
D
• 250°C/W)
Note 3: The LTC1622 is tested in a feedback loop that servos V
FB
to the
feedback point for the error amplifier (V
ITH
= 0.8V).
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
TYPICAL PERFORMANCE CHARACTERISTICS
UW
SUPPLY VOLTAGE (V)
23
SHUTDOWN CURRENT (µA)
10
1622 G01
456789
45
40
35
30
25
20
15
10
5
0
DUTY CYCLE (%)
20 30
TRIP VOLTAGE (mV)
100
1622 G03
40 50 60 70 80 90
110
100
90
80
70
60
50
40
30
V
IN
= 4.2V
UNSYNC
TEMPERATURE (°C)
–55 –35
UNDERVOLTAGE LOCKOUT VOLTAGE (V)
125
1622 G06
–15 5 25 45 65 85 105
2.10
2.05
2.00
1.95
1.90
1.85
1.80
1.75
RUN/SS Current vs Supply Voltage
SUPPLY VOLTAGE (V)
2
SOFT-START CURRENT (µA)
8
1622 G02
46 10
3.50
3.00
2.50
2.00
1.50
1.00 3579
TEMPERATURE (°C)
–55 –35
REFERENCE VOLTAGE (V)
125
1622 G05
–15 5 25 45 65 85 105
0.810
0.805
0.800
0.795
0.790
0.785
0.780
0.775
V
IN
= 4.2V
Shutdown Current
vs Supply Voltage
Maximum Current Sense Voltage
vs Duty Cycle
Undervoltage Lockout Voltage
vs Temperature
Reference Voltage
vs Temperature
TEMPERATURE (°C)
–55 –35
NORMALIZED FREQUENCY (%)
125
1622 G04
–15 5 25 45 65 85 105
10.0
7.5
5.0
2.5
0
–2.5
–5.0
–7.5
–10.0
V
IN
= 4.2V
Normalized Oscillator Frequency
vs Temperature
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V
4
LTC1622
TYPICAL PERFORMANCE CHARACTERISTICS
UW
PIN FUNCTIONS
UUU
SENSE
–
(Pin 1): The Negative Input to the Current Com-
parator.
I
TH
(Pin 2): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 1.2V.
V
FB
(Pin 3): Receives the feedback voltage from an exter-
nal resistive divider across the output capacitor.
RUN/SS (Pin 4): Combination of Soft-Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp time to full output current. The time is approximately
0.45s/µF. Forcing this pin below 0.4V causes all circuitry
to be shut down.
SYNC/MODE (Pin 5): This pin performs three functions.
Greater than 2V on this pin allows Burst Mode operation
at low load currents, while grounding or applying a clock
signal on this pin defeats Burst Mode operation. An
external clock between 625kHz and 750kHz applied to this
pin forces the LTC1622 to operate at the external clock
frequency.
Do not attempt to synchronize below 625kHz
.
Pin 5 has an internal 1µA pull-up current source.
GND (Pin 6): Ground Pin.
PDRV (PIN 7): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to V
IN
.
V
IN
(Pin 8): Main Supply Pin. Must be closely decoupled
to ground Pin 6.
LOAD CURRENT (mA)
1
EFFICIENCY (%)
100
90
80
70
60
50
40 10 100
1622 G07
1000
V
OUT
= 2.5V
R
SENSE
= 0.03Ω
V
IN
= 3.3V
V
IN
= 4.2V
V
IN
= 8.4V
V
IN
= 6V
Efficiency vs Load Current for
Figure 1 with Burst Mode
Operation Defeated
Load Step Transient Response
Burst Enabled
ILOAD = 50mA TO 1.2A
V
IN
= 4.2V
1622 G08
ILOAD = 50mA TO 1.2A
V
IN
= 4.2V
1622 G09
Load Step Transient Response
Burst Inhibited
100mV/DIV
100mV/DIV
5
LTC1622
FUNCTIONAL DIAGRA
UU
W
OPERATIO
U
Main Control Loop
The LTC1622 is a constant frequency current mode switch-
ing regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the R
S
latch (R
S1
) and turned off when
the current comparator (I
COMP
) resets the latch. The peak
inductor current at which I
COMP
resets the R
S
latch is
controlled by the voltage on the I
TH
pin, which is the output
of the error amplifier EA. An external resistive divider
connected between V
OUT
and ground allows EA to receive
an output feedback voltage V
FB
. When the load current
increases, it causes a slight decrease in V
FB
relative to the
0.8V reference, which in turn causes the I
TH
voltage to
increase until the average inductor current matches the
new load current.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 2.5µA
(Refer to Functional Diagram)
current source to charge up the soft-start capacitor C
SS
.
When C
SS
reaches 0.7V, the main control loop is enabled
with the I
TH
voltage clamped at approximately 5% of its
maximum value. As C
SS
continues to charge, I
TH
is gradu-
ally released allowing normal operation to resume.
Comparator OV guards against transient overshoots
>16% by turning off the P-channel power MOSFET and
keeping it off until the fault is removed.
Burst Mode Operation
The LTC1622 can be enabled to go into Burst Mode
operation at low load currents simply by leaving the SYNC/
MODE pin open or connecting it to a voltage of at least 2V.
In this mode, the peak current of the inductor is set as if
V
ITH
= 0.36V (at low duty cycles) even though the voltage
at the I
TH
pin is at lower value. If the inductor’s average
current is greater than the load requirement, the voltage at
+
–
+
–
+
–+
–
+
–
+–
BURST DEFEAT
1µA
5
SYNC/
MODE
3
6
4
2S
RQ
BURST
R
S1
0.12V
SLEEP
SENSE
–
EN
1
0.36V
8
PDRV
OV
1622 BD
7
GND
V
REF
0.8V
RUN/SS
V
IN
SLOPE
COMP
Y = “0” ONLY WHEN X IS A CONSTANT “1”
OTHERWISE Y = “1”
FREQ
SHIFT
OSC
2.5µAg
m
= 0.5m
EA
Ω
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
RUN/
SOFT-START
V
REF
+ 0.12V
SHUTDOWN
V
FB
0.3V
V
IN
V
IN
V
IN
V
CC
ICOMP
I
TH
X
Y
V
IN
0.8V
REFERENCE
UVLO
TRIP = 1.97V
0.8V
V
REF
6
LTC1622
the I
TH
pin will drop. When the I
TH
voltage goes below
0.12V, the sleep signal goes high, turning off the external
MOSFET. The sleep signal goes low when the I
TH
voltage
rises above 0.22V and the LTC1622 resumes normal
operation. The next oscillator cycle will turn the external
MOSFET on and the switching cycle repeats.
Frequency Synchronization
The LTC1622 can be externally driven by a TTL/CMOS
compatible clock signal up to 750kHz.
Do not
synchronize
the LTC1622 below its maximum default operating fre-
quency of 625kHz as this may cause abnormal operation
and an undesired frequency spectrum. The LTC1622 is
synchronized to the rising edge of the clock. The external
clock pulse width must be at least 100ns and not more
than the period minus 200ns.
Synchronization is inhibited when the feedback voltage is
below 0.3V. This is to prevent inductor current buildup
under short-circuit conditions. Burst Mode operation is
deactivated when the LTC1622 is externally driven by a
clock.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the P-channel MOSFET will remain on for more than one
oscillator cycle since the inductor current has not ramped
up to the threshold set by EA. Further reduction in input
supply voltage will eventually cause the P-channel MOSFET
to be turned on 100%, i.e., DC. The output voltage will then
be determined by the input voltage minus the voltage drop
across the MOSFET, the sense resistor and the inductor.
Undervoltage Lockout
To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1622. When the input supply voltage
drops below 2V, the P-channel MOSFET and all circuitry is
turned off except the undervoltage block, which draws
only several microamperes.
OPERATIO
U
(Refer to Functional Diagram)
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator will be reduced to about 110kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s fre-
quency will gradually increase to its nominal value when
the feedback voltage increases above 0.65V. Note that
synchronization is inhibited until the feedback voltage
goes above 0.3V.
Overvoltage Protection
As a further protection, the overvoltage comparator in the
LTC1622 will turn the external MOSFET off when the
feedback voltage has risen 16% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 35mV.
Slope Compensation and Peak Inductor Current
The inductor’s peak current is determined by:
IV
R
PK ITH
SENSE
=
()
10
when the LTC1622 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak induc-
tor current. The amount of reduction is given by the curves
in Figure 2.
DUTY CYCLE (%)
110
100
90
80
70
60
50
40
30
20
10
SF = I
OUT
/I
OUT(MAX)
(%)
1622 F02
0 70 80 90 1006010 20 30 40 50
I
RIPPLE
= 0.4I
PK
AT 5% DUTY CYCLE
I
RIPPLE
= 0.2I
PK
AT 5% DUTY CYCLE
V
IN
= 4.2V
UNSYNC
Figure 2. Maximum Output Current vs Duty Cycle
7
LTC1622
APPLICATIONS INFORMATION
WUUU
Kool Mu is a registered trademark of Magnetics, Inc.
The basic LTC1622 application circuit is shown in Figure
1. External component selection is driven by the load
requirement and begins with the selection of L and R
SENSE
.
Next, the Power MOSFET and the output diode D1 are
selected followed by C
IN
and C
OUT
.
R
SENSE
Selection for Output Current
R
SENSE
is chosen based on the required output current.
With the current comparator monitoring the voltage devel-
oped across R
SENSE
, the threshold of the comparator
determines the inductor’s peak current. The output cur-
rent the LTC1622 can provide is given by:
IR
I
OUT SENSE
RIPPLE
=−
008
2
.
where I
RIPPLE
is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section).
A reasonable starting point for setting ripple current is
I
RIPPLE
= (0.4)(I
OUT
). Rearranging the above equation, it
becomes:
RI
SENSE OUT
=
()( )
1
15 for Duty Cycle < 40%
However, for operation that is above 40% duty cycle, slope
compensation has to be taken into consideration to select
the appropriate value to provide the required amount of
current. Using Figure 2, the value of R
SENSE
is:
RSF
I
SENSE OUT
=
()( )( )
15 100
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple
current. The ripple current, I
RIPPLE
, decreases with higher
inductance or frequency and increases with higher V
IN
or
V
OUT
. The inductor’s peak-to-peak ripple current is given
by:
IVV
fL
VV
VV
RIPPLE IN OUT OUT D
IN D
=−
()
+
+
where f is the operating frequency. Accepting larger values
of I
RIPPLE
allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
I
RIPPLE
= 0.4(I
OUT(MAX)
). Remember, the maximum I
RIPPLE
occurs at the maximum input voltage.
With Burst Mode operation selected on the LTC1622, the
ripple current is normally set such that the inductor
current is continuous during the burst periods. Therefore,
the peak-to-peak ripple current should not exceed:
IR
RIPPLE SENSE
≤0 036.
This implies a minimum inductance of:
LVV
fR
VV
VV
MIN IN OUT
SENSE
OUT D
IN D
=−
+
+
0 036.
(Use V
IN(MAX)
= V
IN
)
A smaller value than L
MIN
could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mu
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase. Ferrite designs have very low core losses and are
8
LTC1622
APPLICATIONS INFORMATION
WUUU
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core materials saturate “hard,” which means that
the inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequently, output voltage
ripple. Do not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mu. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new surface mountable designs that do
not increase the height significantly are available.
Power MOSFET Selection
An external P-channel power MOSFET must be selected
for use with the LTC1622. The main selection criteria for
the power MOSFET are the threshold voltage V
GS(TH)
and
the “on” resistance R
DS(ON)
,reverse transfer capacitance
C
RSS
and total gate charge.
Since the LTC1622 is designed for operation down to low
input voltages, a sublogic level threshold MOSFET (R
DS(ON)
guaranteed at V
GS
= 2.5V) is required for applications that
work close to this voltage. When these MOSFETs are used,
make sure that the input supply to the LTC1622 is less than
the absolute maximum MOSFET V
GS
rating, typically 8V.
The gate drive voltage levels are from ground to V
IN
.
The required minimum R
DS(ON)
of the MOSFET is gov-
erned by its allowable power dissipation. For applications
that may operate the LTC1622 in dropout, i.e., 100% duty
cycle, at its worst case the required R
DS(ON)
is given by:
RP
Ip
DS ON P
OUT MAX
DC
() ()
%= =
()
+
()
100
2
1δ
where P
P
is the allowable power dissipation and δp is the
temperature dependency of R
DS(ON)
. (1 + δp) is generally
given for a MOSFET in the form of a normalized R
DS(ON)
vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less than
100% and the LTC1622 is in continuous mode, the R
DS(ON)
is governed by:
RP
DC I p
DS ON P
OUT
()
≅
()
+
()
2
1δ
where DC is the maximum operating duty cycle of the
LTC1622.
When the LTC1622 is operating in continuous mode, the
MOSFET power dissipation is:
PVV
VV
IpR
KV I C f
MOSFET OUT D
IN D OUT DS ON
IN OUT RSS
=+
+
()
+
()
+
()( )( )()
2
2
1δ()
where K is a constant inversely related to gate drive
current. Because of the high switching frequency, the
second term relating to switching loss is important not to
overlook. The constant K = 3 can be used to estimate the
contributions of the two terms in the MOSFET dissipation
equation.
Output Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As V
IN
approaches V
OUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short circuited. Under this condition the diode must
safely handle I
PEAK
at close to 100% duty cycle. Therefore,
it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.
Under normal load conditions, the average current con-
ducted by the diode is:
IVV
VVI
DIN OUT
IN D OUT
=−+
9
LTC1622
APPLICATIONS INFORMATION
WUUU
∆V I ESR fC
OUT RIPPLE OUT
≈+
1
8
where f is the operating frequency, C
OUT
is the output
capacitance and I
RIPPLE
is the ripple current in the induc-
tor. The output ripple is highest at maximum input voltage
since ∆I
L
increases with input voltage.
The choice of using a smaller output capacitance in-
creases the output ripple voltage due to the frequency
dependent term, but can be compensated for by using
capacitors of very low ESR to maintain low ripple voltage.
The I
TH
pin OPTI-LOOP compensation components can be
optimized to provide stable, high performance transient
response regardless of the output capacitors selected.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for C
OUT
has been
met, the RMS current rating generally far exceeds the
I
RIPPLE(P-P)
requirement.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON, Sanyo
POSCAP, Nichicon PL series and the Panasonic SP series.
Low Supply Operation
Although the LTC1622 can function down to 2V, the
maximum allowable output current is reduced when V
IN
decreases below 3V. Figure 3 shows the amount of change
as the supply is reduced down to 2V. Also shown in
Figure 3 is the effect of V
IN
on V
REF
as V
IN
goes below 2.3V.
Remember the maximum voltage on the I
TH
pin defines
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
VP
I
FD
SC MAX
≈
()
where P
D
is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
C
IN
and C
OUT
Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (V
OUT
+ V
D
)/
(V
IN
+ V
D
). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:
CI
VVV
V
IN MAX OUT IN OUT
IN
Required I
RMS
≈−
()
[]
12/
This formula has a maximum at V
IN
= 2V
OUT
, where I
RMS
= I
OUT
/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1622, ceramic capacitors
can also be used for C
IN
. Always consult the manufacturer
if there is any question.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆V
OUT
) is approximated by:
10
LTC1622
the maximum current sense voltage that sets the maxi-
mum output current.
Setting Output Voltage
The LTC1622 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the output voltage. The regulated
output voltage is determined by:
VR
R
OUT
=+
08 1 2
1
.
For most applications, a 30k resistor is suggested for R1.
To prevent stray pickup, an optional 100pF capacitor is
suggested across R1 located close to LTC1622.
APPLICATIONS INFORMATION
WUUU
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1622 circuits: 1) LTC1622 DC bias current,
2) MOSFET gate charge current, 3) I
2
R losses, 4) voltage
drop of the output diode and 5) transition losses.
1. The V
IN
current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. V
IN
current results in a small loss
which increases with V
IN
.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge dQ moves from V
IN
to ground.
The resulting dQ/dt is a current out of V
IN
which is
typically much larger than the DC supply current. In
continuous mode, I
GATECHG
= f(Qp).
3. I
2
R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but
is “chopped” between the P-channel MOSFET in series
with R
SENSE
and the output diode. The MOSFET R
DS(ON)
plus R
SENSE
multiplied by duty cycle can be summed
with the resistance of the inductor to obtain I
2
R losses.
4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage drop times the diode duty cycle multiplied by
the load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.
5. Transition losses apply to the external MOSFET and
increase with higher operating frequencies and input
voltages. Transition losses can be estimated from:
3
V
FB
V
OUT
LTC1622
100pF R1
1622 F04
R2
Figure 4. Setting Output Voltage
INPUT VOLTAGE (V)
2.0
NORMALIZED VOLTAGE (%)
101
100
99
98
97
96
95 2.2 2.4 2.6 2.8
1622 F03
3.0
V
REF
V
ITH
Figure 3. Line Regulation of VREF and VITH
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
11
LTC1622
APPLICATIONS INFORMATION
WUUU
Transition Loss = 3(V
IN
)
2
I
O(MAX)
C
RSS
(f)
Other losses including C
IN
and C
OUT
ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Run/Soft-Start Function
The RUN/SS pin is a dual purpose pin that provides the
soft-start function and a means to shut down the LTC1622.
Soft-start reduces input surge current from V
IN
by gradu-
ally increasing the internal current limit. Power supply
sequencing can also be accomplished using this pin.
An internal 2.5µA current source charges up an external
capacitor C
SS
. When the voltage on the RUN/SS reaches
0.7V the LTC1622 begins operating. As the voltage on
RUN/SS continues to ramp from 0.7V to 1.8V, the internal
current limit is also ramped at a proportional linear rate.
The current limit begins near 0A (at V
RUN/SS
= 0.7V) and
ends at 0.1/R
SENSE
(V
RUN/SS
≥ 1.8V). The output current
thus ramps up slowly, reducing the starting surge current
required from the input power supply. If the RUN/SS has
been pulled all the way to ground, there will be a delay
before the current limit starts increasing and is given by:
t
DELAY
= 2.8 • 10
5
• C
SS
in seconds
Pulling the RUN/SS pin below 0.4V puts the LTC1622 into
a low quiescent current shutdown (I
Q
< 15µA).
Foldback Current Limiting
As described in the Output Diode Selection, the worst-
case dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.
Foldback current limiting is implemented by adding diode
D
FB
(1N4148 or equivalent) between the output and the I
TH
pin as shown in Figure 5. In a hard short (V
OUT
= 0V), the
current will be reduced to approximately 50% of the
maximum output current.
VFB
ITH
VOUT
LTC1622
R1
1622 F05
R2
DFB
+
Figure 5. Foldback Current Limiting
Design Example
Assume the LTC1622 is used in a single lithium-ion
battery-powered cellular phone application. The V
IN
will be
operating from a maximum of 4.2V down to a minimum of
2.7V. Load current requirement is a maximum of 1.5A but
most of the time it will be on standby mode, requiring only
2mA. Efficiency at both low and high load current is
important. Output voltage is 2.5V.
In the above application, Burst Mode operation is enabled
by connecting Pin 5 to V
IN
.
Maximum VV
VV
OUT D
IN MIN D
Duty Cycle =++=
()
%93
From Figure 2, SF = 57%.
Use the curve of Figure 2 since the operating frequency is
the free running frequency of the LTC1622.
RSF
IA
SENSE OUT
=
()( )( )
=
()( )
=
15 100
057
15 1 5 0 0253
.
..Ω
In the application, a 0.025Ω resistor is used. For the
inductor, the required value is:
L
kHz
H
MIN
=−
+
+
=
42 25
550 0 036
0 025
25 03
42 03 133
..
.
.
..
.. .µ
In the application, a 3.9µH inductor is used to reduce
inductor ripple current and thus, output voltage ripple.
For the selection of the external MOSFET, the R
DS(ON)
must be guaranteed at 2.5V since the LTC1622 has to work
12
LTC1622
APPLICATIONS INFORMATION
WUUU
down to 2.7V. Let’s assume that the MOSFET dissipation
is to be limited to P
P
= 250mW and its thermal resistance
is 50°C/W. Hence the junction temperature at T
A
= 25°C
will be 37.5°C and δp = 0.005 (37.5 – 25) = 0.0625. The
required R
DS(ON)
is then given by:
RP
DC I p
DS ON P
OUT
()
.≅
()
+
()
=
2
1
011
δΩ
The P-channel MOSFET requirement can be met by an
Si6433DQ.
The requirement for the Schottky diode is the most strin-
gent when V
OUT
= 0V, i.e., short circuit. With a 0.025Ω
R
SENSE
resistor, the short-circuit current through the
Schottky is 0.1/0.025 = 4A. An MBRS340T3 Schottky
diode is chosen. With 4A flowing through, the diode is
rated with a forward voltage of 0.4V. Therefore, the worst-
case power dissipated by the diode is 1.6W. The addition
of D
FB
(Figure 5) will reduce the diode dissipation to
approximately 0.8W.
The input capacitor requires an RMS current rating of at
least 0.75A at temperature, and C
OUT
will require an ESR
of 0.1Ω for optimum efficiency.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1622. These items are illustrated graphically in the
layout diagram in Figure 6. Check the following in your
layout:
1. Is the Schottky diode closely connected between ground
at (–) lead of C
IN
and drain of the external MOSFET?
2. Does the (+) plate of C
IN
connect to the sense resistor
as closely as possible? This capacitor provides AC
current to the MOSFET.
3. Is the input decoupling capacitor (0.1µF) connected
closely between V
IN
(Pin 8) and ground (Pin 6)?
4. Connect the end of R
SENSE
as close to V
IN
(Pin 8) as
possible. The V
IN
pin is the SENSE
+
of the current
comparator.
5. Is the trace from the SENSE
–
(Pin 1) to the Sense
resistor kept short? Does the trace connect close to
R
SENSE
?
6. Keep the switching node, SW, away from sensitive
small signal nodes.
7. Does the V
FB
pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of C
OUT
and signal
ground. Optional capacitor C1 should be located as
close as possible to the LTC1622.
R1 and R2 should be located as close as possible to the
LTC1622. R2 should connect to the output as close to
the load as practicable.
Figure 6. LTC1622 Layout Diagram (See PC Board Layout Checklist)
1
2
3
4
8
7
6
5
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
SYNC/
MODE
RUN/
SS
L1
R1 R2
BOLD LINES INDICATE HIGH CURRENT PATHS
R
SENSE
1622 F06
0.1µFM1
SW
C
ITH
R
ITH
C
SS
QUIET SGND
C1
LTC1622
C
IN
+
C
OUT
V
OUT
V
IN
+
13
LTC1622
TYPICAL APPLICATIONS
U
1
2
3
4
8
7
6
5
1
4
8
7
6
5
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
SYNC/
MODE
RUN/
SS
L1
3.3µH
R2
0.025ΩU1
V
OUT
1.8V
1.5A
V
IN
2.5V TO
8.5V
1622 TA01
R1
10K
C3
220pF
C1
47µF
16V
C2
220µF
6V
C4
560pF
R3
93.1k
LTC1622
+
+
R4
75k
C1: AVX TPSD476M016R0150
C2: AVX TPSD227M006R0100
L1: MURATA LQN6C3R3
R2: DALE WSL-1206 0.025Ω
U1: INTERNATIONAL RECTIFIER
FETKY
TM
IRF7422D2
2
3
LTC1622 1.8V/1.5A Regulator with Burst Mode Operation Disabled
FETKY
is a trademark of International Rectifier Corporation.
LTC1622 2.5V/2A Regulator with Burst Mode Operation Enabled
D1 L1
4.7µH
R2
0.02Ω
M1
R3
158k
V
OUT
2.5V
2A
V
IN
3.3V TO
8.5V
R4
75k
1622 TA02
C4
560pF
C1
47µF
16V
× 2
+
C2
150µF
6V
× 2
+
1
2
3
4
8
7
6
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
RUN/
SS
LTC1622
SYNC/
MODE
R1
10k
C3
220pF
5
C1: AVX TPSD476M016R0150
C2: SANYO POSCAP 6TPA47M
D1: MOTOROLA MBR320T3
L1: COILCRAFT D03316-472
M1: SILICONIX Si3443DV
R2: DALE WSL-2010 0.02Ω
14
LTC1622
TYPICAL APPLICATIONS
U
LTC1622 2.5V/3A Regulator with External Frequency Synchronization
D1 L1
4.7µH
R2
0.01Ω
M1
R3
158k
V
IN
3.3V TO
8.5V
V
OUT
2.5V
3A
R4
75k
1622 TA03
C4
560pF
650kHz
1.5V
P-P
C1
47µF
16V
× 2
+
C2
100µF
6.3V
× 2
+
1
2
3
4
8
7
6
5
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
RUN/
SS
LTC1622
SYNC/
MODE
R1
10k
C3
220pF
C1: AVX TPSD476M016R0150
C2: AVX TPSD107M010R0065
D1: MOTOROLA MBR320T3
L1: COILCRAFT D03316-472
M1: SILICONIX Si3443DV
R2: DALE WSL-2512 0.01Ω
Zeta Converter with Foldback Current Limit
R2
0.04Ω
Si3441DV
L1A
6.2µH
L1B
6.2µH
D1
47µF
16V
R3
232k
VOUT
3.3V
VIN
2.5V TO
8.5V
R4
75k
1622 TA04
C4
0.1µF
R1
47k
C3
470pF
C1: AVX TPSD476M016R0150
C2: AVX TPSD107M010R0080
D1: MOTOROLA MBRS320T3
L1A, L1B: BH ELECTRONICS BH511-1012
R2: DALE WSL-1206 0.04Ω
C1
47µF
16V
× 2
+
C2
100µF
10V
+
1
2
3
4
8
7
6
5
SENSE–
ITH
VFB
VIN
PDRV
GND
RUN/
SS
LTC1622
SYNC/
MODE
+
TOP VIEW
1
•
4
32
L1A
L1B
VIN IOUT(MAX)
(V) (A)
2.5 0.45
3.3 0.70
5.0 0.95
6.0 1.00
8.4 1.05
D2
1N4818
15
LTC1622
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTION
U
MS8 Package
8-Lead Plastic MSOP
(LTC DWG # 05-08-1660)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
MSOP (MS8) 1098
* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
0.021 ± 0.006
(0.53 ± 0.015)
0° – 6° TYP
SEATING
PLANE
0.007
(0.18)
0.040 ± 0.006
(1.02 ± 0.15)
0.012
(0.30)
REF
0.006 ± 0.004
(0.15 ± 0.102)
0.034 ± 0.004
(0.86 ± 0.102)
0.0256
(0.65)
BSC
12
34
0.193 ± 0.006
(4.90 ± 0.15)
8765
0.118 ± 0.004*
(3.00 ± 0.102)
0.118 ± 0.004**
(3.00 ± 0.102)
0.016 – 0.050
(0.406 – 1.270)
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 1298
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
16
LTC1622
sn1622 1622fs LT/TP 1001 4K • PRINTED IN USA
LINEAR T ECHNOLOGY CORPORATION 1998
TYPICAL APPLICATION
U
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507
●
www.linear-tech.com
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Small Footprint 3.3V/1A Regulator
1
2
3
4
8
7
6
5
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
SYNC/
MODE D1
RUN/
SS
L1
2.2µH
R2
0.025Ω
M1
R3
232k
V
OUT
3.3V
1A
R1
10k
C3
220pF
C1: MURATA CERAMIC GRM235Y5V106Z
C2: SANYO POSCAP 6TPA47M
D1: MOTOROLA MBRS120LT3
V
IN
3.3V TO
8.5V
R4
75k
1622 TA05
C4
560pF
C1
10µF
16V
CERAMIC
LTC1622
+
C2
47µF
6V
+
L1: COILCRAFT D01608C-222
M1: SILICONIX Si3443DY
R2: DALE WSL-2010 0.025Ω
Efficiency vs Load Current
LOAD CURRENT (mA)
1
EFFICIENCY (%)
100
90
80
70
60
50 10 100 1000
1622 TA05b
V
OUT
= 3.3V
R
SENSE
= 0.025Ω
V
IN
= 3.5V
V
IN
= 4.2V
V
IN
= 6V
Boost Converter 3.3V/2.5A
Efficiency vs Load Current With LTC1622
Configured as Boost Converter
D1
L1
4.6µHR3
105k
R2
0.015Ω
V
IN
3.3V
V
OUT
5V
2.5A
R4
20k
1622 TA06a
C4
0.1µF
C5
150pF
C1
100µF
10V
Si6801DQ
M1
+
C6
0.1µF
1
2
3
4
8
7
6
5
SENSE
–
I
TH
V
FB
V
IN
PDRV
GND
RUN/
SS
LTC1622
SYNC/
MODE
R1
33k
C3
470pF
C1, C2: SANYO POSCAP TPB SERIES
D1: MOTOROLA MBRD835L
L1: SUMIDA CEP123-4R6
C2
220µF
10V
×2
+
M1: SILICONIX Si3442DV
R2: DALE WS-L2512 0.015ΩLOAD CURRENT (mA)
0.001
EFFICIENCY (%)
100
90
80
70
60
50 0.01 0.1 1
1622 TA06b
V
OUT
= 5V
R
SENSE
= 0.015Ω
V
IN
= 3.3V
