AD7641BSTZRL - Analog Devices - Datasheet.Directory

18-Bit, 2 MSPS SAR ADC

AD7641

Rev. 0

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FEATURES

Throughput

2 MSPS (wideband warp and warp mode)

1.5 MSPS (normal mode)

INL: ±2 LSB typical, ±3 LSB max; ±8 ppm of full scale

18-bit resolution with no missing codes

Dynamic range: 95.5 dB

SNR: 93.5 dB typical @ 20 kHz (VREF = 2.5 V)

THD: −112 dB typical @ 20 kHz (VREF = 2.5 V)

2.048 V internal reference: typ drift 10 ppm/°C; TEMP output

Differential input range: ±VREF (VREF up to 2.5 V)

No pipeline delay (SAR architecture)

Parallel (18-, 16-, or 8-bit bus) and

serial 5 V/3.3 V/2.5 V interface

SPI®/QSPI™/MICROWIRE™/DSP compatible

Single 2.5 V supply operation

Power dissipation

75 mW typical @ 2 MSPS with internal REF

2 μW in power-down mode

Pb-free, 48-lead LQFP and 48-lead LFCSP_VQ

Speed upgrade of the AD7674, AD7678, AD7679

APPLICATIONS

Medical instruments

High speed data acquisition/high dynamic data acquisition

Digital signal processing

Spectrum analysis

Instrumentation

Communications

ATE

GENERAL DESCRIPTION

The AD7641 is an 18-bit, 2 MSPS, charge redistribution SAR,

fully differential, analog-to-digital converter (ADC) that

operates from a single 2.5 V power supply. The part contains a

high speed, 18-bit sampling ADC, an internal conversion clock,

an internal reference (and buffer), error correction circuits, and

both serial and parallel system interface ports. It features two

very high sampling rate modes (wideband warp and warp) and

a fast mode (normal) for asynchronous rate applications. The

AD7641 is hardware factory calibrated and tested to ensure ac

parameters, such as signal-to-noise ratio (SNR), in addition to

the more traditional dc parameters of gain, offset, and linearity.

The AD7641 is available in Pb-free only packages with

operation specified from −40°C to +85°C.

FUNCTIONAL BLOCK DIAGRAM

04761-001

CONTROL LOGIC AND

CALIBRATION CIRCUITRY

CLOCK

AD7641

DGNDD

AVDD

AGND

REF REFGND

IN+

IN–

RESET

CNVST

PDBUF

REFBUFIN

PDREF

REF

TEMP

D[17:0]

BUSY

D0/OB/2C

OGND

OVDD

MODE1

MODE0

REF AMP

NORMAL

SERIAL

PORT

PARALLEL

INTERFACE

SWITCHED

CAP DAC

WARP

Figure 1.

Table 1. PulSAR® Selection

Type/kSPS

100 to

250

500 to

570

650 to

1000 >1000

Pseudo

Differential

AD7651,

AD7660,

AD7661

AD7650,

AD7652,

AD7664,

AD7666

AD7653,

AD7667

True Bipolar AD7610,

AD7663

AD7665 AD7612,

AD7671

True

Differential

AD7675 AD7676 AD7677 AD7621,

AD7622,

AD7623

18-Bit

Multichannel/

AD7631,

AD7678

AD7679 AD7634,

AD7674

AD7641,

AD7643

Simultaneous AD7654 AD7655

PRODUCT HIGHLIGHTS

1. Fast Throughput.

The AD7641 is a 2 MSPS, charge redistribution,

18-bit SAR ADC.

2. Superior Linearity.

The AD7641 has no missing 18-bit code.

3. Internal Reference.

The AD7641 has a 2.048 V internal reference with a typical

-chip TEMP sensor.

drift of ±10 ppm/°C and an on

Single-Supply Operation. 4.

2.5 V single supply.

5. ace

arrangement compatible with 2.5 V, 3.3 V, or 5 V logic.

The AD7641 operates from a

Serial or Parallel Interface.

Versatile parallel (16- or 8-bit bus) or 2-wire serial interf

AD7641

Rev. 0 | Page 2 of 28

TABLE OF CONTENTS

Features .............................................................................................. 1

Applications....................................................................................... 1

General Description......................................................................... 1

Functional Block Diagram .............................................................. 1

Product Highlights ........................................................................... 1

Revision History ............................................................................... 2

Specifications..................................................................................... 3

Timing Specifications....................................................................... 5

Absolute Maximum Ratings............................................................ 7

ESD Caution.................................................................................. 7

Pin Configuration and Function Descriptions............................. 8

Terminology .................................................................................... 11

Typical Performance Characteristics ........................................... 12

Appplications Information............................................................ 15

Circuit Information....................................................................15

Converter Operation.................................................................. 15

Modes of Operation ................................................................... 15

Transfer Functions...................................................................... 16

Typical Connection Diagram........................................................ 17

Analog Inputs ............................................................................. 17

Multiplexed Inputs..................................................................... 17

Driver Amplifier Choice ........................................................... 17

Voltage Reference Input ............................................................ 18

Power Supply............................................................................... 20

Conversion Control ................................................................... 20

Interfaces.......................................................................................... 21

Digital Interface.......................................................................... 21

Parallel Interface......................................................................... 21

Serial interface ............................................................................ 22

Master Serial Interface............................................................... 22

Slave Serial Interface.................................................................. 24

Microprocessor Interfacing....................................................... 26

Application Hints ........................................................................... 27

Layout .......................................................................................... 27

Evaluating the AD7641 Performance...................................... 27

Outline Dimensions....................................................................... 28

Ordering Guide .......................................................................... 28

REVISION HISTORY

1/06—Revision 0: Initial Version

AD7641

Rev. 0 | Page 3 of 28

SPECIFICATIONS

AVDD = DVDD = 2.5 V; OVDD = 2.3 V to 3.6 V; VREF = 2.5 V; all specifications TMIN to TMAX, unless otherwise noted.

Table 2.

Parameter Conditions Min Typ Max Unit

RESOLUTION 18 Bits

ANALOG INPUT

Voltage Range VIN+ − VIN− −VREF +VREF V

Operating Input Voltage VIN+, VIN− to AGND −0.1 AVDD1V

Analog Input CMRR fIN = 100 kHz 58 dB

Input Current 2 MSPS throughput 18 μA

Input Impedance2

THROUGHPUT SPEED

Complete Cycle Wideband warp, warp modes 500 ns

Throughput Rate Wideband warp, warp modes 0.001 2 MSPS

Time Between Conversions Wideband warp, warp modes 1 ms

Complete Cycle Normal mode 667 ns

Throughput Rate Normal mode 0 1.5 MSPS

DC ACCURACY

Integral Linearity Error3TMIN to TMAX = −40°C to +70°C −3 ±2 +3 LSB4

Integral Linearity Error TMIN to TMAX = −40°C to +85°C −3.5 ±2 +3.5 LSB4

No Missing Codes 18 Bits

Differential Linearity Error −1 +2 LSB

Transition Noise VREF = 2.5 V 1.6 LSB

Transition Noise VREF = 2.048 V 2.0 LSB

Zero Error, TMIN to TMAX5 −15 +15 LSB

Zero Error Temperature Drift ±0.5 ppm/°C

Gain Error, TMIN to TMAX5 −0.25 +0.25 % of FSR

Gain Error Temperature Drift ±1 ppm/°C

Power Supply Sensitivity AVDD = 2.5 V ± 5% ±16 LSB

AC ACCURACY

Dynamic Range VREF = 2.5 V 95.5 dB6

Signal-to-Noise fIN = 20 kHz, VREF = 2.5 V 93.5 dB

IN = 20 kHz, VREF = 2.048 V 92 dB

IN = 100 kHz, VREF = 2.5 V 93 dB

Spurious-Free Dynamic Range fIN = 20 kHz, VREF = 2.5 V 112 dB

IN = 20 kHz, VREF = 2.048 V 113 dB

IN = 100 kHz, VREF = 2.5 V 101 dB

Total Harmonic Distortion fIN = 20 kHz, VREF = 2.5 V −115 dB

IN = 20 kHz, VREF = 2.048 V −116 dB

IN = 100 kHz, VREF = 2.5 V −101 dB

Signal-to-(Noise + Distortion) fIN = 20 kHz, , VREF = 2.5 V 93.5 dB

IN = 20 kHz, VREF = 2.048 V 92 dB

IN = 100 kHz, , VREF = 2.5 V 92.5 dB

−3 dB Input Bandwidth 50 MHz

SAMPLING DYNAMICS

Aperture Delay 1 ns

Aperture Jitter 5 ps rms

Transient Response Full-scale step 115 ns

INTERNAL REFERENCE PDREF = PDBUF = low

Output Voltage REF @ 25°C 2.038 2.048 2.058 V

Temperature Drift −40°C to +85°C ±10 ppm/°C

Line Regulation AVDD = 2.5 V ± 5% ±15 ppm/V

AD7641

Rev. 0 | Page 4 of 28

Parameter Conditions Min Typ Max Unit

Turn-On Settling Time CREF = 10 μF 5 ms

REFBUFIN Output Voltage REFBUFIN @ 25°C 1.19 V

REFBUFIN Output Resistance 6.33 kΩ

EXTERNAL REFERENCE PDREF = PDBUF = high

Voltage Range REF 1.8 2.048 AVDD + 0.1 V

Current Drain 2 MSPS throughput 180 μA

REFERENCE BUFFER PDREF = high, PDBUF = low

REFBUFIN Input Voltage Range REF = 2.048 V typ 1.05 1.2 1.30 V

REFBUFIN Input Current REFBUFIN = 1.2 V 1 nA

TEMPERATURE PIN

Voltage Output @ 25°C 278 mV

Temperature Sensitivity 1 mV/°C

Output Resistance 4.7 kΩ

DIGITAL INPUTS

Logic Levels

VIL −0.3 +0.6 V

VIH 1.7 5.25 V

IIL −1 +1 μA

IIH −1 +1 μA

DIGITAL OUTPUTS

Data Format7

Pipeline Delay8

VOL I

SINK = 500 μA 0.4 V

VOH I

SOURCE = −500 μA OVDD − 0.3 V

POWER SUPPLIES

Specified Performance

AVDD 2.37 2.5 2.63 V

DVDD 2.37 2.5 2.63 V

OVDD 2.309 3.6 V

Operating Current10 2 MSPS throughput

AVDD11 With internal reference 23 mA

DVDD 2.5 mA

OVDD12 0.5 mA

Power Dissipation11

With Internal Reference10 2 MSPS throughput 75 92 mW

Without Internal Reference10 2 MSPS throughput 68 85 mW

In Power-Down Mode12 PD = high 2 μW

TEMPERATURE RANGE13

Specified Performance TMIN to TMAX −40 +85 °C

1 When using an external reference. With the internal reference, the input range is −0.1 V to VREF.

2 See Analog Inputs section.

3 Linearity is tested using endnotes, not best fit.

4 LSB means least significant bit. With the ±2.048 V input range, 1 LSB is 15.63 μV.

5 See Voltage Reference Input section. These specifications do not include the error contribution from the external reference.

6 All specifications in dB are referred to a full-scale input FS. Tested with an input signal at 0.5 dB below full-scale, unless otherwise specified.

7 Parallel or serial 18-bit.

8 Conversion results are available immediately after completed conversion.

9 See the Absolute Maximum Ratings section.

10 In warp mode. Tested in parallel reading mode.

11 With internal reference, PDREF and PDBUF are low; without internal reference, PDREF and PDBUF are high.

12 With all digital inputs forced to OVDD.

13Consult sales for extended temperature range.

AD7641

Rev. 0 | Page 5 of 28

TIMING SPECIFICATIONS

AVDD = DVDD = 2.5 V; OVDD = 2.3 V to 3.6 V; VREF = 2.5 V; all specifications TMIN to TMAX, unless otherwise noted.

Table 3.

Parameter Symbol Min Typ Max Unit

CONVERSION AND RESET (Refer to Figure 29 and Figure 30)

Convert Pulse Width t1 15 701ns

Time Between Conversions (Warp Mode2/Normal Mode3) t2 500/667 ns

CNVST Low to BUSY High Delay t3 23 ns

BUSY High All Modes (Except Master Serial Read After Convert) t4 385/520 ns

Aperture Delay t5 1 ns

End of Conversion to BUSY Low Delay t6 10 ns

Conversion Time (Warp Mode/Normal Mode) t7 385/520 ns

Acquisition Time (Warp Mode/Normal Mode) t8 115 ns

RESET Pulse Width t9 15 ns

RESET Low to BUSY High Delay4t38 10 ns

BUSY High Time from RESET Low4t39 600 ns

PARALLEL INTERFACE MODES (Refer to Figure 31 to Figure 34 )

CNVST Low to Data Valid Delay (Warp Mode/Normal Mode) t10 385/520 ns

Data Valid to BUSY Low Delay t11 2 ns

Bus Access Request to Data Valid t12 20 ns

Bus Relinquish Time t13 2 15 ns

MASTER SERIAL INTERFACE MODES5 (Refer to Figure 35 and Figure 36)

CS Low to SYNC Valid Delay t14 10 ns

CS Low to Internal SCLK Valid Delay5t15 10 ns

CS Low to SDOUT Delay t16 10 ns

CNVST Low to SYNC Delay (Warp Mode/Normal Mode) t17 14/137 ns

SYNC Asserted to SCLK First Edge Delay t18 0.5 ns

Internal SCLK Period6t19 8 14 ns

Internal SCLK High6t20 2 ns

Internal SCLK Low6t21 3 ns

SDOUT Valid Setup Time6t22 1 ns

SDOUT Valid Hold Time6t23 0 ns

SCLK Last Edge to SYNC Delay6t24 0 ns

CS High to SYNC HI-Z t25 10 ns

CS High to Internal SCLK HI-Z t26 10 ns

CS High to SDOUT HI-Z t27 10 ns

BUSY High in Master Serial Read After Convert6t28 See Table 4 ns

CNVST Low to SYNC Asserted Delay (All Modes) t29 383/500 ns

SYNC Deasserted to BUSY Low Delay t30 13 ns

SLAVE SERIAL INTERFACE MODES (Refer to Figure 38 and Figure 39)

External SCLK Setup Time t31 5 ns

External SCLK Active Edge to SDOUT Delay t32 1 8 ns

SDIN Setup Time t33 5 ns

SDIN Hold Time t34 5 ns

External SCLK Period t35 12.5 ns

External SCLK High t36 5 ns

External SCLK Low t37 5 ns

1 See the Conversion Control section.

2 All timings for wideband warp mode are the same as warp mode.

3 In warp mode only, the maximum time between conversions is 1 ms; otherwise, there is no required maximum time.

4 See the Digital Interface section and the RESET section.

5 In serial interface modes, the SYNC, SCLK, and SDOUT timings are defined with a maximum load CL of 10 pF; otherwise, the load is 60 pF maximum.

6 In serial master read during convert mode. See Table 4 for serial master read after convert mode timing specifications.

AD7641

Rev. 0 | Page 6 of 28

Table 4. Serial Clock Timings in Master Read After Convert Mode

DIVSCLK[1] 0 0 1 1

DIVSCLK[0] Symbol 0 1 0 1 Unit

SYNC to SCLK First Edge Delay Minimum t18 0.5 3 3 3 ns

Internal SCLK Period Minimum t19 8 16 32 64 ns

Internal SCLK Period Maximum t19 14 26 52 103 ns

Internal SCLK High Minimum t20 2 6 15 31 ns

Internal SCLK Low Minimum t21 3 7 16 32 ns

SDOUT Valid Setup Time Minimum t22 1 5 5 5 ns

SDOUT Valid Hold Time Minimum t23 0 0.5 10 28 ns

SCLK Last Edge to SYNC Delay Minimum t24 0 0.5 9 26 ns

BUSY High Width Maximum t24 0.630 0.870 1.350 2.28 μs

04761-002

NOTE

IN SERIAL INTERFACE MODES, THE SYNC, SCLK, AND

SDOUT TIMING ARE DEFINED WITH A MAXIMUM LOAD

OF 10pF; OTHERWISE, THE LOAD IS 60pF MAXIMUM.

500µAI

1.4V

TO OUTPUT

PIN C

50pF

Figure 2. Load Circuit for Digital Interface Timing,

SDOUT, SYNC, and SCLK Outputs, CL = 10 pF

0.8V

DELAY

04761-003

Figure 3. Voltage Reference Levels for Timing

AD7641

Rev. 0 | Page 7 of 28

ABSOLUTE MAXIMUM RATINGS

Table 5.

Parameter Rating

Analog Inputs/Outputs

IN+1, IN−, REF, REFBUFIN, TEMP,

INGND, REFGND to AGND

AVDD + 0.3 V to

AGND − 0.3 V

Ground Voltage Differences

AGND, DGND, OGND ±0.3 V

Supply Voltages

AVDD, DVDD −0.3 V to +2.7 V

OVDD −0.3 V to +3.8 V

AVDD to DVDD ±2.8 V

AVDD, DVDD to OVDD −3.8 V to +2.8 V

Digital Inputs −0.3 V to +5.5 V

PDREF, PDBUF2±20 mA

Internal Power Dissipation3700 mW

Internal Power Dissipation42.5 W

Junction Temperature 125°C

Storage Temperature Range –65°C to +125°C

Stresses above those listed under Absolute Maximum Ratings

may cause permanent damage to the device. This is a stress

rating only; functional operation of the device at these or any

other conditions above those indicated in the operational

section of this specification is not implied. Exposure to absolute

maximum rating conditions for extended periods may affect

device reliability.

1 See Analog Inputs section.

2 See Voltage Reference Input section.

3 Specification is for the device in free air:

48-Lead LQFP; θJA = 91°C/W, θJC = 30°C/W.

4 Specification is for the device in free air:

48-Lead LFCSP; θJA = 26°C/W.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on

the human body and test equipment and can discharge without detection. Although this product features

proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy

electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance

degradation or loss of functionality.

AD7641

Rev. 0 | Page 8 of 28

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

13 14 15 16 17 18 19 20 21 22 23 24

48 47 46 45 44 39 38 3743 42 41 40

PIN 1

IDENTIFIER

TOP VIEW

(Not to Scale)

AGND

CNVST

RESET

DGND

AGND

AVDD

MODE0

MODE1

D0/OB/2C

NC = NO CONNECT

D1/A0

D2/A1

D4/DIVSCLK[0]

BUSY

D17

D16

D15

AD7641

D5/DIVSCLK[1] D14

D6/EXT/INT

D7/INVSYNC

D8/INVSCLK

D9/RDC/SDIN

OGND

OVDD

DVDD

DGND

D10/SDOUT

D11/SCLK

D12/SYNC

D13/RDERROR

PDBUF

PDREF

REFBUFIN

TEMP

AVDD

IN+

AGND

IN–

REFGND

REF

WARP

04761-004

NORMAL

Figure 4. Pin Configuration

Table 6. Pin Function Descriptions

No. Mnemonic Type1Description

1, 36,

41, 42

AGND P Analog Power Ground Pin.

2, 44 AVDD P Input Analog Power Pins. Nominally 2.5 V.

3, 4 MODE[0:1] DI Data Output Interface Mode Selection.

Interface MODE# MODE1 MODE0 Description

0 0 0 18-bit interface

1 0 1 16-bit interface

2 1 0 8-bit (byte) interface

3 1 1 Serial interface

5 D0/OB/2C DI/O When MODE[1:0] = 0 (18-bit interface mode), this pin is Bit 0 of the parallel port data output bus

and the data coding is straight binary. In all other modes, this pin allows the choice of straight

binary/twos complement. When OB/2C is high, the digital output is straight binary; when low,

the MSB is inverted resulting in a twos complement output from its internal shift register.

6 WARP DI

Conversion Mode Selection. When WARP = high and NORMAL = high, this selects wideband warp

mode with slightly improved linearity and THD. When WARP = high and NORMAL = low, this selects

warp mode. In either mode, these are the fastest modes; maximum throughput is achievable, and

a minimum conversion rate must be applied to guarantee full specified accuracy.

7 NORMAL DI Conversion Mode Selection. When NORMAL = low and WARP = low, this input selects normal mode

where full accuracy is maintained independent of the minimum conversion rate.

8 D1/A0 DI/O

When MODE[1:0] = 0, this pin is Bit 1 of the parallel port data output bus. In all other modes, this

input pin controls the form in which data is output as shown in Table 7.

9 D2/A1 DI/O When MODE[1:0] = 0, this pin is Bit 2 of the parallel port data output bus.

When MODE[1:0] = 1 or 2, this input pin controls the form in which data is output as shown in Table 7.

10 D3 D0 When MODE[1:0] = 0, 1, or 2, this output is used as Bit 3 of the parallel port data output bus.

This pin is always an output, regardless of the interface mode.

11, 12 D[4:5] DI/O When MODE[1:0] = 0, 1, or 2, these pins are Bit 4 and Bit 5 of the parallel port data output bus.

or DIVSCLK[0:1] When MODE[1:0] = 3 (serial mode), serial clock division selection. When using serial master read

after convert mode (EXT/INT = low, RDC/SDIN = low), these inputs can be used to slow down the

internally generated serial clock that clocks the data output. In other serial modes, these pins are

high impedance outputs.

AD7641

Rev. 0 | Page 9 of 28

No. Mnemonic Type1Description

13 D6 DI/O When MODE[1:0] = 0, 1, or 2, this output is used as Bit 6 of the parallel port data output bus.

or EXT/INT When MODE[1:0] = 3, (serial mode), serial clock source select. This input is used to select the

internally generated (master) or external (slave) serial data clock.

When EXT/INT = low, master mode. The internal serial clock is selected on SCLK output.

When EXT/INT = high, slave mode. The output data is synchronized to an external clock signal,

gated by CS, connected to the SCLK input.

14 D7 DI/O When MODE[1:0] = 0, 1, or 2, this output is used as Bit 7 of the parallel port data output bus.

or INVSYNC When MODE[1:0] = 3, (serial mode), invert sync select. In serial master mode (EXT/INT = low), this

input is used to select the active state of the SYNC signal.

When INVSYNC = low, SYNC is active high.

When INVSYNC = high, SYNC is active low.

15 D8 DI/O When MODE[1:0] = 0, 1, or 2, this output is used as Bit 8 of the parallel port data output bus.

or INVSCLK When MODE[1:0] = 3, (serial mode), invert SCLK select. In all serial modes, this input is used to

invert the SCLK signal.

16 D9 DI/O When MODE[1:0] = 0, 1, or 2, this output is used as bit 9 of the parallel port data output bus.

or RDC When MODE[1:0] = 3, (serial mode), read during convert. When using serial master mode

(EXT/INT = low), RDC is used to select the read mode.

When RDC = high, the previous conversion result is output on SDOUT during conversion and

the period of SCLK changes (see the Master Serial Interface section).

When RDC = low (read after convert), the current result can be output on SDOUT only when

the conversion is complete.

or SDIN When MODE[1:0] = 3, (serial mode), serial data in. When using serial slave mode, (EXT/INT = high),

SDIN could be used as a data input to daisy-chain the conversion results from two or more ADCs

onto a single SDOUT line. The digital data level on SDIN is output on SDOUT with a delay of 18 SCLK

periods after the initiation of the read sequence.

17 OGND P Input/Output Interface Digital Power Ground.

18 OVDD P Input/Output Interface Digital Power. Nominally at the same supply as the supply of the

host interface (2.5 V or 3 V).

19 DVDD P Digital Power. Nominally at 2.5 V.

20 DGND P Digital Power Ground.

21 D10 DO When MODE[1:0] = 0, 1, or 2, this output is used as Bit 10 of the parallel port data output bus.

or SDOUT When MODE[1:0] = 3, (serial mode), serial data output. In serial mode, this pin is used as the serial

data output synchronized to SCLK. Conversion results are stored in an on-chip register. The AD7641

provides the conversion result, MSB first, from its internal shift register. The data format is

determined by the logic level of OB/2C.

In master mode, EXT/INT = low. SDOUT is valid on both edges of SCLK.

In slave mode, EXT/INT = high:

When INVSCLK = low, SDOUT is updated on SCLK rising edge and valid on the next falling edge.

When INVSCLK = high, SDOUT is updated on SCLK falling edge and valid on the next rising edge.

22 D11 DI/O When MODE[1:0] = 0, 1, or 2, this output is used as Bit 11 of the parallel port data output bus.

or SCLK When MODE[1:0] = 3, (serial mode), serial clock. In all serial modes, this pin is used as the serial

data clock input or output, depending upon the logic state of the EXT/INT pin. The active edge

where the data SDOUT is updated depends on the logic state of the INVSCLK pin.

23 D12 DO When MODE[1:0] = 0, 1, or 2, this output is used as Bit 12 of the parallel port data output bus.

or SYNC When MODE[1:0] = 3, (serial mode), frame synchronization. In serial master mode (EXT/INT= low),

this output is used as a digital output frame synchronization for use with the internal data clock.

When a read sequence is initiated and INVSYNC = low, SYNC is driven high and remains high

while SDOUT output is valid.

When a read sequence is initiated and INVSYNC = high, SYNC is driven low and remains low

while SDOUT output is valid.

24 D13 DO When MODE[1:0] = 0, 1, or 2, this output is used as Bit 13 of the parallel port data output bus.

or RDERROR When MODE[1:0] = 3, (serial mode), read error. In serial slave mode (EXT/INT = high), this output

is used as an incomplete read error flag. If a data read is started and not completed when the

current conversion is complete, the current data is lost and RDERROR is pulsed high.

AD7641

Rev. 0 | Page 10 of 28

No. Mnemonic Type1Description

25 to

D[14:17] DO

Bit 14 to Bit 17 of the parallel port data output bus. These pins are always outputs, regardless of

the interface mode.

29 BUSY DO

Busy Output. Transitions high when a conversion is started and remains high until the conversion

is complete and the data is latched into the on-chip shift register. The falling edge of BUSY can be

used as a data-ready clock signal.

30 DGND P Digital Power Ground.

31 RD DI Read Data. When CS and RD are both low, the interface parallel or serial output bus is enabled.

32 CS DI Chip Select. When CS and RD are both low, the interface parallel or serial output bus is enabled.

CS is also used to gate the external clock in slave serial mode.

33 RESET DI Reset Input. When high, resets the AD7641. Current conversion, if any, is aborted. Falling edge of

RESET enables the calibration mode indicated by pulsing BUSY high. Refer to the Digital Interface

section. If not used, this pin can be tied to DGND.

34 PD DI Power-Down Input. When high, power downs the ADC. Power consumption is reduced and

conversions are inhibited after the current one is completed.

35 CNVST DI Conversion Start. A falling edge on CNVST puts the internal sample-and-hold into the hold state

and initiates a conversion.

37 REF AI/O Reference Output/Input.

When PDREF/PDBUF = low, the internal reference and buffer are enabled producing 2.048 V on this pin.

When PDREF/PDBUF = high, the internal reference and buffer are disabled allowing an externally

supplied voltage reference up to AVDD volts. Decoupling is required with or without the internal

reference and buffer. Refer to the Voltage Reference Input section.

38 REFGND AI Reference Input Analog Ground.

39 IN− AI Differential Negative Analog Input.

40 NC No Connect.

43 IN+ AI Differential Positive Analog Input.

45 TEMP AO Temperature Sensor Analog Output.

46 REFBUFIN AI/O Internal Reference Output/Reference Buffer Input.

When PDREF/PDBUF = low, the internal reference and buffer are enabled producing the 1.2 V (typical)

band gap output on this pin, which needs external decoupling. The internal fixed gain reference

buffer uses this to produce 2.048 V on the REF pin.

When using an external reference with the internal reference buffer (PDBUF = low, PDREF = high),

applying 1.2 V on this pin produces 2.048 V on the REF pin. Refer to the Voltage Reference Input section.

47 PDREF DI Internal Reference Power-Down Input.

When low, the internal reference is enabled.

When high, the internal reference is powered down and an external reference must been used.

48 PDBUF DI Internal Reference Buffer Power-Down Input.

When low, the buffer is enabled (must be low when using internal reference).

When high, the buffer is powered-down.

1 AI = analog input; AI/O = bidirectional analog; AO = analog output; DI = digital input; DI/O = bidirectional digital; DO = digital output; P = power.

Table 7. Data Bus Interface Definition

MODE MODE1 MODE0 D0/OB/2C D1/A0 D2/A1 D[3] D[4:9] D[10:11] D[12:15] D[16:17] Description

0 0 0 R[0] R[1] R[2] R[3] R[4:9] R[10:11] R[12:15] R[16:17] 18-Bit Parallel

1 0 1 OB/2C A0 = 0 R[2] R[3] R[4:9] R[10:11] R[12:15] R[16:17] 16-Bit High Word

1 0 1 OB/2C A0 = 1 R[0] R[1] All Zeros 16-Bit Low Word

2 1 0 OB/2C A0 = 0 A1 = 0 All Hi-Z R[10:11] R[12:15] R[16:17] 8-Bit High Byte

2 1 0 OB/2C A0 = 0 A1 = 1 All Hi-Z R[2:3] R[4:7] R[8:9] 8-Bit Mid Byte

2 1 0 OB/2C A0 = 1 A1 = 0 All Hi-Z R[0:1] All Zeros 8-Bit Low Byte

2 1 0 OB/2C A0 = 1 A1 = 1 All Hi-Z All Zeros R[0:1] 8-Bit Low Byte

3 1 1 OB/2C All Hi-Z Serial Interface Serial Interface

AD7641

Rev. 0 | Page 11 of 28

TERMINOLOGY

Integral Nonlinearity Error (INL)

Linearity error refers to the deviation of each individual code

from a line drawn from negative full scale through positive full

scale. The point used as negative full scale occurs ½ LSB before

the first code transition. Positive full scale is defined as a level

1½ LSB beyond the last code transition. The deviation is

measured from the middle of each code to the true straight line.

Differential Nonlinearity Error (DNL)

In an ideal ADC, code transitions are 1 LSB apart. Differential

nonlinearity is the maximum deviation from this ideal value. It

is often specified in terms of resolution for which no missing

codes are guaranteed.

Gain Error

The first transition (from 000…00 to 000…01) should occur for

an analog voltage ½ LSB above the nominal negative full scale

(−2.0479922 V for the ±2.048 V range). The last transition

(from 111…10 to 111…11) should occur for an analog voltage

1½ LSB below the nominal full scale (+2.0479766 V for the

±2.048 V range). The gain error is the deviation of the

difference between the actual level of the last transition and the

actual level of the first transition from the difference between

the ideal levels.

Zero Error

The zero error is the difference between the ideal midscale

input voltage (0 V) and the actual voltage producing the

midscale output code.

Dynamic Range

It is the ratio of the rms value of the full scale to the rms noise

measured with the inputs shorted together. The value for

dynamic range is expressed in decibels.

Signal-to-Noise Ratio (SNR)

SNR is the ratio of the rms value of the actual input signal to the

rms sum of all other spectral components below the Nyquist

frequency, excluding harmonics and dc. The value for SNR is

expressed in decibels.

Total Harmonic Distortion (THD)

THD is the ratio of the rms sum of the first five harmonic

components to the rms value of a full-scale input signal and is

expressed in decibels.

Signal to (Noise + Distortion) Ratio (SINAD)

SINAD is the ratio of the rms value of the actual input signal to

the rms sum of all other spectral components below the Nyquist

frequency, including harmonics but excluding dc. The value for

SINAD is expressed in decibels.

Spurious-Free Dynamic Range (SFDR)

The difference, in decibels (dB), between the rms amplitude of

the input signal and the peak spurious signal.

Effective Number of Bits (ENOB)

ENOB is a measurement of the resolution with a sine wave

input. It is related to SINAD and is expressed in bits by

ENOB = [(SINADdB − 1.76)/6.02]

Aperture Delay

Aperture delay is a measure of the acquisition performance and

is measured from the falling edge of the CNVST input to when

the input signal is held for a conversion.

Transi ent Response

The time required for the AD7641 to achieve its rated accuracy

after a full-scale step function is applied to its input.

Reference Voltage Temperature Coefficient

It is derived from the typical shift of output voltage at 25°C on a

sample of parts maximum and minimum reference output

voltage (VREF) measured at TMIN, T(25°C), and TMAX. It is

expressed in ppm/°C using

()

(

)

()

C25

Cppm/ ×

−×°

−

=°

REF

REFREF

REF TTV

MinVMaxV

TCV

where:

VREF (Max) = Maximum VREF at TMIN, T(25°C), or TMAX

VREF (Min) = Minimum VREF at TMIN, T(25°C), or TMAX

VREF (25°C) = VREF at 25°C

TMAX = +85°C

TMIN = –40°C

AD7641

Rev. 0 | Page 12 of 28

TYPICAL PERFORMANCE CHARACTERISTICS

3.0

2.5

2.0

–3.0

–2.5

0 262144

CODE

INL (LSB)

04761-005

1.5

1.0

0.5

–0.5

–1.0

–2.0

–1.5

65536 131072 196608

Figure 5. Integral Nonlinearity vs. Code

40000

00 0392322

11 683

272

1FFEF

1FFF0

1FFF1

1FFF2

1FFF3

1FFF4

1FFF5

1FFF6

1FFF7

1FFF8

1FFF9

1FFFA

1FFFB

1FFFC

1FFFD

1FFFE

1FFFF

20000

04761-006

CODE IN HEX

COUNTS

35000

30000

25000

20000

15000

10000

5000

34844

31003

24739

16207

5248

2124

10938

4730

σ = 1.55

Figure 6. Histogram of 261,120 Conversions of a DC Input at

the Code Center (External Reference)

2.0530

2.0480

–55 125

04761-007

TEMPERATURE (°C)

REF

(V)

2.0525

2.0520

2.0515

2.0510

2.0505

2.0500

2.0495

2.0490

2.0485

–35 –15 5 25 45 65 85 105

Figure 7. Typical Reference Voltage Output vs. Temperature (3 Units)

2.0

–2.0

0 262144

CODE

DNL (LSB)

04761-008

1.5

1.0

0.5

–0.5

–1.0

–1.5

65536 131072 196608

Figure 8. Differential Nonlinearity vs. Code

30000

1FED9

1FEDB

1FEDD

1FEDF

1FEE1

1FEE3

1FEE5

1FEE7

1FEE9

1FEEB

1FEDA

1FEDC

1FEDE

1FEE0

1FEE2

1FEE4

1FEE6

1FEE8

1FEEA

04761-009

CODE IN HEX

COUNTS

25000

20000

15000

10000

5000

52 117

912

2105

10846

12922

22225

24731

23436

18995

8219

4688

1171

58951 10 1

σ = 2.02

Figure 9. Histogram of 261,120 Conversions of a DC Input at

the Code Center (Internal Reference)

–20

–55 125

04761-010

TEMPERATURE (°C)

ZERO-ERROR, FULL-SCALE ERROR (LSB)

–2

–4

–6

–8

–10

–12

–14

–16

–18

–35 –15 5 25 45 65 85 105

ZERO-ERROR

–FS

+FS

Figure 10. Zero Error, Positive and Negative Full Scale vs. Temperature

AD7641

Rev. 0 | Page 13 of 28

FREQUENCY (kHz)

AMPLITUDE (dB of Full Scale)

–180

000

04761-011

–20

–40

–60

–80

–100

–120

–140

–160

100 200 300 400 500 600 700 800 900

= 2MSPS

= 20.1kHz

SNR = 93.6dB

THD = –116dB

SFDR = 112dB

SINAD = 93.5dB

–180

01000

FREQUENCY (kHz)

AMPLITUDE (dB of Full Scale)

Figure 11. FFT 20 kHz

1 1000

FREQUENCY (kHz)

SNR, SINAD (dB)

ENOB (Bits)

04761-012

16.0

12.0

15.6

15.2

14.8

14.4

14.0

13.6

13.2

12.8

12.4

ENOB SINAD SNR

100

Figure 12. SNR, SINAD, and ENOB vs. Frequency

–

–140

1 1000

FREQUENCY (kHz)

THD, HARMONICS (dB)

SFDR (dB)

04761-013

–80

–90

–100

–110

–120

–130

120

110

100

10 100

SFDR

THD

THIRD

HARMONIC

SECOND

HARMONIC

Figure 13. THD, Harmonics, and SFDR vs. Frequency

04761-014

= 2MSPS

= 100.8kHz

SNR = 93dB

THD = –101dB

SFDR = 101dB

SINAD = 92.5dB

–20

–40

–60

–80

–100

–120

–140

–160

100 200 300 400 500 600 700 800 900

Figure 14. FFT 100 kHz

–55 125

TEMPERATURE (°C)

SNR, SINAD (dB)

04761-015

ENOB (Bits)

–35 –15 5 25 45 65 85 105

SNR

ENOB

SINAD

Figure 15. SNR, SINAD, and ENOB vs. Temperature

–

–140

–55 125

TEMPERATURE (°C)

THD, HARMONICS (dB)

04761-016

–95

–100

–105

–110

–115

–120

–125

–130

–135

120

SFDR (dB)

115

110

105

100

–35 –15 5 25 45 65 85 105

SFDR

THD

THIRD

HARMONIC

SECOND

HARMONIC

Figure 16. THD, Harmonics, and SFDR vs. Temperature

AD7641

Rev. 0 | Page 14 of 28

96.0

93.0

–60 0

INPUT LEVEL (dB)

SNR, SINAD REFERRED TO FULL SCALE (dB)

04761-017

95.5

95.0

94.5

94.0

93.5

–50 –40 –30 –20 –10

SNR

SINAD

Figure 17. SNR and SINAD vs. Input Level (Referred to Full Scale)

–55 125

TEMPERATURE (°C)

DVDD, OVDD (µA)

04761-018

–35 –15 5 25 45 65 85 105

DVDD

OVDD, 3.3V

OVDD, 2.5V

AVDD

Figure 18. Power-Down Operating Currents vs. Temperature

100000

0.01

10 10M

SAMPLING RATE (SPS)

OPERATING CURRENTS (µA)

04761-019

100 1k 10k 100k 1M

0.1

100

1000

10000

AVDD

DVDD

OVDD = 2.5V, ALL MODES

OVDD = 3.3V, ALL MODES

PDREF = PDBUF = HIGH

Figure 19. Operating Current vs. Sample Rate

04761-020

CL (pF)

12 DELAY (ns)

50 100 150 200

OVDD = 2.5V @ 85°C

OVDD = 3.3V @ 85°C

OVDD = 3.3V @ 25°C

OVDD = 2.5V @ 25°C

Figure 20. Typical Delay vs. Load Capacitance CL

AD7641

Rev. 0 | Page 15 of 28

APPPLICATIONS INFORMATION

SW+

COMP

SW–

IN+

REF

REFGND

LSB

MSB

131,072C 65,536C 4C 2C C C

CONTROL

LOGIC

SWITCHES

CONTROL

BUSY

OUTPUT

CODE

CNVST

IN–

4C 2C C C

LSB

MSB

AGND

131,072C 65,536C

04761-021

Figure 21. ADC Simplified Schematic

CIRCUIT INFORMATION

The AD7641 is a very fast, low power, single-supply, precise

18-bit ADC using successive approximation architecture. The

AD7641 features different modes to optimize performances

according to the applications. In warp mode, the AD7641 is

capable of converting 2,000,000 samples per second (2 MSPS).

The AD7641 provides the user with an on-chip track-and-hold,

successive approximation ADC that does not exhibit any

pipeline or latency, making it ideal for multiple multiplexed

channel applications.

The AD7641 can operate from a single 2.5 V supply and

interface to either 5 V, 3.3 V, or 2.5 V digital logic. It is housed

in a 48-lead LQFP package or a tiny 48-lead LFCSP package,

which combines space savings with flexibility and allows the

AD7641 to be configured as either a serial or a parallel

interface. The AD7641 is pin-to-pin-compatible and is a

speed upgrade of the AD7674, AD7678, and AD7679.

CONVERTER OPERATION

The AD7641 is a successive approximation ADC based on a

charge redistribution DAC. Figure 21 shows the simplified

schematic of the ADC. The capacitive DAC consists of two

identical arrays of 16 binary weighted capacitors that are

connected to the two comparator inputs.

During the acquisition phase, terminals of the array tied to the

comparator’s input are connected to AGND via SW+ and SW−.

All independent switches are connected to the analog inputs.

Therefore, the capacitor arrays are used as sampling capacitors

and acquire the analog signal on the IN+ and IN− inputs. A

conversion phase is initiated once the acquisition phase is complete

and the CNVST input goes low. When the conversion phase

begins, SW+ and SW− are opened first. The two capacitor

arrays are then disconnected from the inputs and connected to

the REFGND input. Therefore, the differential voltage between

the inputs (IN+ and IN−) captured at the end of the acquisition

phase is applied to the comparator inputs, causing the

comparator to become unbalanced. By switching each element

of the capacitor array between REFGND and REF, the comparator

input varies by binary weighted voltage steps (VREF/2, VREF/4

throughVREF/131072). The control logic toggles these switches,

starting with the MSB first, to bring the comparator back into a

balanced condition. After the completion of this process, the

control logic generates the ADC output code and brings BUSY

output low.

MODES OF OPERATION

The AD7641 features three modes of operations: wideband

warp, warp, and normal. Each of these modes is more suitable

for specific applications.

The wideband warp (WARP = high, NORMAL = high) and

warp (WARP = high, NORMAL = low) modes allow the fastest

conversion rate of up to 2 MSPS. However, in these modes, the

full specified accuracy is guaranteed only when the time between

conversions does not exceed 1 ms. If the time between two

consecutive conversions is longer than 1 ms (for instance after

power-up), the first conversion result should be ignored. These

modes make the AD7641 ideal for applications where both high

accuracy and fast sample rates are required. Wideband warp

mode offers slightly improved linearity and THD over warp mode.

Normal mode (NORMAL = low, WARP = low) is the fastest

mode (1.5 MSPS) without any limitation on time between

conversions. This mode makes the AD7641 ideal for

asynchronous applications, such as data acquisition systems,

where both high accuracy and fast sample rates are required.

AD7641

Rev. 0 | Page 16 of 28

TRANSFER FUNCTIONS

Using the OB/2C digital input, except in 18-bit interface mode,

the AD7641 offers two output codings: straight binary and twos

complement. The LSB size with VREF = 2.048 V is 2 × VREF/

262,144, which is 15.623 μV. Refer to Figure 22 and Table 8 for

the ideal transfer characteristic.

04761-022

000...000

000...001

000...010

111. ..101

111...110

111. ..111

ADC CODE (Straight Binary)

ANALOG INPUT

+FSR – 1.5 LSB

+FSR – 1 LSB–FSR + 1 LSB

–FSR

–FSR + 0.5 LSB

Figure 22. ADC Ideal Transfer Function

Table 8. Output Codes and Ideal Input Voltages

Digital Output Code (Hex)

Description

Analog Input

VREF = 2.048 V

Straight

Binary

Twos

Complement

FSR −1 LSB +2.0479844 V 0x3FFFF10x1FFFF1

FSR − 2 LSB +2.0479688 V 0x3FFFE 0x1FFFE

Midscale + 1 LSB +15.625 μV 0x20001 0x00001

Midscale 0 V 0x20000 0x00000

Midscale − 1 LSB −15.625 μV 0x1FFFF 0x3FFFF

−FSR + 1 LSB −2.0479844 V 0x30001 0x20001

−FSR −2.048 V 0x30000 0x20000

2 2

1 This is also the code for overrange analog input (VIN+ − VIN− above

+VREF − VREFGND).

2 This is also the code for underrange analog input (VIN+ − VIN− below

−VREF + VREFGND).

04761-023

100nF 100nF

AVDD

10µF 100nF

AGND DGND DVDD OVDD OGND

CNVST

BUSY

SDOUT

SCLK

RESET

REFBUFIN

10Ω

CLOCK

AD7641

MICROCONVERTER/

MICROPROCESSOR/

DSP

SERIAL

PORT

DIGITAL

INTERFACE

SUPPLY

(2.5V OR 3.3V)

ANALOG

SUPPLY (2.5V)

OVDD

WARP

DIGIT

SUPPLY (2.5V)

IN+

IN–

15Ω

NOTE 5

50Ω

50pF

NOTE 1

ANALOG

INPUT +

2.7nF

15Ω

NOTE 1

MODE0

MODE1

D0/OB/2C

REFGND

REF

PDBUF

PDREF

100nF

ANALOG

INPUT –

NOTE 2

NOTE 3

NOTE 4

NOTE 3

NOTE 7

NOTE 6

NORMAL

10µF

REF

10µF

10kΩ

50pF

1. SEE ANALOG INPUTS SECTION.

. THE AD8021 IS RECOMMENDED. SEE DRIVER AMPLIFIER CHOICE SECTION.

. THE CONFIGURATION SHOWN IS USING THE INTERNAL REFERENCE. SEE VOLTAGE REFERENCE INPUT SECTION.

. A 10µF CERAMIC CAPACITOR (X5R, 1206 SIZE) IS RECOMMENDED (FOR EXAMPLE, PANASONIC ECJ3YB0J106M).

SEE VOLTAGE REFERENCE INPUT SECTION.

5. OPTION, SEE POWER SUPPLY SECTION.

6. OPTION, SEE POWER-UP SECTION.

7. OPTIONAL LOW JITTER CNVST, SEE CONVERSION CONTROL SECTION.

Figure 23. Typical Connection Diagram

AD7641

Rev. 0 | Page 17 of 28

TYPICAL CONNECTION DIAGRAM

Figure 23 shows a typical connection diagram for the AD7641.

Different circuitry shown in this diagram is optional and is

discussed in the following sections.

ANALOG INPUTS

Figure 24 shows an equivalent circuit of the input structure of

the AD7641.

The two diodes, D1 and D2, provide ESD protection for the

analog inputs IN+ and IN−. Care must be taken to ensure that

the analog input signal never exceeds the supply rails by more

than 0.3 V, because this causes the diodes to become forward-

biased and start conducting current. These diodes can handle a

forward-biased current of 100 mA maximum. For instance,

these conditions could eventually occur when the input buffer’s

U1 or U2 supplies are different from AVDD. In such a case, an

input buffer with a short-circuit current limitation can be used

to protect the part.

4761-024

IN+ OR IN–

AGND

Figure 24. AD7641 Simplified Analog Input

The analog input of the AD7641 is a true differential structure.

By using this differential input, small signals common to both

inputs are rejected, as shown in Figure 25, representing the

typical CMRR over frequency with internal and external references.

1 10000

FREQUENCY (kHz)

CMRR (dB)

04761-025

10 100 1000

INT REF

EXT REF

Figure 25. Analog Input CMRR vs. Frequency

During the acquisition phase for ac signals, the impedance of

the analog inputs, IN+ and IN−, can be modeled as a parallel

combination of capacitor CPIN and the network formed by the

series connection of RIN and CIN. CPIN is primarily the pin

capacitance. RIN is typically 175 Ω and is a lumped component

comprised of some serial resistors and the on resistance of the

switches. CIN is typically 12 pF and is mainly the ADC sampling

capacitor. During the conversion phase, when the switches are

opened, the input impedance is limited to CPIN. RIN and CIN

make a 1-pole, low-pass filter that has a typical −3 dB cutoff

frequency of 50 MHz, thereby reducing an undesirable aliasing

effect and limiting the noise coming from the inputs.

Because the input impedance of the AD7641 is very high, the

AD7641 can be directly driven by a low impedance source

without gain error. To further improve the noise filtering achieved

by the AD7641 analog input circuit, an external 1-pole RC filter

between the amplifier’s outputs and the ADC analog inputs can

be used, as shown in Figure 23. However, large source impedances

significantly affect the ac performance, especially the total

harmonic distortion (THD). The maximum source impedance

depends on the amount of THD that can be tolerated. The THD

degrades as a function of the source impedance and the maximum

input frequency.

MULTIPLEXED INPUTS

When using the full 2 MSPS throughput in multiplexed

applications for a full-scale step, the RC filter, as shown in

Figure 23, does not settle in the required acquisition time, t8.

These values are chosen to optimize the best SNR perform-ance

of the AD7641. To use the full 2 MSPS throughput in

multiplexed applicaitons, the RC should be adjusted to satisfy t8

(which is ~ 8.5 × RC time constant). However, lowering R and C

increases the RC filter bandwidth and allows more noise into the

AD7641, which degrades SNR. To preserve the SNR performance

in these applications using the RC filter shown in Figure 23,

the AD7641 should be run with t8 > 350 ns; or approximately

1/(t7 + t8) ~ 1.35 MSPS in wideband and warp modes.

DRIVER AMPLIFIER CHOICE

Although the AD7641 is easy to drive, the driver amplifier

needs to meet the following requirements:

• For multichannel, multiplexed applications, the driver

amplifier and the AD7641 analog input circuit must be

able to settle for a full-scale step of the capacitor array at an

18-bit level (0.0004%). In the amplifier’s data sheet, settling

at 0.1% to 0.01% is more commonly specified. This could

differ significantly from the settling time at a 18-bit level

and should be verified prior to driver selection. The

AD8021 op amp, which combines ultralow noise and high

gain bandwidth, meets this settling time requirement even

when used with gains up to 13.

AD7641

Rev. 0 | Page 18 of 28

Single-to-Differential Driver

• The noise generated by the driver amplifier needs to be

kept as low as possible to preserve the SNR and transition

noise performance of the AD7641. The noise coming from

the driver is filtered by the AD7641 analog input circuit

1-pole, low-pass filter made by RIN and CIN or by the

external filter, if one is used. The SNR degradation due

to the amplifier is

For applications using unipolar analog signals, a single-ended-

to-differential driver, as shown in Figure 26, allows for a

differential input into the part. This configuration, when

provided an input signal of 0 to VREF, produces a differential

±VREF with midscale at VREF/2. The 1-pole filter using R = 10 Ω

and C = 1 nF provides a corner frequency of 16 MHz.

() ()

⎟

⎠

⎞

⎜

⎝

⎛

−

−22

900

20log

3dB

LOSS

SNR

If the application can tolerate more noise, the AD8139

differential driver can be used.

4761-027

AD8021

ANALOG INPUT

(

UNIPOLAR 0

TO 2.048V)

AD8021

IN+

IN–

AD7641

REF

10µF

15Ω

100nF

2.7nF

10pF

5kΩ

590Ω

where:

f–3dB is the input bandwidth of the AD7641 (50 MHz) or

the cutoff frequency of the input RC filter shown in Figure 23

(3.9 MHz), if one is used.

N is the noise factor of the amplifier (1 in buffer

configuration).

eN+ and eN− are the equivalent input voltage noise densities

of the op amps connected to IN+ and IN−, in nV/√Hz.

This approximation can be used when the resistances used

around the amplifier are small. If larger resistances are

used, their noise contributions should also be root-sum

squared.

Figure 26. Single-Ended-to-Differential Driver Circuit

(Internal Reference Buffer Used)

VOLTAGE REFERENCE INPUT

The AD7641 allows the choice of either a very low temperature

drift internal voltage reference or an external reference.

For instance, when using op amps with an equivalent input

noise density of 2.1 nV/√Hz, such as the AD8021, with a

noise gain of 1 when configured as a buffer, degrades the

SNR by only 0.25 dB when using the RC filter in

Unlike many ADCs with internal references, the internal

reference of the AD7641 provides excellent performance and

can be used in almost all applications.

Figure 23,

and by 2.5 dB without it.

Internal Reference (PDBUF = Low, PDREF = Low)

• The driver needs to have a THD performance suitable to

that of the AD7641. Figure 13 gives the THD vs. frequency

that the driver should exceed.

To use the internal reference, the PDREF and PDBUF inputs

must both be low. This produces a 1.2 V band gap output on

REFBUFIN, which is amplified by the internal buffer and

results in a 2.048 V reference on the REF pin.

The AD8021 meets these requirements and is appropriate for

almost all applications. The AD8021 needs a 10 pF external

compensation capacitor that should have good linearity as an

NPO ceramic or mica type. Moreover, the use of a noninverting

1 gain arrangement is recommended and helps to obtain the

best signal-to-noise ratio.

The internal reference is temperature compensated to 2.048 V ±

10 mV. The reference is trimmed to provide a typical drift of

10 ppm/°C. This typical drift characteristic is shown in Figure 7.

The output resistance of REFBUFIN is 6.33 kΩ (minimum)

when the internal reference is enabled. It is necessary to

decouple this with a ceramic capacitor greater than 100 nF.

Therefore, the capacitor provides an RC filter for noise reduction.

The AD8022 can also be used when a dual version is needed

and a gain of 1 is present. The AD829 is an alternative in

applications where high frequency (above 100 kHz) performance is

not required. In applications with a gain of 1, an 82 pF

compensation capacitor is required. The Because the output impedance of REFBUFIN is typically

6.33 kΩ, relative humidity (among other industrial contaminates)

can directly affect the drift characteristics of the reference.

Typically, a guard ring is used to reduce the effects of drift

under such circumstances.

AD8610 is an option

when low bias current is needed in low frequency applications.

AD7641

Rev. 0 | Page 19 of 28

However, because the AD7641 has a fine lead pitch, guarding

this node is not practical. Therefore, in these industrial and

other types of applications, it is recommended to use a conformal

coating, such as Dow Corning® 1-2577 or HumiSeal® 1B73.

For applications that use multiple AD7641 devices, it is more

effective to use an external reference with the internal reference

buffer to buffer the reference voltage. However, because the

reference buffers are not unity gain, ratiometric, simultaneously

sampled designs should use an external reference and external

buffer, such as the

External 1.2 V Reference and Internal Buffer (PDBUF =

Low, PDREF = High) AD8031/AD8032; therefore, preserving the

same reference level for all converters.

To use an external reference along with the internal buffer,

PDREF should be high and PDBUF should be low. This powers

down the internal reference and allows the 1.2 V reference to

be applied to REFBUFIN, producing 2.048 V (typically) on

the REF pin.

The voltage reference temperature coefficient (TC) directly

impacts full scale; therefore, in applications where full-scale

accuracy matters, care must be taken with the TC. For instance,

a ±4 ppm/°C TC of the reference changes full scale by ±1 LSB/°C.

External 2.5 V Reference (PDBUF = High, PDREF = High) Note that VREF can be increased to AVDD + 0.1 V. Because the

input range is defined in terms of VREF, this would essentially

increase the range to 0 V to 2.8 V with an AVDD = 2.7 V.

To use an external 2.5 V reference directly on the REF pin,

PDREF and PDBUF should both be high.

For improved drift performance, an external reference, such as

the Temperature Sensor

AD780 or ADR431, can be used. The advantages of directly

using the external voltage reference are: The TEMP pin measures the temperature of the AD7641. To

improve the calibration accuracy over the temperature range,

the output of the TEMP pin is applied to one of the inputs of

the analog switch (such as,

• The SNR and dynamic range improvement (about 1.7 dB)

resulting from the use of a reference voltage very close to

the supply (2.5 V) instead of a typical 2.048 V reference

when the internal reference is used. This is calculated by

ADG779), and the ADC itself is

used to measure its own temperature. This configuration is

shown in Figure 27.

⎟

⎠

⎞

⎜

⎝

⎛

=50.2

048.2

log20SNR

04761-028

ADG779

AD8021 C

ANALOG INPUT

(UNIPOLAR)

AD7641

IN+ TEMPERATURE

SENSOR

TEMP

• The power savings when the internal reference is powered

down (PDREF high).

PDREF and PDBUF power down the internal reference and

the internal reference buffer, respectively. The input current

of PDREF and PDBUF should never exceed 20 mA. This can

occur when the driving voltage is above AVDD (for instance, at

power-up). In this case, a 125 Ω series resistor is recommended.

Figure 27. Use of the Temperature Sensor

Reference Decoupling

Whether using an internal or external reference, the AD7641

voltage reference input (REF) has a dynamic input impedance;

therefore, it should be driven by a low impedance source with

efficient decoupling between the REF and REFGND inputs.

This decoupling depends on the choice of the voltage reference

but usually consists of a low ESR capacitor connected to REF

and REFGND with minimum parasitic inductance. A 10 μF

(X5R, 1206 size) ceramic chip capacitor (or 47 μF tantalum

capacitor) is appropriate when using either the internal

reference or one of the recommended reference voltages.

The placement of the reference decoupling is also important to

the performance of the AD7641. The decoupling capacitor

should be mounted on the same side as the ADC right at the

REF pin with a thick PCB trace. The REFGND should also connect

to the reference decoupling capacitor with the shortest distance.

AD7641

Rev. 0 | Page 20 of 28

POWER SUPPLY It should be noted that the digital interface remains active even

during the acquisition phase. To reduce the operating digital

supply currents even further, drive the digital inputs close to

the power rails (that is, OVDD and OGND).

The AD7641 uses three sets of power supply pins: an analog

2.5 V supply AVDD, a digital 2.5 V core supply DVDD, and a

digital input/output interface supply OVDD. The OVDD supply

allows direct interface with any logic working between 2.3 V

and 5.25 V. To reduce the number of supplies needed, the digital

core (DVDD) can be supplied through a simple RC filter from

the analog supply, as shown in

CONVERSION CONTROL

The AD7641 is controlled by the CNVST input. A falling edge

on CNVST is all that is necessary to initiate a conversion.

Detailed timing diagrams of the conversion process are shown

Figure 23.

Power Sequencing Figure 29. Once initiated, it cannot be restarted or aborted,

even by the power-down input, PD, until the conversion is

complete. The

The AD7641 is independent of power supply sequencing and

thus free from supply induced voltage latch-up. In addition, it is

very insensitive to power supply variations over a wide

frequency range, as shown in Figure 28.

65.0

45.0

1 10000

FREQUENCY (MHz)

PSRR (dB)

04761-029

10 100 1000

62.5

60.0

57.5

55.0

52.5

50.0

47.5

INT REF

EXT REF

Figure 28. PSRR vs. Frequency

Power-Up

At power-up, or returning to operational mode from the power-

down mode (PD = high), the AD7641 engages an initialization

process. During this time, the first 128 conversions should be

ignored or the RESET input could be pulsed to engage a faster

initialization process. Refer to the Digital Interface section for

RESET and timing details.

A simple power-on reset circuit, as shown in Figure 23, can be

used to minimize the digital interface. As OVDD powers up, the

capacitor is shorted and brings RESET high; it is then charged

returning RESET to low. However, this circuit only works when

powering up the AD7641 because the power-down mode

(PD = high) does not power down any of the supplies and as a

result, RESET is low.

CNVST signal operates independently of CS and

RD signals.

04761-030

BUSY

MODE CONVERT ACQUIREACQUIRE CONVERT

CNVST

Figure 29. Basic Conversion Timing

CNVST

For optimal performance, the rising edge of should not

occur after the maximum CNVST low time, t1, or until the end

of conversion.

Although CNVST is a digital signal, it should be designed with

special care with fast, clean edges and levels with minimum

overshoot and undershoot or ringing.

The CNVST trace should be shielded with ground and a low

value serial resistor (for example, 50 Ω) termination should be

added close to the output of the component that drives this line.

In addition, a 50 pF capacitor is recommended to further reduce

the effects of overshoot and undershoot as shown in Figure 23.

For applications where SNR is critical, the CNVST signal should

have very low jitter. This can be achieved by using a dedicated

oscillator for CNVST generation, or by clocking CNVST with a

high frequency, low jitter clock, as shown in Figure 23.

AD7641

Rev. 0 | Page 21 of 28

INTERFACES

DIGITAL INTERFACE PARALLEL INTERFACE

The AD7641 has a versatile digital interface that can be set up

as either a serial or a parallel interface with the host system. The

serial interface is multiplexed on the parallel data bus. The AD7641

digital interface also accommodates 2.5 V, 3.3 V, or 5 V logic

with either OVDD at 2.5 V or 3.3 V. OVDD defines the logic

high output voltage. In most applications, the OVDD supply pin

of the AD7641 is connected to the host system interface 2.5 V

or 3.3 V digital supply. By using the D0/OB/

The AD7641 is configured to use the parallel interface for an

18-bit, 16-bit, or 8-bit bus width according to Table 7.

Master Parallel Interface

2C input pin, either

twos complement or straight binary coding can be used.

The two signals CS and RD control the interface. When at least

one of these signals is high, the interface outputs are in high

impedance. Usually, CS allows the selection of each AD7641 in

multicircuit applications and is held low in a single AD7641

design. RD is generally used to enable the conversion result on

the data bus.

RESET

The RESET input is used to reset the AD7641 and generate a

fast initialization. A rising edge on RESET aborts the current

conversion (if any) and tristates the data bus. The falling edge of

RESET clears the data bus and engages the initialization process

indicated by pulsing BUSY high. Conversions can take place

after the falling edge of BUSY. Refer to Figure 30 for the RESET

timing details.

4761-031

RESET

DATA

BUSY

CNVST

Figure 30. RESET Timing

Data can be continuously read by tying CS RD

and low, thus

requiring minimal microprocessor connections. However, in

this mode, the data bus is always driven and cannot be used in

shared bus applications, unless the device is held in RESET.

Figure 31 details the timing for this mode.

04761-032

BUSY

DATA

BUS PREVIOUS CONVERSION DATA NEW DATA

CNVST

CS = RD = 0

Figure 31. Master Parallel Data Timing for Reading (Continuous Read)

Slave Parallel Interface

In slave parallel reading mode, the data can be read either after

each conversion, which is during the next acquisition phase, or

during the following conversion, as shown in Figure 32 and

Figure 33, respectively. When the data is read during the

conversion, it is recommended that it is read-only during the

first half of the conversion phase. This avoids any potential

feedthrough between voltage transients on the digital interface

and the most critical analog conversion circuitry.

04761-033

CURRENT

CONVERSION

BUSY

DATA

BUS

Figure 32. Slave Parallel Data Timing for Reading (Read After Convert)

AD7641

Rev. 0 | Page 22 of 28

45761-034

CONVERSION

t13

t12

BUSY

DATA

BUS

CNVST,

CS = 0

SERIAL INTERFACE

The AD7641 is configured to use the serial interface when

MODE[1:0] = 3. The AD7641 outputs 18 bits of data, MSB first,

on the SDOUT pin. This data is synchronized with the 18 clock

pulses provided on the SCLK pin. The output data is valid on

both the rising and falling edge of the data clock.

MASTER SERIAL INTERFACE

Internal Clock

The AD7641 is configured to generate and provide the serial

data clock SCLK when the EXT/INT pin is held low. The

AD7641 also generates a SYNC signal to indicate to the host

when the serial data is valid. The serial clock SCLK and the

SYNC signal can be inverted. Depending on the read during

convert input, RDC/SDIN, the data can be read after each

conversion or during the following conversion.

Figure 33. Slave Parallel Data Timing for Reading (Read During Convert)

16-Bit and 8-Bit Interface (Master or Slave)

In the 16-bit (MODE[1:0] = 1) and 8-bit (MODE[1:0] = 2)

interfaces, the A0/A1 pins allow a glueless interface to a 16- or

8-bit bus, as shown in Figure 35 and

Figure 34. By connecting A0/A1 to an

address line(s), the data can be read in two words for a 16-bit

interface, or three bytes for an 8-bit interface. This interface can

be used in both master and slave parallel reading modes. Refer

Figure 36 show detailed timing diagrams of these two modes.

Usually, because the AD7641 is used with a fast throughput, the

master read during conversion mode is the most recommended

serial mode. In this mode, the serial clock and data toggle at

appropriate instants, minimizing potential feedthrough between

digital activity and critical conversion decisions. In this mode,

the SCLK period changes because the LSBs require more time

to settle and the SCLK is derived from the SAR conversion cycle.

Table 7 for the full details of the interface.

CS, RD

D[17:2] HI-Z HIGH

WORD

LOW

WORD

HI-Z

04761-035

HIGH

BYTE

MID

BYTE

LOW

BYTE

D[17:10]

HI-Z HI-Z

In read after conversion mode, it should be noted that unlike

other modes, the BUSY signal returns low after the 18 data bits

are pulsed out and not at the end of the conversion phase,

resulting in a longer BUSY width. As a result, the maximum

throughput cannot be achieved in this mode.

In addition, in read after convert mode, the SCLK frequency

can be slowed down to accommodate different hosts using the

DIVSCLK[1:0] inputs. Refer to

Figure 34. 8-Bit and 16-Bit Parallel Interface Table 4 for the SCLK timing

details when using these inputs.

AD7641

Rev. 0 | Page 23 of 28

04761-036

BUSY

SYNC

SCLK

DOUT

123 161718

D17 D16 D2 D1 D0

RDC/SDIN = 0 INVSCLK = INVSYNC = 0

CNVST

CS, RD

EXT/INT = 0

DIVSCLK[1:0] = 0

Figure 35. Master Serial Data Timing for Reading (Read After Convert)

04761-037

EXT/INT = 0 RDC/SDIN = 1 INVSCLK = INVSYNC = 0

D17 D16 D2 D1 D0X

123 161718

BUSY

SYNC

SCLK

SDOUT

CNVST

CS, RD

Figure 36. Master Serial Data Timing for Reading (Read Previous Conversion During Convert)

AD7641

Rev. 0 | Page 24 of 28

SLAVE SERIAL INTERFACE

External Clock An example of the concatenation of two devices is shown in

Figure 37. Simultaneous sampling is possible by using a

common

The AD7641 is configured to accept an externally supplied

serial data clock on the SCLK pin when the EXT/ CNVST signal. It should be noted that the RDC/SDIN

input is latched on the edge of SCLK opposite to the one used to

shift out the data on SDOUT. Therefore, the MSB of the

upstream converter just follows the LSB of the downstream

converter on the next SCLK cycle.

INT pin is

held high. In this mode, several methods can be used to read

the data. The external serial clock is gated by CS. When CS and

RD are both low, the data can be read after each conversion or

during the following conversion. The external clock can be

either a continuous or a discontinuous clock. A discontinuous

clock can be either normally high or normally low when

inactive.

4761-038

SCLK

SDOUTRDC/SDIN

AD7641

(DOWNSTREAM)

AD7641

(UPSTREAM)

BUSY

OUT

BUSYBUSY

DATA

OUT

SCLK

RDC/SDIN SDOUT

SCLK IN

CNVST IN

CNVST

CS IN

Figure 38 and Figure 39 show the detailed timing

diagrams of these methods.

While the AD7641 is performing a bit decision, it is important

that voltage transients be avoided on digital input/output pins

or degradation of the conversion result could occur. This is

particularly important during the second half of the conversion

phase because the AD7641 provides error correction circuitry

that can correct for an improper bit decision made during

the first half of the conversion phase. For this reason, it is

recommended that when an external clock is being provided,

a discontinuous clock is toggled only when BUSY is low or,

more importantly, that it does not transition during the latter

half of BUSY high.

Figure 37.Two AD7641 Devices in a Daisy-Chain Configuration

External Clock Data Read During Previous Conversion

Figure 39 shows the detailed timing diagrams of this method.

During a conversion, while

External Discontinuous Clock Data Read After Conversion

Though the maximum throughput cannot be achieved using

this mode, it is the most recommended of the serial slave

modes. Figure 38 shows the detailed timing diagrams of this

method. After a conversion is complete, indicated by BUSY

returning low, the conversion result can be read while both CS

and RD are low. Data is shifted out MSB first with 18 clock

pulses and is valid on the rising and falling edges of the clock.

Among the advantages of this method is the fact that conversion

performance is not degraded because there are no voltage

transients on the digital interface during the conversion process.

Another advantage is the ability to read the data at any speed up

to 80 MHz, which accommodates both the slow digital host

interface and the fast serial reading.

Finally, in this mode only, the AD7641 provides a daisy-chain

feature using the RDC/SDIN pin for cascading multiple converters

together. This feature is useful for reducing component count

and wiring connections when desired, as, for instance, in

isolated multiconverter applications.

CS RD

and are both low, the result

of the previous conversion can be read. The data is shifted out,

MSB first, with 16 clock pulses and is valid on both the rising

and falling edge of the clock. The 18 bits have to be read before

the current conversion is complete; otherwise, RDERROR is

pulsed high and can be used to interrupt the host interface to

prevent incomplete data reading. There is no daisy-chain

feature in this mode, and the RDC/SDIN input should always

be tied either high or low.

To reduce performance degradation due to digital activity, a fast

discontinuous clock (at least 60 MHz when normal mode is

used, or 80 MHz when warp mode is used) is recommended to

ensure that all the bits are read during the first half of the SAR

conversion phase.

It is also possible to begin to read data after conversion and

continue to read the last bits after a new conversion is initiated.

However, this is not recommended when using the fastest

throughput of any mode because the acquisition time, t8, is

only 115 ns.

If the maximum throughput is not used, thus allowing more

acquisition time, then the use of a slower clock speed can be

used to read the data.

AD7641

Rev. 0 | Page 25 of 28

04761-039

SCLK

DOUT D17 D16 D1 D0

D15

X17 X16 X15 X1 X0 Y17 Y16

BUSY

SDIN

INVSCLK = 0

X17 X16

123 1617181920

EXT/INT = 1

CS RD = 0

Figure 38. Slave Serial Data Timing for Reading (Read After Convert)

04761-040

DOUT

SCLK

D1 D0

XD17 D16 D15

12317 18

BUSY

EXT/INT = 1 INVSCLK = 0

CNVST

RD = 0

Figure 39. Slave Serial Data Timing for Reading (Read Previous Conversion During Convert)

AD7641

Rev. 0 | Page 26 of 28

SPI Interface (ADSP-219x)

MICROPROCESSOR INTERFACING

Figure 40 shows an interface diagram between the AD7641 and

an SPI-equipped DSP, the ADSP-219x. To accommodate the

slower speed of the DSP, the AD7641 acts as a slave device and

data must be read after conversion. This mode also allows the

daisy-chain feature. The convert command can be initiated in

response to an internal timer interrupt. The 18-bit output data

are read with three SPI byte access. The reading process can be

initiated in response to the end-of-conversion signal (BUSY

going low) using an interrupt line of the DSP. The serial

peripheral interface (SPI) on the ADSP-219x is configured for

master mode (MSTR) = 1, clock polarity bit (CPOL) = 0, clock

phase bit (CPHA) = 1, and the SPI interrupt enable (TIMOD) =

00 by writing to the SPI control register (SPICLTx). It should be

noted that to meet all timing requirements, the SPI clock should

be limited to 17 Mb/s, allowing it to read an ADC result in less

than 1 μs. When a higher sampling rate is desired, it is

recommended to use one of the parallel interface modes.

The AD7641 is ideally suited for traditional dc measurement

applications supporting a microprocessor, and ac signal processing

applications interfacing to a digital signal processor. The

AD7641 is designed to interface with a parallel 8-bit or 16-bit

wide interface or with a general-purpose serial port or I/O ports

on a microcontroller. A variety of external buffers can be used

with the AD7641 to prevent digital noise from coupling into the

ADC. The SPI Interface (ADSP-219x) section illustrates the use

of the AD7641 with the ADSP-219x SPI-equipped DSP.

04761-041

BUSY

SDOUT

SCLK

CNVST

AD7641

PFx

SPIxSEL (PFx)

MISOx

SCKx

PFx OR TFSx

ADSP-219x

DVDD

MODE0

MODE1

EXT/INT

INVSCLK

ADDITIONAL PINS OMITTED FOR CLARITY.

Figure 40. Interfacing the AD7641 to ADSP-219x

AD7641

Rev. 0 | Page 27 of 28

APPLICATION HINTS

LAYOUT

While the AD7641 has very good immunity to noise on the

power supplies, exercise care with the grounding layout. To

facilitate the use of ground planes that can be easily separated,

design the printed circuit board that houses the AD7641 so that

the analog and digital sections are separated and confined to

certain areas of the board. Digital and analog ground planes

should be joined in only one place, preferably underneath the

AD7641, or as close as possible to the AD7641. If the AD7641 is

in a system where multiple devices require analog-to-digital

ground connections, the connections should still be made at

one point only, a star ground point, established as close as

possible to the AD7641.

To prevent coupling noise onto the die, to avoid radiating noise,

and to reduce feedthrough:

• Do not run digital lines under the device.

• Run the analog ground plane under the AD7641.

• Shield fast switching signals, like CNVST or clocks, with

digital ground to avoid radiating noise to other sections of

the board, and never run them near analog signal paths.

• Avoid crossover of digital and analog signals.

• Run traces on different but close layers of the board, at right

angles to each other, to reduce the effect of feedthrough

through the board.

The power supply lines to the AD7641 should use as large a

trace as possible to provide low impedance paths and reduce the

effect of glitches on the power supply lines. Good decoupling is

also important to lower the impedance of the supplies presented

to the AD7641, and to reduce the magnitude of the supply

spikes. Decoupling ceramic capacitors, typically 100 nF, should

be placed on each of the power supplies pins, AVDD, DVDD,

and OVDD. The capacitors should be placed close to, and

ideally right up against, these pins and their corresponding

ground pins. Additionally, low ESR 10 μF capacitors should be

located in the vicinity of the ADC to further reduce low

frequency ripple.

The DVDD supply of the AD7641 can be either a separate

supply or come from the analog supply, AVDD, or from the

digital interface supply, OVDD. When the system digital supply

is noisy, or fast switching digital signals are present, and no

separate supply is available, it is recommended to connect the

DVDD digital supply to the analog supply AVDD through an

RC filter, and to connect the system supply to the interface

digital supply OVDD and the remaining digital circuitry. Refer

to Figure 23 for an example of this configuration. When DVDD

is powered from the system supply, it is useful to insert a bead

to further reduce high frequency spikes.

The AD7641 has four different ground pins: REFGND, AGND,

DGND, and OGND. REFGND senses the reference voltage and,

because it carries pulsed currents, should have a low impedance

return to the reference. AGND is the ground to which most

internal ADC analog signals are referenced; it must be connected

with the least resistance to the analog ground plane. DGND

must be tied to the analog or digital ground plane depending on the

configuration. OGND is connected to the digital system ground.

The layout of the decoupling of the reference voltage is

important. To minimize parasitic inductances, place the

decoupling capacitor close to the ADC and connect it with

short, thick traces.

EVALUATING THE AD7641 PERFORMANCE

A recommended layout for the AD7641 is outlined in the

documentation of the EVAL-AD7641-CB evaluation board for

the AD7641. The evaluation board package includes a fully

assembled and tested evaluation board, documentation, and

software for controlling the board from a PC via the EVAL-

CONTROL BRD3.

AD7641

Rev. 0 | Page 28 of 28

OUTLINE DIMENSIONS

COMPLIANT TO JEDEC STANDARDS MS-026-BBC

TOP VIEW

(PINS DOWN)

0.27

0.22

0.17

0.50

BSC

LEAD PITCH

7.00

BSC SQ

1.60

MAX

0.75

0.60

0.45

VIEW A

9.00

BSC SQ

PIN 1

0.20

0.09

1.45

1.40

1.35

0.08 MAX

COPLANARITY

VIEW A

ROTATED 90° CCW

SEATING

PLANE

7°

3.5°

0°

0.15

0.05

Figure 41. 48-Lead Low Profile Quad Flat Package [LQFP]

(ST-48)

Dimensions shown in millimeters

COMPLIANT TO JEDEC STANDARDS MO-220-VKKD-2

PIN 1

INDICATOR

TOP

BSC SQ

7.00

BSC SQ

5.25

5.10 SQ

4.95

0.50

0.40

0.30

0.23

0.18

0.50 BSC

12° MAX

0.20 REF

0.80 MAX

0.65 TYP

1.00

0.85

0.80

5.50

REF

0.05 MAX

0.02 NOM

0.60 MAX

0.60 MAX PIN 1

INDICATOR

COPLANARITY

0.08

SEATING

PLANE

0.25 MIN

EXPOSED

(BOTTOM VIEW)

PADDLE CONNECTED TO AGND.

THIS CONNECTION IS NOT

REQUIRED TO MEET THE

ELECTRICAL PERFORMANCES.

Figure 42. 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]

7 mm × 7 mm Body, Very Thin Quad (CP-48-1)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD7641BCPZ −40°C to +85°C 48-Lead Lead Frame Chip Scale Package (LFCSP_VQ) CP-48-1

AD7641BCPZRL −40°C to +85°C 48-Lead Lead Frame Chip Scale Package (LFCSP_VQ) CP-48-1

AD7641BSTZ −40°C to +85°C 48-Lead Low Profile Quad Flat Package (LQFP) ST-48

AD7641BSTZRL −40°C to +85°C 48-Lead Low Profile Quad Flat Package (LQFP) ST-48

EVAL-AD7641CB Evaluation Board

EVAL-CONTROLBRD3 Controller Board

1 Z = Pb-free part.

2 This board can be used as a standalone evaluation board or in conjunction with the EVAL-CONTROL BRD3 for evaluation/demonstration purposes.

3 This board allows a PC to control and communicate with all Analog Devices, Inc. evaluation boards ending in the CB designators.

registered trademarks are the property of their respective owners.

D04761-0-1/06(0)