High-Linearity Analog
Optocouplers
Technical Data
HCNR200
HCNR201
CAUTION: It is advised that normal static precautions be taken in handling and assembly of this component to
prevent damage and/or degradation which may be induced by ESD.
3
4
1
2
V
F
–
+I
F
I
PD1
6
5
I
PD2
8
7
NC
NC
PD2 CATHODE
PD2 ANODE
LED CATHODE
LED ANODE
PD1 CATHODE
PD1 ANODE
Features
• Low Nonlinearity: 0.01%
• K3 (IPD2/IPD1) Transfer Gain
HCNR200: ±15%
HCNR201: ±5%
• Low Gain Temperature
Coefficient: -65 ppm/°C
• Wide Bandwidth – DC to
>1 MHz
• Worldwide Safety Approval
- UL 1577 Recognized
(5 kV rms/1 min Rating)
- CSA Approved
- IEC/EN/DIN EN 60747-5-2
Approved
V
IORM = 1414 V peak
(Option #050)
• Surface Mount Option
Available
(Option #300)
• 8-Pin DIP Package - 0.400"
Spacing
• Allows Flexible Circuit
Design
• Special Selection for
HCNR201: Tighter K1, K3
and Lower Nonlinearity
Available
Applications
• Low Cost Analog Isolation
• Telecom: Modem, PBX
• Industrial Process Control:
Transducer Isolator
Isolator for Thermocouples
4 mA to 20 mA Loop Isolation
• SMPS Feedback Loop, SMPS
Feedforward
• Monitor Motor Supply
Voltage
• Medical
Description
The HCNR200/201 high-linearity
analog optocoupler consists of a
high-performance AlGaAs LED
that illuminates two closely
matched photodiodes. The input
photodiode can be used to
monitor, and therefore stabilize,
the light output of the LED. As a
result, the nonlinearity and drift
characteristics of the LED can be
virtually eliminated. The output
photodiode produces a photocur-
rent that is linearly related to the
light output of the LED. The close
matching of the photodiodes and
advanced design of the package
ensure the high linearity and
stable gain characteristics of the
optocoupler.
The HCNR200/201 can be used to
isolate analog signals in a wide
variety of applications that
require good stability, linearity,
bandwidth and low cost. The
HCNR200/201 is very flexible
and, by appropriate design of the
application circuit, is capable of
operating in many different
modes, including: unipolar/
bipolar, ac/dc and inverting/non-
inverting. The HCNR200/201 is
an excellent solution for many
analog isolation problems.
Schematic
2
Ordering Information:
HCNR20x
0 = ±15% Transfer Gain, 0.25% Maximum Nonlinearity
1 = ±5% Transfer Gain, 0.05% Maximum Nonlinearity
Option yyyy
050 = IEC/EN/DIN EN 60747-5-2 V
IORM = 1414 V peak Option
300 = Gull Wing Surface Mount Lead Option
500 = Tape/Reel Package Option (1 k min.)
XXXE = Lead Free Option
Option data sheets available. Contact your Agilent Technologies sales representative or authorized distributor
for information.
Package Outline Drawings
Figure 1.
Remarks: The notation “#” is used for existing products, while (new) products launched since 15th July
2001 and lead free option will use “-”
0.40 (0.016)
0.56 (0.022)
1
2
3
4
8
7
6
5
1.70 (0.067)
1.80 (0.071)
2.54 (0.100) TYP.
0.51 (0.021) MIN.
5.10 (0.201) MAX.
3.10 (0.122)
3.90 (0.154)
DIMENSIONS IN MILLIMETERS AND (INCHES).
NC
PD1
K
1
11.30 (0.445)
MAX.
PIN
ONE 1.50
(0.059)
MAX.
A
HCNR200Z
YYWW
OPTION
CODE*
DATE
CODE
8 7 6 5
12 3 4
9.00
(0.354)
TYP.
0.20 (0.008)
0.30 (0.012)
0°
15°
11.00
(0.433)
MAX.
10.16
(0.400)
TYP.
K
2
PD2
NC
LED
* MARKING CODE LETTER FOR OPTION NUMBERS.
"V" = OPTION 050
OPTION NUMBERS 300 AND 500 NOT MARKED.
NOTE: FLOATING LEAD PROTRUSION IS 0.25 mm (10 mils) MAX.
3
Gull Wing Surface Mount Option #300
1.00 ± 0.15
(0.039 ± 0.006)
7° NOM.
12.30 ± 0.30
(0.484 ± 0.012)
0.75 ± 0.25
(0.030 ± 0.010)
11.00
(0.433)
5
6
7
8
4
3
2
1
11.15 ± 0.15
(0.442 ± 0.006)
9.00 ± 0.15
(0.354 ± 0.006)
1.3
(0.051)
13.56
(0.534)
2.29
(0.09)
LAND PATTERN RECOMMENDATION
1.78 ± 0.15
(0.070 ± 0.006)
4.00
(0.158)MAX.
1.55
(0.061)
MAX.
2.54
(0.100)
BSC
DIMENSIONS IN MILLIMETERS (INCHES).
LEAD COPLANARITY = 0.10 mm (0.004 INCHES).
NOTE: FLOATING LEAD PROTRUSION IS
0.25 mm (10 mils) MAX.
0.254 + 0.076
- 0.0051
(0.010+ 0.003)
- 0.002)
MAX.
4
Solder Reflow Temperature Profile
Regulatory Information
The HCNR200/201 optocoupler
features a 0.400" wide, eight pin
DIP package. This package was
specifically designed to meet
worldwide regulatory require-
ments. The HCNR200/201 has
been approved by the following
organizations:
UL
Recognized under UL 1577,
Component Recognition Program,
FILE E55361
CSA
Approved under CSA Component
Acceptance Notice #5, File CA
88324
IEC/EN/DIN EN 60747-5-2
Approved under
IEC 60747-5-2:1997 + A1:2002
EN 60747-5-2:2001 + A1:2002
DIN EN 60747-5-2 (VDE 0884
Teil 2):2003-01
(Option 050 only)
Recommended Pb-Free IR Profile
0
TIME (SECONDS)
TEMPERATURE (°C)
200
100
50 150100 200 250
300
0
30
SEC.
50 SEC.
30
SEC.
160°C
140°C
150°C
PEAK
TEMP.
245°C
PEAK
TEMP.
240°C
PEAK
TEMP.
230°C
SOLDERING
TIME
200°C
PREHEATING TIME
150°C, 90 + 30 SEC.
2.5°C ± 0.5°C/SEC.
3°C + 1°C/–0.5°C
TIGHT
TYPICAL
LOOSE
ROOM
TEMPERATURE
PREHEATING RATE 3°C + 1°C/–0.5°C/SEC.
REFLOW HEATING RATE 2.5°C ± 0.5°C/SEC.
217 °C
RAMP-DOWN
6 °C/SEC. MAX.
RAMP-UP
3 °C/SEC. MAX.
150 - 200 °C
260 +0/-5 °C
t 25 °C to PEAK
60 to 150 SEC.
20-40 SEC.
TIME WITHIN 5 °C of ACTUAL
PEAK TEMPERATURE
t
p
t
s
PREHEAT
60 to 180 SEC.
t
L
T
L
T
smax
T
smin
25
T
p
TIME
TEMPERATURE
NOTES:
THE TIME FROM 25 °C to PEAK TEMPERATURE = 8 MINUTES MAX.
T
smax
= 200 °C, T
smin
= 150 °C
5
IEC/EN/DIN EN 60747-5-2 Insulation Characteristics (Option #050 Only)
Description Symbol Characteristic Unit
Installation classification per DIN VDE 0110/1.89, Table 1
For rated mains voltage ≤600 V rms I-IV
For rated mains voltage ≤1000 V rms I-III
Climatic Classification (DIN IEC 68 part 1) 55/100/21
Pollution Degree (DIN VDE 0110 Part 1/1.89) 2
Maximum Working Insulation Voltage VIORM 1414 V peak
Input to Output Test Voltage, Method b* VPR 2651 V peak
VPR = 1.875 x VIORM, 100% Production Test with
tm = 1 sec, Partial Discharge < 5 pC
Input to Output Test Voltage, Method a* VPR 2121 V peak
VPR = 1.5 x VIORM, Type and sample test, tm = 60 sec,
Partial Discharge < 5 pC
Highest Allowable Overvoltage* VIOTM 8000 V peak
(Transient Overvoltage, tini = 10 sec)
Safety-Limiting Values
(Maximum values allowed in the event of a failure,
also see Figure 11)
Case Temperature TS150 °C
Current (Input Current IF, PS = 0) IS400 mA
Output Power PS,OUTPUT 700 mW
Insulation Resistance at TS, VIO = 500 V RS>109Ω
*Refer to the front of the Optocoupler section of the current catalog for a more detailed description of IEC/EN/DIN EN 60747-5-2 and
other product safety regulations.
Note: Optocouplers providing safe electrical separation per IEC/EN/DIN EN 60747-5-2 do so only within the safety-limiting values to
which they are qualified. Protective cut-out switches must be used to ensure that the safety limits are not exceeded.
Insulation and Safety Related Specifications
Parameter Symbol Value Units Conditions
Min. External Clearance L(IO1) 9.6 mm Measured from input terminals to output
(External Air Gap) terminals, shortest distance through air
Min. External Creepage L(IO2) 10.0 mm Measured from input terminals to output
(External Tracking Path) terminals, shortest distance path along body
Min. Internal Clearance 1.0 mm Through insulation distance conductor to
(Internal Plastic Gap) conductor, usually the direct distance
between the photoemitter and photodetector
inside the optocoupler cavity
Min. Internal Creepage 4.0 mm The shortest distance around the border
(Internal Tracking Path) between two different insulating materials
measured between the emitter and detector
Comparative Tracking Index CTI 200 V DIN IEC 112/VDE 0303 PART 1
Isolation Group IIIa Material group (DIN VDE 0110)
Option 300 – surface mount classification is Class A in accordance with CECC 00802.
6
Absolute Maximum Ratings
Storage Temperature .................................................. -55°C to +125°C
Operating Temperature (TA) ........................................ -55°C to +100°C
Junction Temperature (TJ) ............................................................ 125°C
Reflow Temperature Profile ... See Package Outline Drawings Section
Lead Solder Temperature ..................................................260°C for 10s
(up to seating plane)
Average Input Current - IF............................................................ 25 mA
Peak Input Current - IF................................................................. 40 mA
(50 ns maximum pulse width)
Reverse Input Voltage - VR.............................................................. 2.5 V
(IR = 100 µA, Pin 1-2)
Input Power Dissipation ......................................... 60 mW @ TA = 85°C
(Derate at 2.2 mW/°C for operating temperatures above 85°C)
Reverse Output Photodiode Voltage ................................................ 30 V
(Pin 6-5)
Reverse Input Photodiode Voltage................................................... 30 V
(Pin 3-4)
Recommended Operating Conditions
Storage Temperature .................................................... -40°C to +85°C
Operating Temperature ................................................. -40°C to +85°C
Average Input Current - IF....................................................... 1 - 20 mA
Peak Input Current - IF................................................................. 35 mA
(50% duty cycle, 1 ms pulse width)
Reverse Output Photodiode Voltage ........................................... 0 - 15 V
(Pin 6-5)
Reverse Input Photodiode Voltage .............................................. 0 - 15 V
(Pin 3-4)
7
Electrical Specifications
TA = 25°C unless otherwise specified.
Parameter Symbol Device Min. Typ. Max. Units Test Conditions Fig. Note
Transfer Gain K3HCNR200 0.85 1.00 1.15 5 nA < IPD < 50 µA, 2,3 1
0 V < VPD < 15 V
HCNR201 0.95 1.00 1.05 5 nA < IPD < 50 µA, 1,2
0 V < VPD < 15 V
HCNR201 0.93 1.00 1.07 -40°C < T
A < 85°C, 1,2
5 nA < IPD < 50 µA,
0 V < VPD < 15 V
Temperature ∆K3/∆TA -65 ppm/°C-40°C < T
A < 85°C, 2,3
Coefficient of 5 nA < IPD < 50 µA,
Transfer Gain 0 V < VPD < 15 V
DC NonLinearity NLBF HCNR200 0.01 0.25 % 5 nA < IPD < 50 µA, 4,5, 3
(Best Fit) 0 V < VPD < 15 V 6
HCNR201 0.01 0.05 5 nA < IPD < 50 µA, 2,3
0 V < VPD < 15 V
HCNR201 0.01 0.07 -40°C < TA < 85°C, 2,3
5 nA < IPD < 50 µA,
0 V < VPD < 15 V
DC Nonlinearity NLEF 0.016 5 nA < IPD < 50 µA, 4
(Ends Fit) 0 V < VPD < 15 V
Input Photo- K1HCNR200 0.25 0.50 0.75 % IF = 10 mA, 7 2
diode Current 0 V < VPD1 < 15 V
Transfer Ratio HCNR201 0.36 0.48 0.72
(IPD1/IF)
Temperature ∆K1/∆TA-0.3 %/°C-40°C < TA < 85°C, 7
Coefficient IF = 10 mA
of K10 V < VPD1 < 15 V
Photodiode ILK 0.5 25 nA IF = 0 mA, 8
Leakage Current 0 V < VPD < 15 V
Photodiode BVRPD 30 150 V IR = 100 µA
Reverse Break-
down Voltage
Photodiode CPD 22 pF VPD = 0 V
Capacitance
LED Forward VF1.3 1.6 1.85 V IF = 10 mA 9,
Voltage 10
1.2 1.6 1.95 IF = 10 mA,
-40°C < TA < 85°C
LED Reverse BVR2.5 9 V IF = 100 µA
Breakdown
Voltage
Temperature ∆V
F/∆TA-1.7 mV/°CI
F = 10 mA
Coefficient of
Forward Voltage
LED Junction CLED 80 pF f = 1 MHz,
Capacitance V
F = 0 V
8
AC Electrical Specifications
TA = 25°C unless otherwise specified.
Test
Parameter Symbol Device Min. Typ. Max. Units Conditions Fig. Note
LED Bandwidth f -3dB 9 MHz IF = 10 mA
Application Circuit Bandwidth:
High Speed 1.5 MHz 16 7
High Precision 10 kHz 17 7
Application Circuit: IMRR
High Speed 95 dB freq = 60 Hz 16 7, 8
Package Characteristics
TA = 25°C unless otherwise specified.
Test
Parameter Symbol Device Min. Typ. Max. Units Conditions Fig. Note
Input-Output VISO 5000 V rms RH ≤50%, 5, 6
Momentary-Withstand t = 1 min.
Voltage*
Resistance RI-O 1012 1013 ΩV
O = 500 VDC 5
(Input-Output)
1011 TA = 100°C, 5
VIO = 500 VDC
Capacitance CI-O 0.4 0.6 pF f = 1 MHz 5
(Input-Output)
Notes:
1. K3 is calculated from the slope of the
best fit line of IPD2 vs. IPD1 with eleven
equally distributed data points from
5 nA to 50 µA. This is approximately
equal to IPD2/IPD1 at IF = 10 mA.
2. Special selection for tighter K1, K3 and
lower Nonlinearity available.
3. BEST FIT DC NONLINEARITY (NLBF) is
the maximum deviation expressed as a
percentage of the full scale output of a
“best fit” straight line from a graph of
IPD2 vs. IPD1 with eleven equally distrib-
uted data points from 5 nA to 50 µA.
IPD2 error to best fit line is the deviation
below and above the best fit line,
expressed as a percentage of the full
scale output.
4. ENDS FIT DC NONLINEARITY (NLEF)
is the maximum deviation expressed as
a percentage of full scale output of a
straight line from the 5 nA to the 50 µA
data point on the graph of IPD2 vs. IPD1.
5. Device considered a two-terminal
device: Pins 1, 2, 3, and 4 shorted
together and pins 5, 6, 7, and 8 shorted
together.
6. In accordance with UL 1577, each
optocoupler is proof tested by applying
an insulation test voltage of ≥6000 V
rms for ≥1 second (leakage detection
current limit, II-O of 5 µA max.). This
test is performed before the 100%
production test for partial discharge
(method b) shown in the IEC/EN/DIN
EN 60747-5-2 Insulation Characteris-
tics Table (for Option #050 only).
7. Specific performance will depend on
circuit topology and components.
8. IMRR is defined as the ratio of the
signal gain (with signal applied to V
IN of
Figure 16) to the isolation mode gain
(with V
IN connected to input common
and the signal applied between the
input and output commons) at 60 Hz,
expressed in dB.
*The Input-Output Momentary Withstand Voltage is a dielectric voltage rating that should not be interpreted as an input-output
continuous voltage rating. For the continuous voltage rating refer to the VDE 0884 Insulation Characteristics Table (if applicable), your
equipment level safety specification, or Application Note 1074, “Optocoupler Input-Output Endurance Voltage.”
9
Figure 5. NLBF vs. Temperature.
Figure 2. Normalized K3 vs. Input IPD. Figure 3. K3 Drift vs. Temperature. Figure 4. IPD2 Error vs. Input IPD (See
Note 4).
Figure 6. NLBF Drift vs. Temperature. Figure 7. Input Photodiode CTR vs.
LED Input Current.
Figure 8. Typical Photodiode Leakage
vs. Temperature.
Figure 9. LED Input Current vs.
Forward Voltage.
Figure 10. LED Forward Voltage vs.
Temperature.
I
LK
– PHOTODIODE LEAKAGE – nA
10.0
4.0
0.0
T
A
– TEMPERATURE – °C
6.0
2.0
8.0
-25-55 5 35 65 95 125
V
PD
= 15 V
DELTA K3 – DRIFT OF K3 TRANSFER GAIN
0.02
-0.005
-0.02
T
A
– TEMPERATURE – °C
0.01
0.005
-0.01
-0.015 = DELTA K3 MEAN
= DELTA K3 MEAN ± 2 • STD DEV
0.0
0.015
-25-55 5 35 65 95 125
0 V < V
PD
< 15 V
DELTA NLBF – DRIFT OF BEST-FIT NL – % PTS
0.02
-0.005
-0.02
TA – TEMPERATURE – °C
0.01
0.005
-0.01
-0.015 = DELTA NLBF MEAN
= DELTA NLBF MEAN ± 2 • STD DEV
0.0
0.015
-25-55 5 35 65 95 125
0 V < VPD < 15 V
5 nA < IPD < 50 µA
NORMALIZED K1 – INPUT PHOTODIODE CTR
0.0
0.5
0.2
I
F
– LED INPUT CURRENT – mA
2.0 6.0 12.0
0.6
0.4
0.3
4.0 8.0 10.0
0.7
0.8
0.9
1.0
1.1
1.2
14.0 16.0
-55°C
25°C
-40°C
85°C
100°C
NORMALIZED TO K1 CTR
AT I
F
= 10 mA, T
A
= 25°C
0 V < V
PD1
< 15 V
V
F
– LED FORWARD VOLTAGE – V
1.5
1.2
T
A
– TEMPERATURE – °C
1.8
1.7
1.4
1.3
1.6
-25-55 5 35 65 95 125
I
F
= 10 mA
NORMALIZED K3 – TRANSFER GAIN
0.0
1.06
1.00
0.94
IPD1 – INPUT PHOTODIODE CURRENT – µA
10.0 30.0 60.0
1.04
1.02
0.98
0.96
20.0 40.0 50.0
= NORM K3 MEAN
= NORM K3 MEAN ± 2 • STD DEV
NORMALIZED TO BEST-FIT K3 AT TA = 25°C,
0 V < VPD < 15 V
0.0
0.03
0.00
-0.03
I
PD1
– INPUT PHOTODIODE CURRENT – µA
10.0 30.0 60.0
0.02
0.01
-0.01
-0.02
20.0 40.0 50.0
= ERROR MEAN
= ERROR MEAN ± 2 • STD DEV
I
PD2
ERROR FROM BEST-FIT LINE (% OF FS)
T
A
= 25 °C, 0 V < V
PD
< 15 V
NL
BF
– BEST-FIT NON-LINEARITY – %
0.015
0.00
T
A
– TEMPERATURE – °C
0.03
0.025
0.01
0.005
= NL
BF
50TH PERCENTILE
= NL
BF
90TH PERCENTILE
0.02
0.035
-25-55 5 35 65 95 125
0 V < V
PD
< 15 V
5 nA < I
PD
< 50 µA
1.20
100
0.1
0.0001
V
F
– FORWARD VOLTAGE – VOLTS
1.30 1.50
10
1
0.01
0.001
1.40 1.60
I
F
– FORWARD CURRENT – mA
T
A
= 25°C
10
Figure 12. Basic Isolation Amplifier.
I
F
LED
I
PD1
PD1
R1
V
IN
A1
+
-
I
PD2
PD2
R2
A2
-
+
V
OUT
PD1
R1
V
IN
A1
-
+
PD2 PD2
R2
A2
-
+
V
OUT
A) BASIC TOPOLOGY
B) PRACTICAL CIRCUIT
C1
R3
V
CC
LED
C2
Figure 11. Thermal Derating Curve
Dependence of Safety Limiting Value
with Case Temperature per IEC/EN/
DIN EN 60747-5-2.
-
+
V
IN
-
+
V
OUT
V
IN
-
+
-
+
V
OUT
A) POSITIVE INPUT
V
CC
B) POSITIVE OUTPUT
C) NEGATIVE INPUT D) NEGATIVE OUTPUT
Figure 13. Unipolar Circuit Topologies.
0
800
300
0
T
S
– CASE TEMPERATURE – °C
25 75 150
600
500
200
100
50 100 125
P
S
OUTPUT POWER – mV
I
S
INPUT CURRENT – mA
400
700
900
1000
175
11
Figure 15. Loop-Powered 4-20 mA Current Loop Circuits.
Figure 14. Bipolar Circuit Topologies.
-
+
-
+
VOUT
VIN
-
+
-
+
VOUT
A) SINGLE OPTOCOUPLER
VCC1
B) DUAL OPTOCOUPLER
VCC1
IOS1
VCC2
IOS2
VIN
-
+
VCC
-
+
V
OUT
+I
IN
-
+
-
+
+I
OUT
A) RECEIVER
B) TRANSMITTER
PD2
V
IN
-
+
V
CC
-I
IN
R1
R3
PD1
LED
D1
R2
R1
PD1
LED
-I
OUT
R2
R3
PD2
D1 Q1
12
Figure 18. Bipolar Isolation Amplifier.
Figure 16. High-Speed Low-Cost Analog Isolator.
V
IN
V
CC1
+5 V
R1
68 K
PD1
LED
R3
10 K
Q1
2N3906
R4
10
Q2
2N3904
V
CC2
+5 V
R2
68 K
PD2
R5
10 K
Q3
2N3906
R6
10
Q4
2N3904
R7
470
V
OUT
-
+
PD1
2
3A1
7
4
R1
200 K
INPUT
BNC 1%
C3
0.1µ
VCC1 +15 V
C1
47 P
LT1097
R6
6.8 K
R4
2.2 K
R5
270
Q1
2N3906
VEE1 -15 V
C4
0.1µ
R3
33 K
LED D1
1N4150
-
+
PD2
2
3
A2
7
4
C2
33 POUTPUT
BNC
174 K
LT1097
50 K
1 %
VEE2 -15 V
C6
0.1µ
R2
C5
0.1µ
VCC2 +15 V
6
6
Figure 17. Precision Analog Isolation Amplifier.
-
+
V
MAG
-
+
V
IN
OC1
PD1
+
-
OC2
PD1
R1
50 K
D2
C2 10 pf
C1 10 pf
D1 R4
680
R5
680
OC1
LED
OC2
LED
R3
180 K
R2
180 K
BALANCE
C3 10 pf
OC1
PD2
R6
180 K
R7
50 K
GAIN
OC2
PD2
13
-
+
V
MAG
-
+
V
IN
OC1
PD1
+
-
D4
C2 10 pf
C1 10 pf
D3
R4
680 K
OC1
LED
R1
220 K
C3 10 pf
OC1
PD2
R5
180 K
R6
50 K
GAIN
R2
10 K
R3
4.7 K
D1
-
+
D2
+
-R7
6.8 K
V
CC
R8
2.2 K
V
SIGN
OC2
6N139
Figure 20. SPICE Model Listing.
Figure 19. Magnitude/Sign Isolation Amplifier.
H
.SUBCKT HCNR200
14
Figure 21. 4 to 20 mA HCNR200 Receiver Circuit.
Figure 22. 4 to 20 mA HCNR200 Transmitter Circuit.
-
+VOUT
-
+
V
CC
5.5 V
R1
10 kΩ
+ILOOP
HCNR200
PD 1
-ILOOP
R2
10 kΩ
R4
180 Ω
2N3906
Z1
5.1 V
0.1 µF
R3
25 Ω
0.001 µF
R5
80 kΩ
LM158
HCNR200
PD 2
0.001 µF
2
HCNR200
LED
LM158
DESIGN EQUATIONS:
V
OUT
/ ILOOP = K3 (R5 R3) / R1 + R3)
K3 = K2 / K1 = CONSTANT = 1
NOTE: THE TWO OP-AMPS SHOWN ARE TWO SEPARATE LM158, AND NOT TWO CHANNELS IN A SINGLE
DUAL PACKAGE, OTHERWISE THE LOOP SIDE AND OUTPUT SIDE WILL NOT BE PROPERLY ISOLATED.
-
+
V
CC
R3
10 kΩ
+ILOOP
HCNR200
PD 2
-ILOOP
R4
10 kΩ
R6
140 Ω
2N3904
Z1
5.1 V
0.1 µF
R7
3.2 kΩ
0.001 µF
R1
80 kΩ
LM158
HCNR200
PD 1
0.001 µF
1
R8
100 kΩ
VIN
V
CC
5.5 V
R2
150 Ω
HCNR200
LED
2N3906 2N3904
2N3904
R5
25 Ω
LM158
-
+
DESIGN EQUATIONS:
(ILOOP / V
IN
) = K3 (R5 + R3) / R5 R1)
K3 = K2 / K1 = CONSTANT = 1
NOTE: THE TWO OP-AMPS SHOWN ARE TWO SEPARATE LM158, AND NOT TWO CHANNELS IN A SINGLE
DUAL PACKAGE, OTHERWISE THE LOOP SIDE AND OUTPUT SIDE WILL NOT BE PROPERLY ISOLATED.
15
Theory of Operation
Figure 1 illustrates how the
HCNR200/201 high-linearity
optocoupler is configured. The
basic optocoupler consists of an
LED and two photodiodes. The
LED and one of the photodiodes
(PD1) is on the input leadframe
and the other photodiode (PD2) is
on the output leadframe. The
package of the optocoupler is
constructed so that each photo-
diode receives approximately the
same amount of light from the
LED.
An external feedback amplifier
can be used with PD1 to monitor
the light output of the LED and
automatically adjust the LED
current to compensate for any
non-linearities or changes in light
output of the LED. The feedback
amplifier acts to stabilize and
linearize the light output of the
LED. The output photodiode then
converts the stable, linear light
output of the LED into a current,
which can then be converted back
into a voltage by another
amplifier.
Figure 12a illustrates the basic
circuit topology for implementing
a simple isolation amplifier using
the HCNR200/201 optocoupler.
Besides the optocoupler, two
external op-amps and two
resistors are required. This simple
circuit is actually a bit too simple
to function properly in an actual
circuit, but it is quite useful for
explaining how the basic isolation
amplifier circuit works (a few
more components and a circuit
change are required to make a
practical circuit, like the one
shown in Figure 12b).
The operation of the basic circuit
may not be immediately obvious
just from inspecting Figure 12a,
particularly the input part of the
circuit. Stated briefly, amplifier
A1 adjusts the LED current (IF),
and therefore the current in PD1
(IPD1), to maintain its “+” input
terminal at 0 V. For example,
increasing the input voltage would
tend to increase the voltage of the
“+” input terminal of A1 above 0
V. A1 amplifies that increase,
causing IF to increase, as well as
IPD1. Because of the way that PD1
is connected, IPD1 will pull the “+”
terminal of the op-amp back
toward ground. A1 will continue
to increase IF until its “+”
terminal is back at 0 V. Assuming
that A1 is a perfect op-amp, no
current flows into the inputs of
A1; therefore, all of the current
flowing through R1 will flow
through PD1. Since the “+” input
of A1 is at 0 V, the current
through R1, and therefore IPD1 as
well, is equal to VIN/R1.
Essentially, amplifier A1 adjusts IF
so that
IPD1 = VIN/R1.
Notice that IPD1 depends ONLY on
the input voltage and the value of
R1 and is independent of the light
output characteristics of the LED.
As the light output of the LED
changes with temperature, ampli-
fier A1 adjusts IF to compensate
and maintain a constant current
in PD1. Also notice that IPD1 is
exactly proportional to VIN, giving
a very linear relationship between
the input voltage and the
photodiode current.
The relationship between the input
optical power and the output
current of a photodiode is very
linear. Therefore, by stabilizing
and linearizing IPD1, the light
output of the LED is also
stabilized and linearized. And
since light from the LED falls on
both of the photodiodes, IPD2 will
be stabilized as well.
The physical construction of the
package determines the relative
amounts of light that fall on the
two photodiodes and, therefore,
the ratio of the photodiode
currents. This results in very
stable operation over time and
temperature. The photodiode
current ratio can be expressed as
a constant, K, where
K = IPD2/IPD1.
Amplifier A2 and resistor R2 form
a trans-resistance amplifier that
converts IPD2 back into a voltage,
V
OUT, where
V
OUT = IPD2*R2.
Combining the above three
equations yields an overall
expression relating the output
voltage to the input voltage,
V
OUT/VIN = K*(R2/R1).
Therefore the relationship
between VIN and V
OUT is constant,
linear, and independent of the
light output characteristics of the
LED. The gain of the basic isola-
tion amplifier circuit can be
adjusted simply by adjusting the
ratio of R2 to R1. The parameter
K (called K3 in the electrical
specifications) can be thought of
as the gain of the optocoupler and
is specified in the data sheet.
Remember, the circuit in
Figure 12a is simplified in order
to explain the basic circuit opera-
tion. A practical circuit, more like
Figure 12b, will require a few
additional components to stabilize
the input part of the circuit, to
limit the LED current, or to
16
second circuit requires two
optocouplers, separate gain
adjustments for the positive and
negative portions of the signal,
and can exhibit crossover distor-
tion near zero volts. The correct
circuit to choose for an applica-
tion would depend on the
requirements of that particular
application. As with the basic
isolation amplifier circuit in
Figure 12a, the circuits in Figure
14 are simplified and would
require a few additional compo-
nents to function properly. Two
example circuits that operate with
bipolar input signals are
discussed in the next section.
As a final example of circuit
design flexibility, the simplified
schematics in Figure 15 illustrate
how to implement 4-20 mA
analog current-loop transmitter
and receiver circuits using the
HCNR200/201 optocoupler. An
important feature of these circuits
is that the loop side of the circuit
is powered entirely by the loop
current, eliminating the need for
an isolated power supply.
The input and output circuits in
Figure 15a are the same as the
negative input and positive output
circuits shown in Figures 13c and
13b, except for the addition of R3
and zener diode D1 on the input
side of the circuit. D1 regulates
the supply voltage for the input
amplifier, while R3 forms a
current divider with R1 to scale
the loop current down from 20
mA to an appropriate level for the
input circuit (<50 µA).
As in the simpler circuits, the
input amplifier adjusts the LED
current so that both of its input
terminals are at the same voltage.
The loop current is then divided
optimize circuit performance.
Example application circuits will
be discussed later in the data
sheet.
Circuit Design Flexibility
Circuit design with the HCNR200/
201 is very flexible because the
LED and both photodiodes are
accessible to the designer. This
allows the designer to make perf-
ormance trade-offs that would
otherwise be difficult to make with
commercially available isolation
amplifiers (e.g., bandwidth vs.
accuracy vs. cost). Analog isola-
tion circuits can be designed for
applications that have either
unipolar (e.g., 0-10 V) or bipolar
(e.g., ±10 V) signals, with
positive or negative input or
output voltages. Several simplified
circuit topologies illustrating the
design flexibility of the HCNR200/
201 are discussed below.
The circuit in Figure 12a is
configured to be non-inverting
with positive input and output
voltages. By simply changing the
polarity of one or both of the
photodiodes, the LED, or the op-
amp inputs, it is possible to
implement other circuit configu-
rations as well. Figure 13
illustrates how to change the
basic circuit to accommodate
both positive and negative input
and output voltages. The input
and output circuits can be
matched to achieve any combina-
tion of positive and negative
voltages, allowing for both
inverting and non-inverting
circuits.
All of the configurations described
above are unipolar (single polar-
ity); the circuits cannot accommo-
date a signal that might swing
both positive and negative. It is
possible, however, to use the
HCNR200/201 optocoupler to
implement a bipolar isolation
amplifier. Two topologies that
allow for bipolar operation are
shown in Figure 14.
The circuit in Figure 14a uses two
current sources to offset the
signal so that it appears to be
unipolar to the optocoupler.
Current source IOS1 provides
enough offset to ensure that IPD1
is always positive. The second
current source, IOS2, provides an
offset of opposite polarity to
obtain a net circuit offset of zero.
Current sources IOS1 and IOS2 can
be implemented simply as
resistors connected to suitable
voltage sources.
The circuit in Figure 14b uses two
optocouplers to obtain bipolar
operation. The first optocoupler
handles the positive voltage
excursions, while the second
optocoupler handles the negative
ones. The output photodiodes are
connected in an antiparallel
configuration so that they
produce output signals of
opposite polarity.
The first circuit has the obvious
advantage of requiring only one
optocoupler; however, the offset
performance of the circuit is
dependent on the matching of IOS1
and IOS2 and is also dependent on
the gain of the optocoupler.
Changes in the gain of the opto-
coupler will directly affect the
offset of the circuit.
The offset performance of the
second circuit, on the other hand,
is much more stable; it is inde-
pendent of optocoupler gain and
has no matched current sources
to worry about. However, the
17
between R1 and R3. IPD1 is equal
to the current in R1 and is given
by the following equation:
IPD1 = ILOOP*R3/(R1+R3).
Combining the above equation
with the equations used for Figure
12a yields an overall expression
relating the output voltage to the
loop current,
V
OUT/ILOOP
= K*(R2*R3)/(R1+R3).
Again, you can see that the
relationship is constant, linear,
and independent of the charac-
teristics of the LED.
The 4-20 mA transmitter circuit in
Figure 15b is a little different
from the previous circuits, partic-
ularly the output circuit. The
output circuit does not directly
generate an output voltage which
is sensed by R2, it instead uses
Q1 to generate an output current
which flows through R3. This
output current generates a
voltage across R3, which is then
sensed by R2. An analysis similar
to the one above yields the
following expression relating
output current to input voltage:
ILOOP/VIN = K*(R2+R3)/(R1*R3).
The preceding circuits were pre-
sented to illustrate the flexibility
in designing analog isolation
circuits using the HCNR200/201.
The next section presents several
complete schematics to illustrate
practical applications of the
HCNR200/201.
Example Application
Circuits
The circuit shown in Figure 16 is
a high-speed low-cost circuit
designed for use in the feedback
path of switch-mode power
supplies. This application requires
good bandwidth, low cost and
stable gain, but does not require
very high accuracy. This circuit is
a good example of how a designer
can trade off accuracy to achieve
improvements in bandwidth and
cost. The circuit has a bandwidth
of about 1.5 MHz with stable gain
characteristics and requires few
external components.
Although it may not appear so at
first glance, the circuit in Figure
16 is essentially the same as the
circuit in Figure 12a. Amplifier A1
is comprised of Q1, Q2, R3 and
R4, while amplifier A2 is
comprised of Q3, Q4, R5, R6 and
R7. The circuit operates in the
same manner as well; the only
difference is the performance of
amplifiers A1 and A2. The lower
gains, higher input currents and
higher offset voltages affect the
accuracy of the circuit, but not
the way it operates. Because the
basic circuit operation has not
changed, the circuit still has good
gain stability. The use of discrete
transistors instead of op-amps
allowed the design to trade off
accuracy to achieve good
bandwidth and gain stability at
low cost.
To get into a little more detail
about the circuit, R1 is selected to
achieve an LED current of about
7-10 mA at the nominal input
operating voltage according to the
following equation:
IF = (VIN/R1)/K1,
where K1 (i.e., IPD1/IF) of the
optocoupler is typically about
0.5%. R2 is then selected to
achieve the desired output voltage
according to the equation,
V
OUT/VIN = R2/R1.
The purpose of R4 and R6 is to
improve the dynamic response
(i.e., stability) of the input and
output circuits by lowering the
local loop gains. R3 and R5 are
selected to provide enough
current to drive the bases of Q2
and Q4. And R7 is selected so that
Q4 operates at about the same
collector current as Q2.
The next circuit, shown in
Figure 17, is designed to achieve
the highest possible accuracy at a
reasonable cost. The high
accuracy and wide dynamic range
of the circuit is achieved by using
low-cost precision op-amps with
very low input bias currents and
offset voltages and is limited by
the performance of the opto-
coupler. The circuit is designed to
operate with input and output
voltages from 1 mV to 10 V.
The circuit operates in the same
way as the others. The only major
differences are the two compensa-
tion capacitors and additional
LED drive circuitry. In the high-
speed circuit discussed above, the
input and output circuits are
stabilized by reducing the local
loop gains of the input and output
circuits. Because reducing the
loop gains would decrease the
accuracy of the circuit, two
compensation capacitors, C1 and
C2, are instead used to improve
circuit stability. These capacitors
also limit the bandwidth of the
circuit to about 10 kHz and can
be used to reduce the output
noise of the circuit by reducing its
bandwidth even further.
The additional LED drive circuitry
(Q1 and R3 through R6) helps to
maintain the accuracy and band-
width of the circuit over the entire
range of input voltages. Without
these components, the transcon-
ductance of the LED driver would
decrease at low input voltages
and LED currents. This would
reduce the loop gain of the input
circuit, reducing circuit accuracy
and bandwidth. D1 prevents
excessive reverse voltage from
being applied to the LED when
the LED turns off completely.
No offset adjustment of the circuit
is necessary; the gain can be
adjusted to unity by simply
adjusting the 50 kohm poten-
tiometer that is part of R2. Any
OP-97 type of op-amp can be
used in the circuit, such as the
LT1097 from Linear Technology
or the AD705 from Analog
Devices, both of which offer pA
bias currents, µV offset voltages
and are low cost. The input
terminals of the op-amps and the
photodiodes are connected in the
circuit using Kelvin connections
to help ensure the accuracy of the
circuit.
The next two circuits illustrate
how the HCNR200/201 can be
used with bipolar input signals.
The isolation amplifier in
Figure 18 is a practical implemen-
tation of the circuit shown in
Figure 14b. It uses two opto-
couplers, OC1 and OC2; OC1
handles the positive portions of
the input signal and OC2 handles
the negative portions.
Diodes D1 and D2 help reduce
crossover distortion by keeping
both amplifiers active during both
positive and negative portions of
the input signal. For example,
when the input signal positive,
optocoupler OC1 is active while
OC2 is turned off. However, the
amplifier controlling OC2 is kept
active by D2, allowing it to turn
on OC2 more rapidly when the
input signal goes negative,
thereby reducing crossover
distortion.
Balance control R1 adjusts the
relative gain for the positive and
negative portions of the input
signal, gain control R7 adjusts the
overall gain of the isolation
amplifier, and capacitors C1-C3
provide compensation to stabilize
the amplifiers.
The final circuit shown in
Figure 19 isolates a bipolar
analog signal using only one
optocoupler and generates two
output signals: an analog signal
proportional to the magnitude of
the input signal and a digital
signal corresponding to the sign
of the input signal. This circuit is
especially useful for applications
where the output of the circuit is
going to be applied to an analog-
to-digital converter. The primary
advantages of this circuit are very
good linearity and offset, with
only a single gain adjustment and
no offset or balance adjustments.
To achieve very high linearity for
bipolar signals, the gain should be
exactly the same for both positive
and negative input polarities. This
circuit achieves excellent linearity
by using a single optocoupler and
a single input resistor, which
guarantees identical gain for both
positive and negative polarities of
the input signal. This precise
matching of gain for both polari-
ties is much more difficult to
obtain when separate components
are used for the different input
polarities, such as is the previous
circuit.
The circuit in Figure 19 is actually
very similar to the previous
circuit. As mentioned above, only
one optocoupler is used. Because
a photodiode can conduct current
in only one direction, two diodes
(D1 and D2) are used to steer the
input current to the appropriate
terminal of input photodiode PD1
to allow bipolar input currents.
Normally the forward voltage
drops of the diodes would cause a
serious linearity or accuracy
problem. However, an additional
amplifier is used to provide an
appropriate offset voltage to the
other amplifiers that exactly
cancels the diode voltage drops to
maintain circuit accuracy.
Diodes D3 and D4 perform two
different functions; the diodes
keep their respective amplifiers
active independent of the input
signal polarity (as in the previous
circuit), and they also provide the
feedback signal to PD1 that
cancels the voltage drops of
diodes D1 and D2.
Either a comparator or an extra
op-amp can be used to sense the
polarity of the input signal and
drive an inexpensive digital
optocoupler, like a 6N139.
It is also possible to convert this
circuit into a fully bipolar circuit
(with a bipolar output signal) by
using the output of the 6N139 to
drive some CMOS switches to
switch the polarity of PD2
depending on the polarity of the
input signal, obtaining a bipolar
output voltage swing.
HCNR200/201 SPICE
Model
Figure 20 is the net list of a
SPICE macro-model for the
HCNR200/201 high-linearity
optocoupler. The macro-model
accurately reflects the primary
characteristics of the HCNR200/
201 and should facilitate the
design and understanding of
circuits using the HCNR200/201
optocoupler.
18
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Copyright © 2005 Agilent Technologies, Inc.
Obsoletes 5989-0286EN
March 1, 2005
5989-2137EN
