May 2011 Doc ID 018774 Rev 1 1/46
46
A5974D
Up to 2.5 A step down switching regulator
for automotive applications
Features
■Qualified following the AEC-Q100
requirements (see PPAP for more details)
■2.5 A DC output current
■Operating input voltage from 4 V to 36 V
■3.3 V / (±2%) reference voltage
■Output voltage adjustable from 1.235 V to 35 V
■Low dropout operation: 100% duty cycle
■250 kHz internally fixed frequency
■Voltage feedforward
■Zero load current operation
■Internal current limiting
■Inhibit for zero current consumption
■Synchronization
■Protection against feedback disconnection
■Thermal shutdown
Application
■Dedicated to automotive applications
Description
The A5974D is a step down monolithic power
switching regulator with a minimum switch current
limit of 3.1 A so it is able to deliver up to 2.5 A DC
current to the load depending on the application
conditions. The output voltage can be set from
1.235 V to 35 V. The high current level is also
achieved thanks to an HSOP8 package with
exposed frame, that allows to reduce the Rth(JA)
down to approximately 40 °C/W. The device uses
an internal p-channel DMOS transistor (with a
typical RDS(on) of 250 mΩ) as switching element
to minimize the size of the external components.
An internal oscillator fixes the switching frequency
at 250 kHz. Having a minimum input voltage of 4
V only it fits the automotive applications requiring
the device operation even in cold crank
conditions. Pulse by pulse current limit with the
internal frequency modulation offers an effective
constant current short circuit protection.
HSOP8 - exposed pad
Figure 1. Application schematic
small signal
power plane
=4V TO 36V
TP3
GND
TP3
GND
D1
STPS3L40U
D1
STPS3L40U
VOUT 1
VCC
8
VREF
6
EX-PAD
9
INH
3
COMP
4
GND
7
FB 5
SYNC
2
U1
A5974D
U1
A5974D
C9
330uF
C9
330uF
TP2
VOUT
TP2
VOUT
C3
100p
C3
100p
TP4
GND
TP4
GND
L1
15uH
L1
15uH
C4
33nF
C4
33nF
R3
10K
R3
10K
R1
5.6K
R1
5.6K
C2
68nF
50V
C2
68nF
50V
C1
10uF
50V
C1
10uF
50V
TP1
VIN
TP1
VIN
R2
3.3K
R2
3.3K
www.st.com
Contents A5974D
2/46 Doc ID 018774 Rev 1
Contents
1 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.1 Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
2 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.1 Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
3 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
4 Datasheet parameters over the temperature range . . . . . . . . . . . . . . . . 8
5 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
5.1 Power supply and voltage reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
5.2 Voltages monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5.3 Oscillator and synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5.4 Current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
5.5 Error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
5.6 PWM comparator and power stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
5.7 Inhibit function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
5.8 Thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
6 Additional features and protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
6.1 Feedback disconnection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
6.2 Output overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
6.3 Zero load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
7 Closing the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
7.1 Error amplifier and compensation network . . . . . . . . . . . . . . . . . . . . . . . . 16
7.2 LC filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
7.3 PWM comparator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
8 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
A5974D Contents
Doc ID 018774 Rev 1 3/46
8.1 Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
8.2 Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
8.3 Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.3.1 Thermal resistance RthJA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
8.3.2 Thermal impedance ZTHJ-A(t) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
8.4 R.M.S. current of the embedded power MOSFET . . . . . . . . . . . . . . . . . . 29
8.5 Short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
8.6 Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
8.7 Positive buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
8.8 Negative buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
8.9 Floating boost current generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
8.10 Synchronization example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
8.11 Compensation network with MLCC at the output . . . . . . . . . . . . . . . . . . . 38
8.12 External SOFT_START network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
9 Typical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
10 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
11 Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
12 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
Pin settings A5974D
4/46 Doc ID 018774 Rev 1
1 Pin settings
1.1 Pin connection
Figure 2. Pin connection (top view)
1.2 Pin description
/54
39.#
).(
#/-0
6##
62%&
'.$
&"
Table 1. Pin description
N Pin Description
1 OUT Regulator output.
2 SYNCH Master/slave synchronization.
3INH
A logical signal (active high) disables the device. If INH not used the pin
must be grounded. When it is open an internal pull-up disable the
device.
4 COMP E/A output for frequency compensation.
5FB
Feedback input. Connecting directly to this pin results in an output
voltage of 1.23 V. An external resistive divider is required for higher
output voltages.
6 VREF 3.3 V VREF
. No cap is requested for stability.
7 GND Ground.
8 VCC Unregulated DC input voltage.
A5974D Electrical data
Doc ID 018774 Rev 1 5/46
2 Electrical data
2.1 Maximum ratings
2.2 Thermal data
Table 2. Absolute maximum ratings
Symbol Parameter Value Unit
V8Input voltage 40 V
V1
OUT pin DC voltage
OUT pin peak voltage at Δt = 0.1 μs
-1 to 40
-5 to 40
V
V
I1Maximum output current int. limit.
V4, V5Analog pins 4 V
V3INH -0.3 to VCC V
V2SYNCH -0.3 to 4 V
PTOT Power dissipation at TA ≤ 60 °C 2.25 W
TJOperating junction temperature range -40 to 150 °C
TSTG Storage temperature range -55 to 150 °C
Table 3. Thermal data
Symbol Parameter Value Unit
RthJA Maximum thermal resistance junction-ambient 40 (1)
1. Package mounted on evaluation board
°C/W
Electrical characteristics A5974D
6/46 Doc ID 018774 Rev 1
3 Electrical characteristics
TJ = -40 °C to 125 °C, VCC = 12 V, unless otherwise specified.
Table 4. Electrical characteristics
Symbol Parameter Test condition Min Typ Max Unit
VCC
Operating input
voltage range V0 = 1.235 V; I0 = 2 A 4 36 V
RDS(on)
MOSFET on
resistance 0.250 0.5 Ω
IL
Maximum limiting
current VCC = 5 V 3.1 3.6 4.1 A
fSW Switching frequency 212 250 280 kHz
Duty cycle 0 100 %
Dynamic characteristics (see test circuit)
V5Voltage feedback 4.4 V < VCC < 36 V,
20 mA < I0 < 2 A 1.198 1.235 1.272 V
DC characteristics
Iqop
Total operating
quiescent current 35mA
IqQuiescent current Duty cycle = 0; VFB = 1.5 V 2.5 mA
Iqst-by
Total stand-by
quiescent current
Vinh > 2.2 V 50 100 μA
VC C = 36 V;
Vinh > 2.2 V 50 100 μA
Inhibit
INH threshold voltage Device ON 0.8 V
Device OFF 2.2 V
Error amplifier
VOH
High level output
voltage VFB = 1 V 3.5 V
VOL
Low level output
voltage VFB = 1.5 V 0.4 V
Io source Source output current VCOMP = 1.9 V; VFB = 1 V 190 300 μA
Io sink Sink output current VCOMP = 1.9 V; VFB = 1.5 V 1 1.5 mA
IbSource bias current 2.5 4 μA
DC open loop gain RL = ∞50 65 dB
gm Transconductance ICOMP = -0.1 mA to 0.1 mA;
VCOMP = 1.9 V 2.3 mS
A5974D Electrical characteristics
Doc ID 018774 Rev 1 7/46
Synch function
High input voltage VCC = 4.4 to 36 V; 2.5 VREF V
Low input voltage VCC = 4.4 to 36 V; 0.74 V
Slave synch current(1) Vsynch = 0.74 V
Vsynch = 2.33 V
0.11
0.21
0.25
0.45 mA
Master output
amplitude Isource = 3 mA 2.75 3 V
Output pulse width no load, Vsynch = 1.65 V 0.20 0.35 μs
Reference section
Reference voltage IREF = 0 to 5 mA
VCC = 4.4 V to 36 V 3.2 3.3 3.399 V
Line regulation IREF = 0 mA
VCC = 4.4 V to 36 V 510mV
Load regulation IREF = 0 mA 8 15 mV
Short circuit current 5 18 35 mA
1. Guaranteed by design
Table 4. Electrical characteristics (continued)
Symbol Parameter Test condition Min Typ Max Unit
Datasheet parameters over the temperature range A5974D
8/46 Doc ID 018774 Rev 1
4 Datasheet parameters over the temperature range
The 100% of the population in the production flow is tested at three different ambient
temperatures (-40 °C; +25 °C, +125 °C) to guarantee the datasheet parameters inside the
junction temperature range (-40 °C; +125 °C).
The device operation is so guaranteed when the junction temperature is inside the (-40 °C;
+150 °C) temperature range. The designer can estimate the silicon temperature increase
respect to the ambient temperature evaluating the internal power losses generated during
the device operation (please refer to the Chapter 2.2).
However the embedded thermal protection disables the switching activity to protect the
device in case the junction temperature reaches the TSHTDWN (+150 °C±10 °C)
temperature.
All the datasheet parameters can be guaranteed to a maximum junction temperature of
+125 °C to avoid triggering the thermal shutdown protection during the testing phase
because of self heating.
A5974D Functional description
Doc ID 018774 Rev 1 9/46
5 Functional description
The main internal blocks are shown in the device block diagram in Figure 3. They are:
●A voltage regulator supplying the internal circuitry. From this regulator, a 3.3 V
reference voltage is externally available.
●A voltage monitor circuit which checks the input and the internal voltages.
●A fully integrated sawtooth oscillator with a frequency of 250 kHz ± 15%, including also
the voltage feed forward function and an input/output synchronization pin.
●Two embedded current limitation circuits which control the current that flows through
the power switch. The pulse-by-pulse current limit forces the power switch OFF cycle
by cycle if the current reaches an internal threshold, while the frequency shifter reduces
the switching frequency in order to significantly reduce the duty cycle.
●A transconductance error amplifier.
●A pulse width modulator (PWM) comparator and the relative logic circuitry necessary to
drive the internal power.
●A high side driver for the internal P-MOS switch.
●An inhibit block for stand-by operation.
●A circuit to implement the thermal protection function.
Figure 3. Block diagram
5.1 Power supply and voltage reference
The internal regulator circuit (shown in Figure 4) consists of a start-up circuit, an internal
voltage pre-regulator, the Bandgap voltage reference and the Bias block that provides
current to all the blocks. The Starter supplies the start-up currents to the entire device when
the input voltage goes high and the device is enabled (inhibit pin connected to ground). The
pre-regulator block supplies the Bandgap cell with a pre-regulated voltage VREG that has a
very low supply voltage noise sensitivity.
).()")4
6/,4!'%3
-/.)4/2
0%!+4/0%!+
#522%.4,)-)4
4(%2-!,
3(54$/7.
%! 07-
6
/3#),,!4/2
$
#K
1
&2%15%.#9
3()&4%2
42)--).'
3500,9
6 6
$2)6%2
6
2%&
"5&&%2
,0$-/3
0/7%2
&"
39.#
#/-0
).(
62%&
45/$.'
6##
!-V
Functional description A5974D
10/46 Doc ID 018774 Rev 1
5.2 Voltages monitor
An internal block continuously senses the Vcc, Vref and Vbg. If the voltages go higher than
their thresholds, the regulator begins operating. There is also a hysteresis on the VCC
(UVLO).
Figure 4. Internal circuit
5.3 Oscillator and synchronization
Figure 5 shows the block diagram of the oscillator circuit.
The clock generator provides the switching frequency of the device, which is internally fixed
at 250 kHz. The frequency shifter block acts to reduce the switching frequency in case of
strong overcurrent or short circuit. The clock signal is then used in the internal logic circuitry
and is the input of the ramp generator and synchronizer blocks.
The ramp generator circuit provides the sawtooth signal, used for PWM control and the
internal voltage feed-forward, while the synchronizer circuit generates the synchronization
signal. The device also has a synchronization pin which can work both as master and slave.
Beating frequency noise is an issue when more than one voltage rail is on the same board.
A simple way to avoid this issue is to operate all the regulators at the same switching
frequency.
The synchronization feature of a set of the A5974D is simply get connecting together their
SYNCH pin. The device with highest switching frequency will be the MASTER and it
provides the synchronization signal to the others. Therefore the SYNCH is a I/O pin to
deliver or recognize a frequency signal. The synchronization circuitry is powered by the
internal reference (VREF) so a small filtering capacitor (≥ 100 nF) connected between VREF
pin and the signal ground of the Master device is suggested for its proper operation.
However when a set of synchronized devices populates a board it is not possible to know in
advance the one working as Master, so the filtering capacitor have to be designed for whole
set of devices.
When one or more devices are synchronized to an external signal, its amplitude have to be
in comply with specifications given in the Ta b l e 4 . The frequency of the synchronization
signal must be, at a minimum, higher than the maximum guaranteed natural switching
frequency of the device (275 kHz, see Ta b le 4 ) while the duty cycle of the synchronization
signal can vary from approximately 10% to 90%. The small capacitor under VREF pin is
required for this operation.
34!24%2
)#")!3
02%2%'5,!4/2
"!.$'!0
62%'
62%&
6##
A5974D Functional description
Doc ID 018774 Rev 1 11/46
Figure 5. Oscillator circuit block diagram
Figure 6. Synchronization example
5.4 Current protection
The A5974D features two types of current limit protection: pulse-by-pulse and frequency
foldback.
The schematic of the current limitation circuitry for the pulse-by-pulse protection is shown in
Figure 7. The output power PDMOS transistor is split into two parallel PDMOS transistors.
The smallest one includes a resistor in series, RSENSE. The current is sensed through
RSENSE and if it reaches the threshold, the mirror becomes unbalanced and the PDMOS is
switched off until the next falling edge of the internal clock pulse. Due to this reduction of the
ON time, the output voltage decreases. Since the minimum switch ON time necessary to
sense the current in order to avoid a false overcurrent signal is too short to obtain a
sufficiently low duty cycle at 250 kHz (see Chapter 8.5), the output current in strong
overcurrent or short circuit conditions could be not properly limited. For this reason the
switching frequency is also reduced, thus keeping the inductor current under its maximum
&2%15%.#9
3()&4%2
#,/#+
'%.%2!4/2
2!-0
'%.%2!4/2
39.#(2/.):!4/2
#,/#+
2!-0
)BIAS?OSC
39.#
T
&2%15%.#9
3()&4%2
#,/#+
'%.%2!4/2
2!-0
'%.%2!4/2
39.#(2/.):%2
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH OUT
GND
COMP
FB
SS/INH
A5973D
SYNCH
A5974D A5974D
A5974D A5974D
Functional description A5974D
12/46 Doc ID 018774 Rev 1
threshold. The frequency shifter (Figure 5) functions based on the feedback voltage. As the
feedback voltage decreases (due to the reduced duty cycle), the switching frequency
decreases also.
Figure 7. Current limitation circuitry
5.5 Error amplifier
The voltage error amplifier is the core of the loop regulation. It is a transconductance
operational amplifier whose non inverting input is connected to the internal voltage
reference (1.235 V), while the inverting input (FB) is connected to the external divider or
directly to the output voltage. The output (COMP) is connected to the external compensation
network. The uncompensated error amplifier has the following characteristics:
The error amplifier output is compared to the oscillator sawtooth to perform PWM control.
5.6 PWM comparator and power stage
This block compares the oscillator sawtooth and the error amplifier output signals to
generate the PWM signal for the driving stage.
The power stage is a highly critical block, as it functions to guarantee a correct turn ON and
turn OFF of the PDMOS. The turn ON of the power element, or more accurately, the rise
time of the current at turn ON, is a very critical parameter. At a first approach, it appears that
the faster the rise time, the lower the turn ON losses.
However, there is a limit introduced by the recovery time of the recirculation diode.
DRIVER
NOT
A1
PWM
VCC
OUT
A1/A2=95
I
L
RSENSE
AM00008v1
I
OFF
II
RTH
A2
Table 5. Uncompensated error amplifier characteristics
Description Values
Transconductance 2300 µS
Low frequency gain 65 dB
Minimum sink/source voltage 1500 µA/300 µA
Output voltage swing 0.4 V/3.65 V
Input bias current 2.5 µA
A5974D Functional description
Doc ID 018774 Rev 1 13/46
In fact, when the current of the power element is equal to the inductor current, the diode
turns OFF and the drain of the power is able to go high. But during its recovery time, the
diode can be considered a high value capacitor and this produces a very high peak current,
responsible for numerous problems:
●Spikes on the device supply voltage that cause oscillations (and thus noise) due to the
board parasites.
●Turn ON overcurrent leads to a decrease in the efficiency and system reliability.
●Major EMI problems.
●Shorter freewheeling diode life.
The fall time of the current during turn OFF is also critical, as it produces voltage spikes (due
to the parasites elements of the board) that increase the voltage drop across the PDMOS.
In order to minimize these problems, a new driving circuit topology has been used and the
block diagram is shown in Figure 8. The basic idea is to change the current levels used to
turn the power switch ON and OFF, based on the PDMOS and the gate clamp status.
This circuitry allows the power switch to be turned OFF and ON quickly and addresses the
freewheeling diode recovery time problem. The gate clamp is necessary to ensure that VGS
of the internal switch does not go higher than VGSmax. The ON/OFF Control block protects
against any cross conduction between the supply line and ground.
Figure 8. Driving circuitry
5.7 Inhibit function
The inhibit feature is used to put the device in standby mode. With the INH pin higher than
2.2 V the device is disabled and the power consumption is reduced to less than 100 µA.
With the INH pin lower than 0.8 V, the device is enabled. If the INH pin is left floating, an
internal pull up ensures that the voltage at the pin reaches the inhibit threshold and the
device is disabled. The pin is also Vcc compatible.
Vgsmax
GATE
STOP
DRIVE
DRAIN
OFF
ON
PDMOS
VOUT
DRAIN
VCC
ILOAD
C
ESR
AM00009v1
IOFF
ION
ON/OFF
CONTROL
CLAMP
L
Functional description A5974D
14/46 Doc ID 018774 Rev 1
5.8 Thermal shutdown
The shutdown block generates a signal that turns OFF the power stage if the temperature of
the chip goes higher than a fixed internal threshold (150±10 °C). The sensing element of the
chip is very close to the PDMOS area, ensuring fast and accurate temperature detection. A
hysteresis of approximately 20 °C keeps the device from turning ON and OFF continuously.
A5974D Additional features and protection
Doc ID 018774 Rev 1 15/46
6 Additional features and protection
6.1 Feedback disconnection
If the feedback is disconnected, the duty cycle increases towards the maximum allowed
value, bringing the output voltage close to the input supply. This condition could destroy the
load.
To avoid this hazardous condition, the device is turned OFF if the feedback pin is left
floating.
6.2 Output overvoltage protection
Overvoltage protection, or OVP, is achieved by using an internal comparator connected to
the feedback, which turns OFF the power stage when the OVP threshold is reached. This
threshold is typically 30% higher than the feedback voltage.
When a voltage divider is required to adjust the output voltage (Figure 19), the OVP
intervention will be set at:
Equation 1
Where R1 is the resistor connected between the output voltage and the feedback pin, and
R2 is between the feedback pin and ground.
6.3 Zero load
Due to the fact that the internal power is a PDMOS, no boostrap capacitor is required and so
the device works properly even with no load at the output. In this case it works in burst
mode, with a random burst repetition rate.
VOVP 1.3 R1R2
+
R2
--------------------
•VFB
•=
Closing the loop A5974D
16/46 Doc ID 018774 Rev 1
7 Closing the loop
Figure 9. Block diagram of the loop
7.1 Error amplifier and compensation network
The output L-C filter of a step-down converter contributes with 180° degrees phase shift in
the control loop. For this reason a compensation network between the COMP pin and
GROUND is added. The simplest compensation network together with the equivalent circuit
of the error amplifier are shown in Figure 10. RC and CC introduce a pole and a zero in the
open loop gain. CP does not significantly affect system stability but it is useful to reduce the
noise of the COMP pin.
The transfer function of the error amplifier and its compensation network is:
Equation 2
Where Avo = Gm · Ro
9&&
H[WHUQDOUHVLVWRUGLYLGH
U
/&ILOWHU
LQWHUQDOVZLWFK
/5
6(16(
&
287
)%
FRPSHQVDWLRQQHWZRUN
3:0FRPSDUDWRU
5
9
5()
5
5
HUURU
DPSOLILHU
&
3
5
&
&
&
A0s() AV0 1s+RcCc
••()•
s2R0
•C0Cp
+()RcCcsR
0Cc
•R0C0Cp
+()RcCc
•+•+()1+•+•••
---------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------=
A5974D Closing the loop
Doc ID 018774 Rev 1 17/46
Figure 10. Error amplifier equivalent circuit and compensation network
The poles of this transfer function are (if Cc >> C0+CP):
Equation 3
Equation 4
whereas the zero is defined as:
Equation 5
FP1 is the low frequency which sets the bandwidth, while the zero FZ1 is usually put near to
the frequency of the double pole of the L-C filter (see below). FP2 is usually at a very high
frequency.
+
-
CP
RC
CC
FB
COMP
ΔVR0
0.8MΩ
GmΔV
V+
E/A
RC
CC
CP
C0
10pF
FP1
1
2π• R0
•Cc
•
-------------------------------------=
FP2
1
2π• Rc
•C0Cp
+()•
--------------------------------------------------------=
FZ1
1
2π• Rc
•Cc
•
-------------------------------------=
Closing the loop A5974D
18/46 Doc ID 018774 Rev 1
7.2 LC filter
The transfer function of the L-C filter is given by:
Equation 6
where RLOAD is defined as the ratio between VOUT and IOUT
.
If RLOAD>>ESR, the previous expression of ALC can be simplified and becomes:
Equation 7
The zero of this transfer function is given by:
Equation 8
F0 is the zero introduced by the ESR of the output capacitor and it is very important to
increase the phase margin of the loop.
The poles of the transfer function can be calculated through the following expression:
Equation 9
In the denominator of ALC the typical second order system equation can be recognized:
Equation 10
If the damping coefficient δ is very close to zero, the roots of the equation become a double
root whose value is ωn.
Similarly for ALC the poles can usually be defined as a double pole whose value is:
Equation 11
7.3 PWM comparator
The PWM gain is given by the following formula:
ALC s() RLOAD 1 ESR COUT s••+()•
s2LC
OUT ESR RLOAD
+()s ESR COUT
•RLOAD L+•()RLOAD
+•+•••
----------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------=
ALC s() 1 ESR COUT
•s•+
LC
OUT
•s2ESR COUT
•s1+•+•
----------------------------------------------------------------------------------------------=
FO
1
2π• ESR•COUT
•
----------------------------------------------------=
FPLC1 2,
ESR COUT ESR COUT
•()
24L•COUT
•–±•–
2L•COUT
•
------------------------------------------------------------------------------------------------------------------------------------------=
s22δ•ω
n
•sω2n
+•+
FPLC
1
2π• LC
OUT
••
----------------------------------------------=
A5974D Closing the loop
Doc ID 018774 Rev 1 19/46
Equation 12
where VOSCMAX is the maximum value of a sawtooth waveform and VOSCMIN is the
minimum value. A voltage feed forward is implemented to ensure a constant GPWM. This is
obtained by generating a sawtooth waveform directly proportional to the input voltage VCC.
Equation 13
Where K is equal to 0.076. Therefore the PWM gain is also equal to:
Equation 14
This means that even if the input voltage changes, the error amplifier does not change its
value to keep the loop in regulation, thus ensuring a better line regulation and line transient
response.
In summary, the open loop gain can be expressed as:
Equation 15
Example:
Considering RC = 10 kΩ, CC = 33 nF and CP = 100 pF, the poles and zeroes of A0 are:
FP1 = 6 Hz
FP2 = 150 kHz
FZ1 = 480 Hz
If L = 15 µH, DCR =56 mΩ, COUT = 330 µF and ESR = 25 mΩ, the poles and zeroes of
ALC become:
FPLC = 2.2 kHz
FZ ESR= 20 kHz
Finally R1 = 5.6 kΩ and R2 = 3.3 kΩ.
The gain and phase bode diagrams are plotted respectively in Figure 11 and Figure 12.
GPWM s() Vcc
VOSCMAX VOSCMIN
–()
-------------------------------------------------------------=
VOSCMAX VOSCMIN
–KV
CC
•=
GPWM s() 1
K
----const==
Gs() GPWM s() R2
R1R2
+
--------------------
•AOs()•ALC
•s()=
Closing the loop A5974D
20/46 Doc ID 018774 Rev 1
Figure 11. Module plot
Figure 12. Phase plot
The cut-off frequency and the phase margin are:
Equation 16
FC33KHz=Phase margin = 49°
A5974D Application information
Doc ID 018774 Rev 1 21/46
8 Application information
8.1 Component selection
●Input capacitor
The input capacitor must be able to support the maximum input operating voltage and the
maximum RMS input current.
Since step-down converters draw current from the input in pulses, the input current is
squared and the height of each pulse is equal to the output current. The input capacitor has
to absorb all this switching current, which can be up to the load current divided by two (worst
case, with duty cycle of 50%). For this reason, the quality of these capacitors has to be very
high to minimize the power dissipation generated by the internal ESR, thereby improving
system reliability and efficiency. The critical parameter is usually the RMS current rating,
which must be higher than the RMS input current. The maximum RMS input current (flowing
through the input capacitor) is:
Equation 17
Where η is the expected system efficiency, D is the duty cycle and IO is the output DC
current. This function reaches its maximum value at D = 0.5 and the equivalent RMS current
is equal to IO divided by 2 (considering η = 1). The maximum and minimum duty cycles are:
Equation 18
and
Equation 19
IRMS IOD2D
2
•
η
----------------– D2
η2
-------+•=
DMAX
VOUT VF
+
VINMIN VSW
–
-------------------------------------=
DMIN
VOUT VF
+
VINMAX VSW
–
--------------------------------------=
Application information A5974D
22/46 Doc ID 018774 Rev 1
Where VF is the freewheeling diode forward voltage and VSW the voltage drop across the
internal PDMOS. Considering the range DMIN to DMAX, it is possible to determine the max
IRMS going through the input capacitor. Capacitors that can be considered are:
Electrolytic capacitors:
These are widely used due to their low price and their availability in a wide range of
RMS current ratings.
The only drawback is that, considering ripple current rating requirements, they are
physically larger than other capacitors.
Ceramic capacitors:
If available for the required value and voltage rating, these capacitors usually have a
higher RMS current rating for a given physical dimension (due to very low ESR).
The drawback is the considerably high cost.
Tantalum capacitors:
Very good, small tantalum capacitors with very low ESR are becoming more available.
However, they can occasionally burn if subjected to very high current during charge.
Therefore, it is better to avoid this type of capacitor for the input filter of the device. They
can, however, be subjected to high surge current when connected to the power supply.
●Output capacitor
The output capacitor is very important to meet the output voltage ripple requirement.
Using a small inductor value is useful to reduce the size of the choke but it increases the
current ripple. So, to reduce the output voltage ripple, a low ESR capacitor is required.
Nevertheless, the ESR of the output capacitor introduces a zero in the open loop gain,
which helps to increase the phase margin of the system. If the zero goes to a very high
frequency, its effect is negligible. For this reason, ceramic capacitors and very low ESR
capacitors in general should be avoided.
Tantalum and electrolytic capacitors are usually a good choice for this purpose. A list of
some tantalum capacitor manufacturers is provided in Table 7.: Output capacitor selection.
●Inductor
Table 6. List of ceramic capacitors for the A5974D
Manufacturer Series Capacitor value (µ) Rated voltage (V)
TAIYO YUDEN UMK325BJ106MM-T 10 50
MURATA GRM42-2 X7R 475K 50 4.7 50
Table 7. Output capacitor selection
Manufacturer Series Cap value (µF) Rated voltage (V) ESR (mΩ)
Sanyo POSCAP(1)
1. POSCAP capacitors have some characteristics which are very similar to tantalum.
TAE 47 to 680 2.5 to 10 25 to 35
TV 68 to 330 4 to 6.3 25 to 40
AVX TPS 100 to 470 4 to 35 50 to 200
KEMET T494/5 100 to 470 4 to 20 30 to 200
Sprague 595D 220 to 390 4 to 20 160 to 650
A5974D Application information
Doc ID 018774 Rev 1 23/46
The inductor value is very important as it fixes the ripple current flowing through the output
capacitor. The ripple current is usually fixed at 20 - 40% of Iomax, which is 0.6 - 1.2 A with
IOmax = 3 A. The approximate inductor value is obtained using the following formula:
Equation 20
where TON is the ON time of the internal switch, given by D · T. For example, with
VOUT = 3.3 V, VIN = 12 V and ΔIO = 0.9 A, the inductor value is about 12 µH. The peak
current through the inductor is given by:
Equation 21
and it can be observed that if the inductor value decreases, the peak current (which must be
lower than the current limit of the device) increases. So, when the peak current is fixed, a
higher inductor value allows a higher value for the output current. In the Table 8.: Inductor
selection, some inductor manufacturers are listed.
8.2 Layout considerations
The layout of switching DC-DC converters is very important to minimize noise and
interference. Power-generating portions of the layout are the main cause of noise and so
high switching current loop areas should be kept as small as possible and lead lengths as
short as possible.
High impedance paths (in particular the feedback connections) are susceptible to
interference, so they should be as far as possible from the high current paths. An layout
example is provided in Figure 13 below.
The input and output loops are minimized to avoid radiation and high frequency resonance
problems. The feedback pin connections to the external divider are very close to the device
to avoid pick-up noise. Another important issue is the ground plane of the board. Since the
package has an exposed pad, it is very important to connect it to an extended ground plane
in order to reduce the thermal resistance junction-to-ambient.
Table 8. Inductor selection
Manufacturer Series Inductor value (µH) Saturation current (A)
Coilcraft DO3316T 5.6 to 12 3.5 to 4.7
Coilcraft MSS1260T 5.6 to 15 3.5 to 8
Wurth Elektronik WE-PD L 4.7 to 27 3.55 to 6
LVIN VOUT
–()
ΔI
---------------------------------- TON
•=
IPK IO
ΔI
2
-----+=
Application information A5974D
24/46 Doc ID 018774 Rev 1
Figure 13. Layout example
8.3 Thermal considerations
8.3.1 Thermal resistance RthJA
RthJ-A is the equivalent static thermal resistance junction to ambient of the device; it can be
calculated as the parallel of many paths of heat conduction from the junction to the ambient.
For this device the path through the exposed pad is the one conducting the largest amount
of heat. The static RthJA measured on the application is about 40 °C/W.
The junction temperature of device will be:
Equation 22
The dissipated power of the device is tied to three different sources:
●Conduction losses due to the not insignificant RDSON, which are equal to:
Equation 23
Where D is the duty cycle of the application. Note that the duty cycle is theoretically given by
the ratio between VOUT and VIN, but in practice it is substantially higher than this value to
$'
&LQ ' &RXW
/
,QKLELWVLJQDO
WRRXWSXWYROWDJH
9LQ
9RXW
*QG
5
5
9(5<60$//+,*+&855(17
&,5&8/$7,1*3$7+720,1,0,=(
5$',$7,21$1'+,*+)5(48(1&<
5(621$1&(352%/(06
287387&$3$&,725
',5(&7/<
&211(&7('72
+($9<*5281'
&203(16$7,211(7:25.)$5
)520+,*+&855(173$7+6
0,1,0816,=(2))(('%$&.
3,1&211(&7,21672$92,'
3,&.83
&211(&7,2172
*5281'3/$1(
7+528*+9,$6
(;7(1'('*5281'3/$1(
217+(%277206,'(
TJTARthJA–PTOT
•+=
PON RDSON IOUT
()•2D•=
A5974D Application information
Doc ID 018774 Rev 1 25/46
compensate for the losses in the overall application. For this reason, the switching losses
related to the RDSON increases compared to an ideal case.
●Switching losses due to turning ON and OFF. These are derived using the following
equation:
Equation 24
Where TRISE and TFALL represent the switching times of the power element that cause the
switching losses when driving an inductive load (see Figure 14). TSW is the equivalent
switching time.
Figure 14. Switching losses
●Quiescent current losses.
Equation 25
Where IQ is the quiescent current.
Example:
–V
IN = 12 V
–V
OUT = 3.3 V
–I
OUT = 2.5 A
RDS(on) has a typical value of 0.25 @ 25 °C and increases up to a maximum value of 0.5. @
150 °C. We can consider a value of 0.4 Ω.
TSW is approximately 70 ns.
IQ has a typical value of 2.5 mA @ VIN = 12 V.
The overall losses are:
PSW VIN IOUT
TON TOFF
+()
2
------------------------------------
•• FSW VIN
=•IOUT TSW
•FSW
••=
PQVIN IQ
•=
Application information A5974D
26/46 Doc ID 018774 Rev 1
Equation 26
The junction temperature of device will be:
Equation 27
Equation 28
8.3.2 Thermal impedance ZTHJ-A(t)
The thermal impedance of the system, considered as the device in HSO8 package soldered
on the application board, takes on an important rule when the maximum output power is
limited by the static thermal performance and not by the electrical performance of the
device. Therefore the embedded power elements could manage an higher current but the
system is already taking away the maximum power generated by the internal losses.
In case the output power increases the thermal shutdown will be triggered because the
junction temperature triggers the designed thermal shutdown threshold.
The RTH is a static parameter of the package: it sets the maximum power loss which can be
generated from the system given the operation conditions.
If we suppose, as an example, TA = 60 °C, 140 °C is the maximum operating temperature
before triggering the thermal shutdown and RTH = 40 °C/W so the maximum power loss
achievable with the thermal performance of the system will be:
Figure 15. represents the estimation of Power losses for different output voltages at VIN=5V
and TAMB=60°C. The calculations are performed considering the RDS(on) of the power
element equal to 0.4A
PTOT RDSON IOUT
()•2DV
IN IOUT
•TSW
•FSW VIN IQ=•+•+•=
0.4 2.52
•0.3 12 2.5•70•10 9–
•250•10312 2.5•10 3–
•+•+•1.3W≅=
TJTARthJA–PTOT
•+=
TJ60 1.3 42 115°C≅•+=
PMAX DC
ΔT
RTH
-----------
TJ MAX TAMB
–
RTH
--------------------------------------80
40
------ 2W== ==
A5974D Application information
Doc ID 018774 Rev 1 27/46
Figure 15. Power losses estimation (VIN = 5 V, fSW = 250 kHz)
The red trace represents the maximum power which can be taken away as calculated
above, whilst the rest of the traces are the total internal losses for different output voltage.
The embedded conduction losses are proportional to the duty cycle required for the
conversion. Assuming the input voltage constant, the switching losses are proportional to
the output current while the quiescent losses can be considered as constant.
As a consequence in Figure 15. the maximum power losses is for VOUT=3.3V where the
system can manage a continuos output current up to 2.35 A. The device could deliver a
continuos output current up to 2.5 A to the load, however the maximum power loss of 2 W is
reached with an output current of 2.35 A, so the maximum output power is derated.
Figure 16. plots the power losses for VIN=12V and main output rails.
Figure 16. Power losses estimation (VIN = 12V, fSW = 250 kHz)
Application information A5974D
28/46 Doc ID 018774 Rev 1
At VIN=12V and VOUT=5V can deliver 2.5A continuously (see Figure 17.) because the total
power loss is now lower than 2W ((Δ switching loss +Δ quiescent loss) < Δ conduction loss).
As a consequence, the calculation of the internal power losses must be done for each
specific operating condition given by the final application.
In applications where the current to the output is pulsed, the thermal impedance should be
considered instead of the thermal resistance.
The thermal impedance of the system could be much lower than the thermal resistance,
which is a static parameter. As a consequence the maximum power losses can be higher
than 2 W if a pulsed output power is requested from the load:
So, depending on the pulse duration and its frequency, the maximum output current can be
delivered to the load.
The characterization of the thermal impedance is strictly dependent on the layout of the
board. In Figure 17. the measurement of the thermal impedance of the evaluation board of
the A5974D is provided.
Figure 17. Measurement of the thermal impedance of the evaluation board
As it can be see, for example, for load pulses with duration of 1 second, the actual thermal
impedance is lower than 20 °C/W. This means that, for short pulses, the device can deliver
an higher output current value.
PMAX t() ΔT
ZTH t()
-----------------
TJ MAX TAMB
–
ZTH t()
--------------------------------------==
H H   








7LPH>V @
1RUPDOL]HGWHPSHUDWXUHULVH>&@
A5974D Application information
Doc ID 018774 Rev 1 29/46
8.4 R.M.S. current of the embedded power MOSFET
As the A5974D embeds the high side switch and so the internal power dissipation is
sometimes the bottleneck for the output current capability (refer to Chapter 8.3 for the
estimation of the operating temperature).
Nevertheless, as mentioned in the general description on page 1 the device can manage a
continuos output current of 2.5 A in most of the application conditions.
However the rated maximum RMS current of the power elements is 2 A, where:
and the real duty cycle D:
Fixing the limit of 2 A for IRMS HS the maximum output current can be derived, as illustrated
in Figure 18..
Figure 18. Maximum continuos output current vs. duty cycle
8.5 Short-circuit protection
In overcurrent protection mode, when the peak current reaches the current limit, the device
reduces the TON down to its minimum value (approximately 250 nsec) and the switching
frequency to approximately one third of its nominal value even when synchronized to an
external signal (see Section 5.4: Current protection). In these conditions, the duty cycle is
strongly reduced and, in most applications, this is enough to limit the current to ILIM. In any
event, in case of heavy short-circuit at the output (VO = 0 V) and depending on the
IRMS HS ILOAD D⋅=
DVOUT RDS ON LS DCR+()ILOAD
⋅+
VIN RDS ON LS RDS ON HS
–()ILOAD
⋅+
----------------------------------------------------------------------------------------------------=
Application information A5974D
30/46 Doc ID 018774 Rev 1
application conditions (Vcc value and parasitic effect of external components) the current
peak could reach values higher than ILIM. This can be understood considering the inductor
current ripple during the ON and OFF phases:
●ON phase
Equation 29
●OFF phase
Equation 30
where VD is the voltage drop across the diode, DCRL is the series resistance of the inductor.
In short-circuit conditions VOUT is negligible so during TOFF the voltage across the inductor
is very small as equal to the voltage drop across parasitic components (typically the DCR of
the inductor and the VFW of the free wheeling diode) while during TON the voltage applied
the inductor is instead maximized as approximately equal to VIN.
So the Equation 29 and the Equation 30 in overcurrent conditions can be simplified to:
Equation 31
considering TON that has been already reduced to its minimum.
Equation 32
considering that fSW has been already reduced to one third of the nominal.
In case a short circuit at the output is applied and VIN = 12 V the inductor current is
controlled in most of the applications (see Figure 19). When the application must sustain the
short-circuit condition for an extended period, the external components (mainly the inductor
and diode) must be selected based on this value.
In case the VIN is very high, it could occur that the ripple current during TOFF (Equation 32)
does not compensate the current increase during TON(Equation 31). The Figure 21 shows
an example of a power up phase with VIN = VIN MAX = 36 V where ΔIL TON > ΔIL TOFF so the
current escalates and the balance between Equation 31 and Equation 32 occurs at a current
slightly higher than the current limit. This must be taken into account in particular to avoid
the risk of an abrupt inductor saturation.
IL TON
ΔVIN Vout
–DCRLRDSON
+()I⋅–
L
------------------------------------------------------------------------------------ TON
()=
IL TOFF
ΔVDVout DCRLI•++()–
L
--------------------------------------------------------------- TOFF
()=
IL TON
ΔVIN DCRLRDSON
+()I⋅–
L
---------------------------------------------------------------- TON MIN
()
VIN
L
---------250ns()≅=
IL TOFF
ΔVDVout DCRLI•++()–
L
--------------------------------------------------------------- 3T⋅SW
()
VDVout DCRLI•++()–
L
--------------------------------------------------------------- 12μs()≅=
A5974D Application information
Doc ID 018774 Rev 1 31/46
Figure 19. Short-circuit current VIN = 12 V
Figure 20. Short-circuit current VIN = 24 V
Figure 21. Short-circuit current VIN = 36 V
Application information A5974D
32/46 Doc ID 018774 Rev 1
8.6 Application circuit
Figure 22 shows the evaluation board application circuit, where the input supply voltage,
VCC, can range from 4 V to 36 V and the output voltage is adjustable from 1.235 V to 6.3 V
due to the voltage rating of the output capacitor,.
Figure 22. Evaluation board application circuit
Table 9. Component list
Reference Part number Description Manufacturer
C1 UMK325BJ106MM-T 10 µF, 50 V Taiyo Yuden
C2 68 nF, 5%, 0603
C3 100 pF, 5%, 0603
C4 33 nF, 5%, 0603
C10 POSCAP 6TVB330ML 330 µH, 25 mΩSanyo
R1 5.6 kΩ, 1%, 0.1 W 0603
R2 3.3 kΩ, 1%, 0.1 W 0603
R3 10 kΩ, 1%, 0.1 W 0603
D1 STPS3L40U 3 A, 40 V STMicroelectronics
L1 MSS1246T-153 15 µH, IRMS 20°C 2.85A Coilcraft
9729
VPDOOVLJQDO
SRZHUSODQH
9
73
*1'
73
*1'
&
10
&
10
&
10
&
10
5
10
5
10
5
.
5
.
&
Q)
&
Q)
5
10
5
10
5
.
5
.
73
9,1
73
9,1
73
6<1&
73
6<1&
&
S
&
S
&
Q)
9
&
Q)
9
'
6736/8
'
6736/8
5 105 10
-3-3
73
9287
73
9287
73
9
73
9
& 10& 10
&
Q
&
Q
73
*1'
73
*1'
5
10
5
10
& 10& 10
5 .5 .
&
X
&
X
9287
9&&
95()
(;3$'
,1+
&203
*1'
)%
6<1&
5 105 10
& 10& 10
&
X)
9
&
X)
9
/ X+/ X+
4
10
4
10
+
P&
U(
N&
A5974D Application information
Doc ID 018774 Rev 1 33/46
Figure 23. PCB layout (component side)
Figure 24. PCB layout (bottom side)
Figure 25. PCB layout (front side)
Application information A5974D
34/46 Doc ID 018774 Rev 1
8.7 Positive buck-boost regulator
The device can be used to implement a step-up/down converter with a positive output
voltage.
The output voltage is given by:
Equation 33
where the ideal duty cycle D for the buck boost converter is:
Equation 34
However, due to power losses in the passive elements, the real duty cycle is always higher
than this. The real value (that can be measured in the application) should be used in the
following formulas.
The peak current flowing in the embedded switch is:
Equation 35
while its average current is equal to:
Equation 36
This is due to the fact that the current flowing through the internal power switch is delivered
to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Ta b l e 4 for details) while the average current must be lower than the rated
DC current of the device.
As a consequence, the maximum output current is:
Equation 37
where ISW MAX represents the rated current of the device.
The current capability is reduced by the term (1-D) and so, for example, with a duty cycle of
0.5, and considering an average current through the switch of 3 A, the maximum output
current deliverable to the load is 1.5 A.
The figure below shows the schematic circuit of this topology for a 12 V output voltage and
5 V input.
VOUT VIN
D
1D–
-------------
⋅=
DVOUT
VIN VOUT
+
------------------------------=
ISW
ILOAD
1D–
---------------
IRIPPLE
2
--------------------+
ILOAD
1D–
---------------
VIN
2L⋅
-----------D
fSW
---------
⋅+==
ISW
ILOAD
1D–
---------------=
IOUT MAX ISW MAX 1D–()⋅≅
A5974D Application information
Doc ID 018774 Rev 1 35/46
Figure 26. Positive buck-boost regulator
8.8 Negative buck-boost regulator
In Figure 27, the schematic circuit for a standard buck-boost topology is shown. The output
voltage is:
Equation 38
where the ideal duty cycle D for the buck boost converter is:
Equation 39
The considerations given in Section 8.8 for the real duty cycle are still valid here.
Also the Equation 35 till Equation 37 can be used to calculate the maximum output current.
So, as an example, considering the conversion VIN = 12 V to VOUT = -5 V, ILOAD = 0.5 A:
Equation 40
Equation 41
An important thing to take into account is that the ground pin of the device is connected to
the negative output voltage. Therefore, the device is subjected to a voltage equal to VIN-VO,
which must be lower than 36 V (the maximum operating input voltage).
6).6
#
U&
6
#ERAMIC
$
3403,5
6##
#/-0
'.$
/54
&"
).(
39.# 62%&
!$
2
K
5
,
U(
K
K
#
N&
6
#

U&

6
6/546!
#
P&
$
3403,5
-
34..%,
6IN66OUT6)OUT!%FFICIENCY

VOUT VIN
–D
1D–
-------------
⋅=
DV–OUT
VIN VOUT
–
------------------------------=
D5
512+
----------------0.706==
ISW
ILOAD
1D–
--------------- 0.5
10.706–
------------------------1.7A== =
Application information A5974D
36/46 Doc ID 018774 Rev 1
Figure 27. Negative buck-boost regulator
8.9 Floating boost current generator
The A5974D doesn’t support a nominal boost conversion as this topology requires a low
side switch, however a floating boost can be useful in applications where the load can be
floating. A typical example is a current generator for LEDs driving as the LED does not
require a connection to the ground.
Figure 28. Floating boost topology
Figure 29. 350mA LED boost current source
6).6
#
U&
6
#ERAMIC
$
6##
#/-0
'.$
/54
&"
).(
39.# 62%&
2
K
,
U(
K
K
#
N&
6
#
U&
6
%32¾
! 6/546!
#
P&
#
U&
6
#ERAMIC
¾
 6/54
6).
GND small signal
ILED = 350mA
GND power
Rsense
KZM 330uF 25V (8x11,5)
KZM 330uF 25V (8x11,5)
VIN = 12VAC
C7
NM
C7
NM
TP3VAC TP3VAC
C8
470n
C8
470n
R3
15
R3
15
L1 22uHL1 22uH
R1
220K
R1
220K
C6
220n
C6
220n
C2
330u
C2
330u
C10
NM
C10
NM
C5
10uF
C5
10uF
D6 STPS3L40UD6 STPS3L40U
C3
470n
C3
470n
TP2VLED- TP2VLED-
D5 STPS3L40UD5 STPS3L40U
R5
2K
R5
2K
D1 STPS3L40UD1 STPS3L40U
C4
1u
C4
1u
R2
6.8k
R2
6.8k
D2 STPS3L40UD2 STPS3L40U
TP6VLED+ TP6VLED+
D3
BZX84-C33
D3
BZX84-C33
D4 STPS3L40UD4 STPS3L40U
TP1VDC TP1VDC
TP5GND TP5GND
VOUT
1
VCC
8
VREF
6
EX-PAD
9
INH
3
COMP
4
GND
7
FB
5
SYNC
2
U1 L5973DU1 L5973D
TP4VAC TP4VAC
R4
1.8
R4
1.8
Q1
BC327
Q1
BC327
C1
330u
C1
330u
C9 4.7u 50VC9 4.7u 50V
C11 NMC11 NM
A5974D Application information
Doc ID 018774 Rev 1 37/46
The device is powered from the output voltage so the maximum voltage drop across the
LEDs and resistor sense is 36 V.
The output voltage is given by:
Equation 42
where the ideal duty cycle D for the boost converter is:
Equation 43
As for positive and inverting buck boost (see Chapter 8.7 and Chapter 8.8.) the measured
real duty cycle has to be used to calculate the switch current level.
The peak current flowing in the embedded switch is:
Equation 44
while its average current is equal to:
Equation 45
This is due to the fact that the current flowing through the internal power switch is delivered
to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Ta b l e 4 for details) while the average current must be lower than the rated
DC current of the device.
As a consequence, the maximum output current is:
Equation 46
where ISW MAX represents the rated current of the device.
Figure 29 shows a tested circuit to implement a boost current source for high current LED
driving (350mA). To implement a boost conversion the LEDs string must be composed of a
minimum device number having a total voltage drop larger than maximum input voltage.
The input voltage can be either a DC or AC thanks to the input bridge rectifier. In case of a
DC voltage source D1, D2, D3, D4, C1, C2 can be removed from the circuit and 1μF
capacitor value can be used for C5.
VOUT
VIN
1D–
-------------=
DVOUT VIN
–
VOUT
------------------------------=
ISW
ILOAD
1D–
---------------
IRIPPLE
2
--------------------+
ILOAD
1D–
---------------
VIN
2L⋅
-----------D
fSW
---------
⋅+==
ISW
ILOAD
1D–
---------------=
IOUT MAX ISW MAX 1D–()⋅≅
Application information A5974D
38/46 Doc ID 018774 Rev 1
8.10 Synchronization example
See Chapter 5.3 for details.
Figure 30. Synchronization example
8.11 Compensation network with MLCC at the output
The A5974D standard compensation network (please refer to Figure 1. and Chapter 7)
introduces a single zero and a low frequency pole in the system bandwidth, so an high ESR
output capacitor must be selected to compensate the 180 degree phase shift given by the
LC double pole.
The selection of the output capacitor has to guarantee that the zero introduced by this
component is inside the designed system bandwidth and close to the frequency of the
double pole introduced by the LC filter. A general rule for the selection of this compound for
the system stability is provided in Equation 47.
Equation 47
MLCCs (multiple layer ceramic capacitor) with values in the range of 10 µF-22 µF and rated
voltages in the range of 10 V-25 V are available today at relatively low cost from many
manufacturers.
These capacitors have very low ESR values (a few mΩ) and thus are occasionally used for
the output filter in order to reduce the voltage ripple and the overall size of the application.
However the zero given by the output capacitor falls outside the designed bandwidth and so
the system becomes unstable with the standard compensation network.
The Figure 31 shows the type III compensation network stabilizing the system with ceramic
capacitors at the output (the optimum components value depends on the application). This
configuration introduces two zeros and a low frequency pole in the designed bandwidth so
guarantee a proper phase margin.
An excel worksheet supporting the compensation network design with ceramic output
capacitor is available at www.st.com in the A5974D product page. The tool, once selected
the power components, properly places the singularities and calculates the value of the
external components.
6##
'.$
/54
&"
).(
39.# 62%&
!$
#/-0
6##
'.$
/54
&"
).(
39.# 62%&
!$
#/-0
6).
fZ ESR
1
2πESR COUT
⋅⋅ ⋅
------------------------------------------------= bandwidth<
fLC f<Z ESR 10 fLC
⋅<
A5974D Application information
Doc ID 018774 Rev 1 39/46
Figure 31. MLCC compensation network circuit
8.12 External SOFT_START network
At start-up the device can quickly increase the current up to the current limit in order to
charge the output capacitor. If soft ramp-up of the output voltage is required, an external
soft-start network can be implemented as shown in Figure 32. The capacitor C is charged
up to an external reference through R and the BJT clamps the COMP pin.
This clamps the duty cycle, limiting the slew rate of the output voltage.
Figure 32. Soft-start network example
= 4V TO 36V
small signal
power plane
GNDGND
C8
470nF
C8
470nF
COUTCOUTR6
220K
R6
220K
C11
68nF
C11
68nF
R8
NM
R8
NM
R2R2
VINVIN
C3C3
C2
68nF
50V
C2
68nF
50V
D1
STPS3L40U
D1
STPS3L40U
R5R5
VOUTVOUT
C7C7
GNDGND
C6C6
R1R1
VOUT
1
VCC
8
VREF
6
EX-PAD
9
INH
3
COMP
4
GND
7
FB
5
SYNC
2
R4R4 C5C5
C1
10uF
50V
C1
10uF
50V
L1L1
Q1
BC327
Q1
BC327
=4V TO 36V
small signal
power plane
TP3
GND
TP3
GND
D1D1
Q1
BC327
Q1
BC327
R2
220K
R2
220K
TP1
VIN
TP1
VIN
C2C2
TP2
VOUT
TP2
VOUT
C3C3
C4
470nF
C4
470nF
TP4
GND
TP4
GND
VOUT 1
VCC
8
VREF
6
EX-PAD
9
INH
3
COMP
4
GND
7
FB 5
SYNC
2
U1
L597xD
U1
L597xD
L1L1
C5C5
R3R3
R1R1
C1
50V
C1
50V
R4R4
Typical characteristics A5974D
40/46 Doc ID 018774 Rev 1
9 Typical characteristics
Figure 33. Line regulator Figure 34. Shutdown current vs junction
temperature
Figure 35. Output voltage vs junction
temperature
Figure 36. Switching frequency vs
junction temperature
Figure 37. Quiescent current vs junction
temperature











9F F 9
7M &
7M &
9FF9
9R9
9R9





   
,VKGP$
9FF 9
7M&
7M&
9FF 9






   
9R9
9FF9





  
7M&
)VZ.+]
9FF9
9R 9




 

7M&
,TP$
9FF9
'&
A5974D Typical characteristics
Doc ID 018774 Rev 1 41/46
Figure 38. Junction temperature vs
output current VIN = 5 V
Figure 39. Junction temperature vs
output current VIN = 12 V
Figure 40. Efficiency vs output current
VIN = 12 V
Figure 41. Efficiency vs output current
VIN = 5 V
Package mechanical data A5974D
42/46 Doc ID 018774 Rev 1
10 Package mechanical data
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK®
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
Table 10. HSOP8 mechanical data
Dim.
mm inch
Min. Typ. Max. Min. Typ. Max.
A 1.70 0.0669
A1 0.00 0.10 0.00 0.0039
A2 1.25 0.0492
b 0.31 0.51 0.0122 0.0201
c 0.17 0.25 0.0067 0.0098
D 4.80 4.90 5.00 0.1890 0.1929 0.1969
D1 3 3.1 3.2 0.118 0.122 0.126
E 5.80 6.00 6.20 0.2283 0.2441
E1 3.80 3.90 4.00 0.1496 0.1575
E2 2.31 2.41 2.51 0.091 0.095 0.099
e1.27
h 0.25 0.50 0.0098 0.0197
L 0.40 1.27 0.0157 0.0500
k0° (min), 8° (max)
ccc 0.10 0.0039
A5974D Package mechanical data
Doc ID 018774 Rev 1 43/46
Figure 42. Package dimensions
$
Ordering information A5974D
44/46 Doc ID 018774 Rev 1
11 Ordering information
Table 11. Ordering information
Order codes Package Packaging
A5974D HSOP8 Tu b e
A5974DTR Tape and reel
A5974D Revision history
Doc ID 018774 Rev 1 45/46
12 Revision history
Table 12. Document revision history
Date Revision Changes
17-May-2011 1 Initial release
A5974D
46/46 Doc ID 018774 Rev 1
Please Read Carefully:
Information in this document is provided solely in connection with ST products. STMicroelectronics NV and its subsidiaries (“ST”) reserve the
right to make changes, corrections, modifications or improvements, to this document, and the products and services described herein at any
time, without notice.
All ST products are sold pursuant to ST’s terms and conditions of sale.
Purchasers are solely responsible for the choice, selection and use of the ST products and services described herein, and ST assumes no
liability whatsoever relating to the choice, selection or use of the ST products and services described herein.
No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted under this document. If any part of this
document refers to any third party products or services it shall not be deemed a license grant by ST for the use of such third party products
or services, or any intellectual property contained therein or considered as a warranty covering the use in any manner whatsoever of such
third party products or services or any intellectual property contained therein.
UNLESS OTHERWISE SET FORTH IN ST’S TERMS AND CONDITIONS OF SALE ST DISCLAIMS ANY EXPRESS OR IMPLIED
WARRANTY WITH RESPECT TO THE USE AND/OR SALE OF ST PRODUCTS INCLUDING WITHOUT LIMITATION IMPLIED
WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS
OF ANY JURISDICTION), OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT.
UNLESS EXPRESSLY APPROVED IN WRITING BY AN AUTHORIZED ST REPRESENTATIVE, ST PRODUCTS ARE NOT
RECOMMENDED, AUTHORIZED OR WARRANTED FOR USE IN MILITARY, AIR CRAFT, SPACE, LIFE SAVING, OR LIFE SUSTAINING
APPLICATIONS, NOR IN PRODUCTS OR SYSTEMS WHERE FAILURE OR MALFUNCTION MAY RESULT IN PERSONAL INJURY,
DEATH, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE. ST PRODUCTS WHICH ARE NOT SPECIFIED AS "AUTOMOTIVE
GRADE" MAY ONLY BE USED IN AUTOMOTIVE APPLICATIONS AT USER’S OWN RISK.
Resale of ST products with provisions different from the statements and/or technical features set forth in this document shall immediately void
any warranty granted by ST for the ST product or service described herein and shall not create or extend in any manner whatsoever, any
liability of ST.
ST and the ST logo are trademarks or registered trademarks of ST in various countries.
Information in this document supersedes and replaces all information previously supplied.
The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners.
© 2011 STMicroelectronics - All rights reserved
STMicroelectronics group of companies
Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan -
Malaysia - Malta - Morocco - Philippines - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America
www.st.com