LM3153MHE-3.3/NOPB - NATIONAL SEMICONDUCTOR

LM3151,LM3152,LM3153

LM3151/LM3152/LM3153 SIMPLE SWITCHER ® CONTROLLER, High Input Voltage

Synchronous Step-Down

Literature Number: SNVS562G

LM3151/LM3152/LM3153

March 4, 2011

SIMPLE SWITCHER® CONTROLLER, High Input Voltage

Synchronous Step-Down

General Description

The LM3151/2/3 SIMPLE SWITCHER® Controller is an easy

to use and simplified step down power controller capable of

providing up to 12A of output current in a typical application.

Operating with an input voltage range from 6V-42V, the

LM3151/2/3 features a fixed output voltage of 3.3V, and fea-

tures switching frequencies of 250 kHz, 500 kHz, and 750

kHz. The synchronous architecture provides for highly effi-

cient designs. The LM3151/2/3 controller employs a Constant

On-Time (COT) architecture with a proprietary Emulated Rip-

ple Mode (ERM) control that allows for the use of low ESR

output capacitors, which reduces overall solution size and

output voltage ripple. The Constant On-Time (COT) regula-

tion architecture allows for fast transient response and re-

quires no loop compensation, which reduces external com-

ponent count and reduces design complexity.

Fault protection features such as thermal shutdown, under-

voltage lockout, over-voltage protection, short-circuit protec-

tion, current limit, and output voltage pre-bias startup allow for

a reliable and robust solution.

The LM3151/2/3 SIMPLE SWITCHER® concept provides for

an easy to use complete design using a minimum number of

external components and National’s WEBENCH® online de-

sign tool. WEBENCH® provides design support for every step

of the design process and includes features such as external

component calculation with a new MOSFET selector, electri-

cal simulation, thermal simulation, and Build-It boards for

prototyping.

Features

■PowerWise® step-down controller

■6V to 42V Wide input voltage range

■Fixed output voltage of 3.3V

■Fixed switching frequencies of 250 kHz/500 kHz/750 kHz

■No loop compensation required

■Fully WEBENCH® enabled

■Low external component count

■Constant On-Time control

■Ultra-Fast transient response

■Stable with low ESR capacitors

■Output voltage pre-bias startup

■Valley current limit

■Programmable soft-start

Typical Applications

■Telecom

■Networking Equipment

■Routers

■Security Surveillance

■Power Modules

Typical Application

30053201

SIMPLE SWITCHER® is a registered trademark of National Semiconductor Corporation

LM3151/LM3152/LM3153 SIMPLE SWITCHER® CONTROLLER, High Input Voltage Synchronous

Step-Down

Connection Diagram

30053202

eTSSOP-14

Ordering Information

Order Number Package Type NSC Package

Drawing

Input

Voltage

Range

Output

Voltage

Switching

Frequency Supplied As

LM3151MH-3.3

eTSSOP-14 MXA14A 6V - 42V 3.3V 250KHz

94 Units per Anti-Static

Tube

LM3151MHE-3.3 250 Units in Tape and

Reel

LM3151MHX-3.3 2500 Units in Tape and

Reel

LM3152MH-3.3

eTSSOP-14 MXA14A 6V - 33V 3.3V 500KHz

94 Units per Anti-Static

Tube

LM3152MHE-3.3 250 Units in Tape and

Reel

LM3152MHX-3.3 2500 Units in Tape and

Reel

LM3153MH-3.3

eTSSOP-14 MXA14A 8V - 18V 3.3V 750KHz

94 Units per Anti-Static

Tube

LM3153MHE-3.3 250 Units in Tape and

Reel

LM3153MHX-3.3 2500 Units in Tape and

Reel

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LM3151/LM3152/LM3153

Pin Descriptions

Pin Name Description Function

1 VCC Supply Voltage for

FET Drivers

Nominally regulated to 5.95V. Connect a 1 µF to 2.2 µF decoupling capacitor from this

pin to ground.

2 VIN Input Supply Voltage Supply pin to the device. Nominal input range is 6V to 42V. See ordering information

for Vin limitations.

3 EN Enable To enable the IC apply a logic high signal to this pin greater than 1.26V typical or leave

floating. To disable the part, ground the EN pin.

4 FB Feedback

Internally connected to the resistor divider network which sets the fixed output voltage.

This pin also senses the output voltage faults such a over-voltage and short circuit

conditions.

5,9 SGND Signal Ground Ground for all internal bias and reference circuitry. Should be connected to PGND at a

single point.

6 SS Soft-Start An internal 7.7 µA current source charges an external capacitor to provide the soft-start

function.

7,8 N/C Not Connected Internally not electrically connected. These pins may be left unconnected or connected

to ground.

10 SW Switch Node Switch pin of controller and high-gate driver lower supply rail. A boost capacitor is also

connected between this pin and BST pin

11 HG High-Side Gate

Drive

Gate drive signal to the high-side NMOS switch. The high-side gate driver voltage is

supplied by the differential voltage between the BST pin and SW pin.

12 BST Connection for

Bootstrap Capacitor

High-gate driver upper supply rail. Connect a 0.33 µF-0.47 µF capacitor from SW pin to

this pin. An internal diode charges the capacitor during the high-side switch off-time. Do

not connect to an external supply rail.

13 LG Low-Side Gate Drive Gate drive signal to the low-side NMOS switch. The low-side gate driver voltage is

supplied by VCC.

14 PGND Power Ground Synchronous rectifier MOSFET source connection. Tie to power ground plane. Should

be tied to SGND at a single point.

EP EP Exposed Pad Exposed die attach pad should be connected directly to SGND. Also used to help

dissipate heat out of the IC.

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LM3151/LM3152/LM3153

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN to GND -0.3V to 47V

SW to GND -3V to 47V

BST to SW -0.3V to 7V

BST to GND -0.3V to 52V

All Other Inputs to GND -0.3V to 7V

ESD Rating (Note 2) 2kV

Storage Temperature Range -65°C to +150°C

Operating Ratings (Note 1)

VIN 6V to 42V

Junction Temperature Range (TJ)−40°C to + 125°C

EN 0V to 5V

Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the

junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical

correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.

Unless otherwise stated the following conditions apply: VIN = 18V.

Symbol Parameter Conditions Min Typ Max Units

Start-Up Regulator, VCC

VCC CVCC = 1 µF, 0 mA to 40 mA 5.65 5.95 6.25 V

VIN - VCC VIN - VCC Dropout Voltage IVCC = 2 mA, Vin = 5.5V 40 mV

IVCC = 30 mA, Vin = 5.5V 330

IVCCL VCC Current Limit (Note 3) VCC = 0V 65 100 mA

VCCUVLO

VCC Under-voltage Lockout threshold

(UVLO) VCC Increasing 4.75 5.1 5.40 V

VCC-UVLO-HYS VCC UVLO Hysteresis VCC Decreasing 475 mV

tCC-UVLO-D VCC UVLO Filter Delay 3 µs

IIN Input Operating Current No Switching 3.6 5.2 mA

IIN-SD

Input Operating Current, Device

Shutdown VEN = 0V 32 55 µA

GATE Drive

IQ-BST Boost Pin Leakage VBST – VSW = 6V 2 nA

RDS-HG-Pull-Up HG Drive Pull–Up On-Resistance IHG Source = 200 mA 5 Ω

RDS-HG-Pull-Down HG Drive Pull–Down On-Resistance IHG Sink = 200 mA 3.4 Ω

RDS-LG-Pull-Up LG Drive Pull–Up On-Resistance ILG Source = 200 mA 3.4 Ω

RDS-LG-Pull-Down LG Drive Pull–Down On-Resistance ILG Sink = 200 mA 2 Ω

Soft-Start

ISS SS Pin Source Current VSS = 0V 5.9 7.7 9.5 mA

ISS-DIS SS Pin Discharge Current 200 µA

Current Limit

VCL Current Limit Voltage Threshold 175 200 225 mV

ON/OFF Timer

tON-MIN ON Timer Minimum Pulse Width 200 ns

tOFF OFF Timer Minimum Pulse Width 370 525 ns

Enable Input

VEN EN Pin Input Threshold Trip Point VEN Rising 1.14 1.20 1.26 V

VEN-HYS EN Pin threshold Hysteresis VEN Falling 120 mV

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LM3151/LM3152/LM3153

Symbol Parameter Conditions Min Typ Max Units

Boost Diode

VfForward Voltage IBST = 2 mA 0.7 V

IBST = 30 mA 1 V

Thermal Characteristics

TSD

Thermal Shutdown Rising 165 °C

Thermal Shutdown Hysteresis Falling 15 °C

θJA Junction to Ambient

4 Layer JEDEC Printed Circuit

Board, 9 Vias, No Air Flow

°C/W

2 Layer JEDEC Printed Circuit

Board. No Air Flow

140

θJC Junction to Case No Air Flow 4 °C/W

3.3V Output Option

Symbol Parameter Conditions Min Typ Max Units

VOUT Output Voltage 3.234 3.3 3.366 V

VOUT-OV Output Voltage Over-Voltage Threshold 3.83 4.00 4.17 V

VIN-MAX Maximum Input Voltage (Note 4)

LM3151-3.3 42

VLM3152-3.3 33

LM3153-3.3 18

VIN-MIN Minimum Input Voltage (Note 4)

LM3151-3.3 6

VLM3152-3.3 6

LM3153-3.3 8

fSSwitching Frequency

LM3151-3.3, RON = 115 kΩ 250

kHz

LM3152-3.3, RON = 51 kΩ 500

LM3153-3.3, RON = 32 kΩ 750

tON On-Time

LM3151-3.3, RON = 115 kΩ 730

LM3152-3.3, RON = 51 kΩ 400

LM3153-3.3, RON = 32 kΩ 330

RFB FB Resistance to Ground 566 kΩ

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but does not guarantee specific performance limits. For guaranteed specifications and conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test Method is per JESD-22-A114.

Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.

Note 4: The input voltage range is dependent on minimum on-time, off-time, and therefore frequency, and is also affected by optimized MOSFET selection.

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LM3151/LM3152/LM3153

Simplified Block Diagram

30053203

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LM3151/LM3152/LM3153

Typical Performance Characteristics

Boost Diode Forward Voltage vs. Temperature

30053240

Quiescent Current vs. Temperature

30053242

Soft-Start Current vs. Temperature

30053243

VCC Current Limit vs. Temperature

30053247

VCC Dropout vs. Temperature

30053248

VCC vs. Temperature

30053249

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LM3151/LM3152/LM3153

VCL vs. Temperature

30053282

On-Time vs. Temperature (250 kHz)

30053283

On-Time vs. Temperature (500 kHz)

30053284

On-Time vs. Temperature (750 kHz)

30053286

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LM3151/LM3152/LM3153

Theory of Operation

The LM3151/2/3 synchronous step-down SIMPLE SWITCH-

ER® Controller employs a Constant On-Time (COT) archi-

tecture which is a derivative of the hysteretic control scheme.

COT relies on a fixed switch on-time to regulate the output.

The on-time of the high-side switch is set internally by resistor

RON. The LM3151/2/3 automatically adjusts the on-time in-

versely with the input voltage to maintain a constant frequen-

cy. Assuming an ideal system and VIN is much greater than

1V, the following approximations can be made:

The on-time, tON:

Where K = 100 pC, and RON is specified in the electrical char-

acteristics table.

Control is based on a comparator and the on-timer, with the

output voltage feedback (FB) attenuated and then compared

with an internal reference of 0.6V. If the attenuated FB level

is below the reference, the high-side switch is turned on for a

fixed time, tON, which is determined by the input voltage and

the internal resistor, RON. Following this on-time, the switch

remains off for a minimum off-time, tOFF, as specified in the

Electrical Characteristics table or until the attenuated FB volt-

age is less than 0.6V. This switching cycle will continue while

maintaining regulation. During continuous conduction mode

(CCM), the switching frequency depends only on duty cycle

and on-time. The duty cycle can be calculated as:

Where the switching frequency of a COT regulator is:

Typical COT hysteretic controllers need a significant amount

of output capacitor ESR to maintain a minimum amount of

ripple at the FB pin in order to switch properly and maintain

efficient regulation. The LM3151/2/3 however utilizes propri-

etary, Emulated Ripple Mode Control Scheme (ERM) that

allows the use of ceramic output capacitors without additional

equivalent series resistance (ESR) compensation. Not only

does this reduce the need for output capacitor ESR, but also

significantly reduces the amount of output voltage ripple seen

in a typical hysteretic control scheme. The output ripple volt-

age can become so low that it is comparable to voltage-mode

and current-mode control schemes.

Regulation Comparator

The output voltage is sampled through the FB pin and then

divided down by two internal resistors and compared to the

internal reference voltage of 0.6V by the error comparator. In

normal operation, an on-time period is initiated when the sam-

pled output voltage at the input of the error comparator falls

below 0.6V. The high-side switch stays on for the specified

on-time, causing the sampled voltage on the error comparator

input to rise above 0.6V. After the on-time period, the high-

side switch stays off for the greater of the following:

1) Minimum off time as specified in the electrical characteris-

tics table

2) The error comparator sampled voltage falls below 0.6V

Over-Voltage Comparator

The over-voltage comparator is provided to protect the output

from over-voltage conditions due to sudden input line voltage

changes or output loading changes. The over-voltage com-

parator continuously monitors the attenuated FB voltage ver-

sus a 0.72V internal reference. If the voltage at FB rises above

0.72V the on-time pulse is immediately terminated. This con-

dition can occur if the input or the output load changes sud-

denly. Once the over-voltage protection is activated, the HG

and LG signals remain off until the attenuated FB voltage falls

below 0.72V.

Current Limit

Current limit detection occurs during the off-time by monitor-

ing the current through the low-side switch. If during the off-

time the current in the low-side switch exceeds the user

defined current limit value, the next on-time cycle is immedi-

ately terminated. Current sensing is achieved by comparing

the voltage across the low-side switch against an internal ref-

erence value, VCL, of 200 mV. If the voltage across the low-

side switch exceeds 200 mV, the current limit comparator will

trigger logic to terminate the next on-time cycle. The current

limit ICL, can be determined as follows:

Where IOCL is the user-defined average output current limit

value, RDS(ON)max is the resistance value of the low-side FET

at the expected maximum FET junction temperature, VCL is

the internal current limit reference voltage and Tj is the junc-

tion temperature of the LM3151/2/3.

Figure 1 illustrates the inductor current waveform. During nor-

mal operation, the output current ripple is dictated by the

switching of the FETs. The current through the low-side

switch, Ivalley, is sampled at the end of each switching cycle

and compared to the current limit threshold voltage, VCL. The

valley current can be calculated as follows:

Where IOUT is the average output current and ΔIL is the peak-

to-peak inductor ripple current.

If an overload condition occurs, the current through the low-

side switch will increase which will cause the current limit

comparator to trigger the logic to skip the next on-time cycle.

The IC will then try to recover by checking the valley current

during each off-time. If the valley current is greater than or

equal to ICL, then the IC will keep the low-side FET on and

allow the inductor current to further decay.

Throughout the whole process, regardless of the load current,

the on-time of the controller will stay constant and thereby the

positive ripple current slope will remain constant. During each

on-time the current ramps up an amount equal to:

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LM3151/LM3152/LM3153

The valley current limit feature prevents current runaway con-

ditions due to propagation delays or inductor saturation since

the inductor current is forced to decay following any overload

conditions.

30053212

FIGURE 1. Inductor Current - Current Limit Operation

Short-Circuit Protection

The LM3151/2/3 will sense a short-circuit on the output by

monitoring the output voltage. When the attenuated feedback

voltage has fallen below 60% of the reference voltage, Vref x

0.6 (≈ 0.36V), short-circuit mode of operation will start. During

short-circuit operation, the SS pin is discharged and the out-

put voltage will fall to 0V. The SS pin voltage, VSS, is then

ramped back up at the rate determined by the SS capacitor

and ISS until VSS reaches 0.7V. During this re-ramp phase, if

the short-circuit fault is still present the output current will be

equal to the set current limit. Once the soft-start voltage

reaches 0.7V the output voltage is sensed again and if the

attenuated VFB is still below Vref x 0.6 then the SS pin is dis-

charged again and the cycle repeats until the short-circuit fault

is removed.

Soft-Start

The soft-start (SS) feature allows the regulator to gradually

reach a steady-state operating point, which reduces start-up

stresses and current surges. At turn-on, while VCC is below

the under-voltage threshold, the SS pin is internally grounded

and VOUT is held at 0V. The SS capacitor is used to slowly

ramp VFB from 0V to it's final output voltage as programmed

by the internal resistor divider. By changing the soft-start ca-

pacitor value, the duration of start-up can be changed ac-

cordingly. The start-up time can be calculated using the

following equation:

Where tSS is measured in seconds, Vref = 0.6V and ISS is the

soft-start pin source current, which is typically 7.7 µA (refer to

electrical characteristics table).

An internal switch grounds the SS pin if VCC is below the

under-voltage lockout threshold, if a thermal shutdown oc-

curs, or if the EN pin is grounded. By using an externally

controlled switch, the output voltage can be shut off by

grounding the SS pin.

During startup the LM3151/2/3 will operate in diode emulation

mode, where the low-side gate LG will turn off and remain off

when the inductor current falls to zero. Diode emulation mode

allows for start up into a pre-biased output voltage. When soft-

start is greater than 0.7V, the LM3151/2/3 will remain in

continuous conduction mode. During diode emulation mode

at current limit the low-gate will remain off when the inductor

current is off.

The soft start time should be greater than the rise time spec-

ified by,

tSS ≥ (VOUT x COUT) / (IOCL - IOUT)

Enable/Shutdown

The EN pin can be activated by either leaving the pin floating

due to an internal pull up resistor to VIN or by applying a logic

high signal to the EN pin of 1.26V or greater. The LM3151/2/3

can be remotely shut down by taking the EN pin below 1.02V.

Low quiescent shutdown is achieved when VEN is less than

0.4V. During low quiescent shutdown the internal bias circuit-

ry is turned off.

The LM3151/2/3 has certain fault conditions that can trigger

shutdown, such as over-voltage protection, current limit, un-

der-voltage lockout, or thermal shutdown. During shutdown,

the soft-start capacitor is discharged. Once the fault condition

is removed, the soft-start capacitor begins charging, allowing

the part to start up in a controlled fashion. In conditions where

there may be an open drain connection to the EN pin, it may

be necessary to add a 1000 pF bypass capacitor to this pin.

This will help decouple noise from the EN pin and prevent

false disabling.

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LM3151/LM3152/LM3153

Thermal Protection

The LM3151/2/3 should be operated such that the junction

temperature does not exceed the maximum operating junc-

tion temperature. An internal thermal shutdown circuit, which

activates at 165°C (typical), takes the controller to a low-pow-

er reset state by disabling the buck switch and the on-timer,

and grounding the SS pin. This feature helps prevent catas-

trophic failures from accidental device overheating. When the

junction temperature falls back below 150°C the SS pin is re-

leased and normal operation resumes.

Design Guide

The design guide provides the equations required to design

with the LM3151/2/3 SIMPLE SWITCHER® Controller.

WEBENCH® design tool can be used with or in place of this

section for a more complete and simplified design process.

1. Define Power Supply Operating Conditions

a. Maximum and Minimum DC Input voltage

b. Maximum Expected Load Current during normal operation

c. Target Switching Frequency

2. Determine which IC Controller to Use

The desired input voltage range will determine which version

of the LM3151/2/3 controller will be chosen. The higher

switching frequency options allow for physically smaller in-

ductors but efficiency may decrease.

3. Determine Inductor Required Using Figure 2

To use the nomograph below calculate the inductor volt-mi-

crosecond constant ET from the following formula:

Where fS is in kHz units. The intersection of the Load Current

and the Volt-microseconds lines on the chart below will de-

termine which inductors are capable for use in the design. The

chart shows a sample of parts that can be used. The offline

calculator tools and WEBENCH® will fully calculate the re-

quirements for the components needed for the design.

30053252

FIGURE 2. Inductor Nomograph

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LM3151/LM3152/LM3153

TABLE 1. Inductor Selection Table

Inductor Designator Inductance (µH) Current (A) Part Name Vendor

L01 47 7-9

L02 33 7-9 SER2817H-333KL COILCRAFT

L03 22 7-9 SER2814H-223KL COILCRAFT

L04 15 7-9 7447709150 WURTH

L05 10 7-9 RLF12560T-100M7R5 TDK

L06 6.8 7-9 B82477-G4682-M EPCOS

L07 4.7 7-9 B82477-G4472-M EPCOS

L08 3.3 7-9 DR1050-3R3-R COOPER

L09 2.2 7-9 MSS1048-222 COILCRAFT

L10 1.5 7-9 SRU1048-1R5Y BOURNS

L11 1 7-9 DO3316P-102 COILCRAFT

L12 0.68 7-9 DO3316H-681 COILCRAFT

L13 33 9-12

L14 22 9-12 SER2918H-223 COILCRAFT

L15 15 9-12 SER2814H-153KL COILCRAFT

L16 10 9-12 7447709100 WURTH

L17 6.8 9-12 SPT50H-652 COILCRAFT

L18 4.7 9-12 SER1360-472 COILCRAFT

L19 3.3 9-12 MSS1260-332 COILCRAFT

L20 2.2 9-12 DR1050-2R2-R COOPER

L21 1.5 9-12 DR1050-1R5-R COOPER

L22 1 9-12 DO3316H-102 COILCRAFT

L23 0.68 9-12

L24 0.47 9-12

L25 22 12-15 SER2817H-223KL COILCRAFT

L26 15 12-15

L27 10 12-15 SER2814L-103KL COILCRAFT

L28 6.8 12-15 7447709006 WURTH

L29 4.7 12-15 7447709004 WURTH

L30 3.3 12-15

L31 2.2 12-15

L32 1.5 12-15 MLC1245-152 COILCRAFT

L33 1 12-15

L34 0.68 12-15 DO3316H-681 COILCRAFT

L35 0.47 12-15

L36 0.33 12-15 DR73-R33-R COOPER

L37 22 15-

L38 15 15- SER2817H-153KL COILCRAFT

L39 10 15- SER2814H-103KL COILCRAFT

L40 6.8 15-

L41 4.7 15- SER2013-472ML COILCRAFT

L42 3.3 15- SER2013-362L COILCRAFT

L43 2.2 15-

L44 1.5 15- HA3778-AL COILCRAFT

L45 1 15- B82477-G4102-M EPCOS

L46 0.68 15-

L47 0.47 15-

L48 0.33 15-

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LM3151/LM3152/LM3153

4. Determine Output Capacitance

Typical hysteretic COT converters similar to the LM3151/2/3

require a certain amount of ripple that is generated across the

ESR of the output capacitor and fed back to the error com-

parator. Emulated Ripple Mode control built into the

LM3151/2/3 will recreate a similar ripple signal and thus the

requirement for output capacitor ESR will decrease compared

to a typical Hysteretic COT converter. The emulated ripple is

generated by sensing the voltage signal across the low-side

FET and is then compared to the FB voltage at the error com-

parator input to determine when to initiate the next on-time

period.

COmin = 70 / (fs2 x L)

The maximum ESR allowed to prevent over-voltage protec-

tion during normal operation is:

ESRmax = (80 mV x L) / ETmin

ETmin is calculated using VIN-MIN

The minimum ESR must meet both of the following criteria:

ESRmin ≥ (15 mV x L) / ETmax

ESRmin ≥ [ETmax / (VIN - VOUT)]/ CO

ETmax is calculated using VIN-MAX.

Any additional parallel capacitors should be chosen so that

their effective impedance will not negatively attenuate the

output ripple voltage.

5. MOSFET Selection

The high-side and low-side FETs must have a drain to source

(VDS) rating of at least 1.2 x VIN.

The gate drive current from VCC must not exceed the mini-

mum current limit of VCC. The drive current from VCC can be

calculated with:

IVCCdrive = Qgtotal x fS

Where, Qgtotal is the combined total gate charge of the high-

side and low-side FETs.

Use the following equations to calculate the current limit, ICL,

as shown in Figure 1.

Tj is the junction temperature of the LM3151/2/3.

The plateau voltage of the FET VGS vs Qg curve, as shown in

Figure 3 must be less than VCC - 750 mV.

30053281

FIGURE 3. Typical MOSFET Gate Charge Curve

See following design example for estimated power dissipation

calculation.

6. Calculate Input Capacitance

The main parameters for the input capacitor are the voltage

rating, which must be greater than or equal to the maximum

DC input voltage of the power supply, and its rms current rat-

ing. The maximum rms current is approximately 50% of the

maximum load current.

Where, ΔVIN-MAX is the maximum allowable input ripple volt-

age. A good starting point for the input ripple voltage is 5% of

VIN.

When using low ESR ceramic capacitors on the input of the

LM3151/2/3 a resonant circuit can be formed with the

impedance of the input power supply and parasitic impedance

of long leads/PCB traces to the LM3151/2/3 input capacitors.

It is recommended to use a damping capacitor under these

circumstances, such as aluminum electrolytic that will prevent

ringing on the input. The damping capacitor should be chosen

to be approximately 5 times greater than the parallel ceramic

capacitors combination. The total input capacitance should

be greater than 10 times the input inductance of the power

supply leads/pcb trace. The damping capacitor should also

be chosen to handle its share of the rms input current which

is shared proportionately with the parallel impedance of the

ceramic capacitors and aluminum electrolytic at the

LM3151/2/3 switching frequency.

The CBYP capacitor should be placed directly at the VIN pin.

The recommended value is 0.1 µF.

7. Calculate Soft-Start Capacitor

Where tSS is the soft-start time in seconds and Vref = 0.6V.

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LM3151/LM3152/LM3153

8. CVCC, and CBST and CEN

CVCC should be placed directly at the VCC pin with a recom-

mended value of 1 µF to 2.2 µF. For input voltage ranges that

include voltages below 8V a 1 µF capacitor must be used for

CVCC. CBST creates a voltage used to drive the gate of the

high-side FET. It is charged during the SW off-time. The rec-

ommended value for CBST is 0.47 µF. The EN bypass capac-

itor, CEN, recommended value is 1000 pF when driving the EN

pin from open drain type of signal.

Design Example

30053261

FIGURE 4. Design Example Schematic

1. Define Power Supply Operating Conditions

a. VOUT = 3.3V

b. VIN-MIN = 6V, VIN-TYP = 12V, VIN-MAX = 24V

c. Typical Load Current = 12A, Max Load Current = 15A

d. Soft-Start time tSS = 5 ms

2. Determine which IC Controller to Use

The LM3151 and LM3152 allow for the full input voltage

range. However, from buck converter basic theory, the higher

switching frequency will allow for a smaller inductor. There-

fore, the LM3152-3.3 500 kHz part is chosen so that a smaller

inductor can be used.

3. Determine Inductor Required

a. ET = (24-3.3) x (3.3/24) x (1000/500) = 5.7 V µs

b. From the inductor nomograph a 12A load and 5.7 V µs cal-

culation corresponds to a L44 type of inductor.

c. Using the inductor designator L44 in Table 1 the Coilcraft

HA3778-AL 1.65 µH inductor is chosen.

4. Determine Output Capacitance

The voltage rating on the output capacitor should be greater

than or equal to the output voltage. As a rule of thumb most

capacitor manufacturers suggests not to exceed 90% of the

capacitor rated voltage. In the case of multilayer ceramics the

capacitance will tend to decrease dramatically as the applied

voltage is increased towards the capacitor rated voltage. The

capacitance can decrease by as much as 50% when the ap-

plied voltage is only 30% of the rated voltage. The chosen

capacitor should also be able to handle the rms current which

is equal to:

For this design the chosen ripple current ratio, r = 0.3, repre-

sents the ratio of inductor peak-to-peak current to load current

Iout. A good starting point for ripple ratio is 0.3 but it is ac-

ceptable to choose r between 0.25 to 0.5. The nomographs

in this datasheet all use 0.3 as the ripple current ratio.

Irmsco = 1A

tON = (3.3V/12V) / 500 kHz = 550 ns

Minimum output capacitance is:

COmin = 70 / (fS2 x L)

COmin = 70 / (500 kHz2 x 1.65 µH) = 169 µF

The maximum ESR allowed to prevent over-voltage protec-

tion during normal operation is:

ESRmax = (80 mV x L) / ET

ESRmax = (80 mV x 1.65 µH) / 5.7 V µs

ESRmax = 23 mΩ

The minimum ESR must meet both of the following criteria:

ESRmin ≥ (15 mV x L) / ET

www.national.com 14

LM3151/LM3152/LM3153

ESRmin ≥ [ET / (VIN - VOUT)] / CO

ESRmin ≥ (15 mV x 1.65 µH) / 5.7 V µs = 4.3 mΩ

ESRmin ≥ [5.7 V µs / (12 - 3.3)] / 169 µF = 3.9 mΩ

Based on the above criteria two 150 µF polymer aluminum

capacitors with a ESR = 12 mΩ each for a effective ESR in

parallel of 6 mΩ was chosen from Panasonic. The part num-

ber is EEF-UE0J151P.

5. MOSFET Selection

The LM3151/2/3 are designed to drive N-channel MOSFETs.

For a maximum input voltage of 24V we should choose N-

channel MOSFETs with a maximum drain-source voltage,

VDS, greater than 1.2 x 24V = 28.8V. FETs with maximum

VDS of 30V will be the first option. The combined total gate

charge Qgtotal of the high-side and low-side FET should satisfy

the following:

Qgtotal ≤ IVCCL / fs

Qgtotal ≤ 65 mA / 500 kHz

Qgtotal ≤ 130 nC

Where IVCCL is the minimum current limit of VCC, over the

temperature range, specified in the electrical characteristics

table. The MOSFET gate charge Qg is gathered from reading

the VGS vs Qg curve of the MOSFET datasheet at the VGS =

5V for the high-side, M1, MOSFET and VGS = 6V for the low-

side, M2, MOSFET.

The Renesas MOSFET RJK0305DPB has a gate charge of

10 nC at VGS = 5V, and 12 nC at VGS = 6V. This combined

gate charge for a high-side, M1, and low-side, M2, MOSFET

12 nC + 10 nC = 22 nC is less than 130 nC calculated

Qgtotal.

The calculated MOSFET power dissipation must be less than

the max allowed power dissipation, Pdmax, as specified in the

MOSFET datasheet. An approximate calculation of the FET

power dissipated Pd, of the high-side and low-side FET is

given by:

High-Side MOSFET

The max power dissipation of the RJK0305DPB is rated as

45W for a junction temperature that is 125°C higher than the

case temperature and a thermal resistance from the FET

junction to case, θJC, of 2.78°C/W. When the FET is mounted

onto the PCB, the PCB will have some additional thermal re-

sistance such that the total system thermal resistance of the

FET package and the PCB, θJA, is typically in the range of 30°

C/W for this type of FET package. The max power dissipation,

Pdmax, with the FET mounted onto a PCB with a 125°C junc-

tion temperature rise above ambient temperature and θJA =

30°C/W, can be estimated by:

Pdmax = 125°C / 30°C/W = 4.1W

The system calculated Pdh of 0.674W is much less than the

FET Pdmax of 4.1W and therefore the RJK0305DPB max al-

lowable power dissipation criteria is met.

Low-Side MOSFET

Primary loss is conduction loss given by:

Pdl = Iout2 x RDS(ON) x (1-D) = 122 x 0.01 x (1-0.275) = 1W

Pdl is also less than the Pdmax specified on the RJK0305DPB

MOSFET datasheet.

However, it is not always necessary to use the same MOS-

FET for both the high-side and low-side. For most applications

it is necessary to choose the high-side MOSFET with the low-

est gate charge and the low-side MOSFET is chosen for the

lowest allowed RDS(ON). The plateau voltage of the FET VGS

vs Qg curve must be less than VCC - 750 mV.

The current limit, IOCL, is calculated by estimating the RDS

(ON) of the low-side FET at the maximum junction temperature

of 100°C. Then the following calculation of IOCL is:

IOCL = ICL + ΔIL / 2

ICL = 200 mV / 0.014 = 14.2A

IOCL = 14.2A + 3.6 / 2 = 16A

6. Calculate Input Capacitance

The input capacitor should be chosen so that the voltage rat-

ing is greater than the maximum input voltage which for this

example is 24V. Similar to the output capacitor, the voltage

rating needed will depend on the type of capacitor chosen.

The input capacitor should also be able to handle the input

rms current which is approximately 0.5 x IOUT. For this exam-

ple the rms input current is approximately 0.5 x 12A = 6A.

The minimum capacitance with a maximum 5% input ripple

ΔVIN-MAX = (0.05 x 12) = 0.6V:

CIN = [12 x 0.275 x (1-0.275)] / [500 kHz x 0.6] = 8 µF

To handle the large input rms current 2 ceramic capacitors

are chosen at 10 µF each with a voltage rating of 50V and

case size of 1210, that can handle 3A of rms current each. A

100 µF aluminum electrolytic is chosen to help dampen input

ringing.

CBYP = 0.1 µF ceramic with a voltage rating greater than max-

imum VIN

7. Calculate Soft-Start Capacitor

The soft start-time should be greater than the input voltage

rise time and also satisfy the following equality to maintain a

smooth transition of the output voltage to the programmed

regulation voltage during startup.

tSS ≥ (VOUT x COUT) / (IOCL - IOUT)

5 ms > (3.3V x 300 µF) / (1.2 x 12A - 12A)

5 ms > 0.412 ms

The desired soft-start time, tSS, of 5 ms satisfies the equality

as shown above. Therefore, the soft-start capacitor, CSS, is

calculated as:

CSS = (7.7 µA x 5 ms) / 0.6V = 0.064 µF

15 www.national.com

LM3151/LM3152/LM3153

Let CSS = 0.068 µF, which is the next closest standard value.

This should be a ceramic cap with a voltage rating greater

than 10V.

8. CVCC, CEN, and CBST

CVCC = 1µF ceramic with a voltage rating greater than 10V

CEN = 1000 pF ceramic with a voltage rating greater than 10V

CBST = 0.47 µF ceramic with a voltage rating greater than 10V

Bill of Materials

Designator Value Parameters Manufacturer Part Number

CBST 0.47 µF Ceramic, X7R, 16V, 10% TDK C2012X7R1C474K

CBYP 0.1 µF Ceramic, X7R, 50V, 10% TDK C2012X7R1H104K

CEN 1000 pF Ceramic, X7R, 50V, 10% TDK C1608X7R1H102K

CIN1 100 µF AL, EEV-FK, 63V, 20% Panasonic EEV-FK1J101P

CIN2, CIN3 10 µF Ceramic, X5R, 35V, 10% Taiyo Yuden GMK325BJ106KN-T

COUT1, COUT2 150 µF AL, UE, 6.3V, 20% Panasonic EEF-UE0J151R

CSS 0.068 µF Ceramic, 16V, 10% 0603YC683KAT2A

CVCC 1 µF Ceramic, X7R, 16V, 10% Kemet C0805C105K4RACTU

L1 1.65 µH Shielded Drum Core, A, 2.53 mΩCoilcraft Inc. HA3778-AL

M1, M2 30V 8 nC, RDS(ON) @4.5V = 10 mΩRenesas RJK0305DB

U1 National Semiconductor LM3152MH-3.3

www.national.com 16

LM3151/LM3152/LM3153

PCB Layout Considerations

It is good practice to layout the power components first, such

as the input and output capacitors, FETs, and inductor. The

first priority is to make the loop between the input capacitors

and the source of the low side FET to be very small and tie

the grounds of each directly to each other and then to the

ground plane through vias. As shown in the figure below,

when the input cap ground is tied directly to the source of the

low side FET, parasitic inductance in the power path, along

with noise coupled into the ground plane, are reduced.

The switch node is the next item of importance. The switch

node should be made only as large as required to handle the

load current. There are fast voltage transitions occurring in

the switch node at a high frequency, and if the switch node is

made too large it may act as an antennae and couple switch-

ing noise into other parts of the circuit. For high power designs

it is recommended to use a multi-layer board. The FET’s are

going to be the largest heat generating devices in the design,

and as such, care should be taken to remove the heat. On

multi layer boards using exposed-pad packages for the FET’s

such as the power-pak SO-8, vias should be used under the

FETs to the same plane on the interior layers to help dissipate

the heat and cool the FETs. For the typical single FET Power-

Pak type FETs the high-side FET DAP is Vin. The Vin plane

should be copied to the other interior layers to the bottom layer

for maximum heat dissipation. Likewise, the DAP of the low-

side FET is connected to the SW node and it’s shape should

be duplicated to the interior layers down to the bottom layer

for maximum heat dissipation.

See the Evaluation Board application note AN-1900 for an

example of a typical multilayer board layout, and the Demon-

stration Board Reference Design App Note for a typical 2 layer

board layout. Each design allows for single sided component

mounting.

30053258

FIGURE 5. Schematic of Parasitics

30053280

FIGURE 6. PCB Placement of Power Stage

17 www.national.com

LM3151/LM3152/LM3153

Physical Dimensions inches (millimeters) unless otherwise noted

14-Lead eTSSOP Package

NS Package Number MXA14A

www.national.com 18

LM3151/LM3152/LM3153

Notes

19 www.national.com

LM3151/LM3152/LM3153

Notes

LM3151/LM3152/LM3153 SIMPLE SWITCHER® CONTROLLER, High Input Voltage Synchronous

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