MCP6V27-E/SN - MICROCHIP - Datasheet.Directory

MCP6V26/7/8

Features

• High DC Precision:

-V

OS Drift: ±50 nV/°C (maximum)

-V

OS: ±2 µV (maximum)

-A

OL: 125 dB (minimum)

- PSRR: 125 dB (minimum)

- CMRR: 120 dB (minimum)

-E

ni: 1.0 µVP-P (typical), f = 0.1 Hz to 10 Hz

-E

ni: 0.32 µVP-P (typical), f = 0.01 Hz to 1 Hz

• Low Power and Supply Voltages:

-I

Q: 620 µA/amplifier (typical)

- Wide Supply Voltage Range: 2.3V to 5.5V

•Easy to Use:

- Rail-to-Rail Input/Output

- Gain Bandwidth Product: 2 MHz (typical)

- Unity Gain Stable

- Available in Single and Dual

- Single with Chip Select (CS): MCP6V28

• Extended Temperature Range: -40°C to +125°C

Typical Applications

• Portable Instrumentation

• Sensor Conditioning

• Temperature Measurement

• DC Offset Correction

• Medical Instrumentation

Design Aids

• SPICE Macro Models

•FilterLab

® Software

• Microchip Advanced Part Selector (MAPS)

• Analog Demonstration and Evaluation Boards

• Application Notes

Related Parts

Parts with lower power, lower bandwidth and higher

noise:

• MCP6V01/2/3: Spread clock

• MCP6V06/7/8: Non-spread clock

Description

The Microchip Technology Inc. MCP6V26/7/8 family of

operational amplifiers provides input offset voltage

correction for very low offset and offset drift. These

devices have a wide gain bandwidth product (2 MHz,

typical) and strongly reject switching noise. They are

unity gain stable, have no 1/f noise, and have good

power supply rejection ratio (PSRR) and common

mode rejection ratio (CMRR). These products operate

with a single supply voltage as low as 2.3V, while

drawing 620 µA/amplifier (typical) of quiescent current.

The Microchip Technology Inc. MCP6V26/7/8 op amps

are offered as a single (MCP6V26), single with Chip

Select (CS) (MCP6V28) and dual (MCP6V27). They

were designed using an advanced CMOS process.

Package Types (top view)

VIN+

VIN–

VSS

VDD

VOUT

5NC

NCNC

VINA+

VINA–

VSS

VOUTA VDD

VOUTB

VINB–

VINB+

MCP6V26

MSOP, SOIC

MCP6V27

MSOP, SOIC

VIN+

VIN–

VSS

VDD

VOUT

5NC

MCP6V28

MSOP, SOIC

VINA+

VINA–

VSS

VOUTA VDD

VOUTB

VINB–

VINB+

MCP6V27

4×4 DFN *

* Includes Exposed Thermal Pad (EP); see Ta b l e 3 - 1 .

VIN+

VIN–

VSS

NC NC

VDD

VOUT

MCP6V26

2×3 TDFN *

VIN+

VIN–

VSS

NC CS

VDD

VOUT

MCP6V28

2×3 TDFN *

620 µA, 2 MHz Auto-Zeroed Op Amps

MCP6V26/7/8

Typical Application Circuit

Offset Voltage Correction for Power Driver

10 nF

10 kΩ

10 kΩ10 kΩ

MCP661

VDD/2

500 kΩ

VIN VOUT

10 kΩ

MCP6V26

5kΩ

VDD/2

MCP6V26/7/8

1.0 ELECTRICAL

CHARACTERISTICS

1.1 Absolute Maximum Ratings †

VDD –V

SS ..............................................................................6.5V

Current at Input Pins †† ......................................................±2 mA

Analog Inputs (VIN+ and VIN–) †† .......... VSS – 1.0V to VDD+1.0V

All other Inputs and Outputs .................. VSS – 0.3V to VDD+0.3V

Difference Input voltage ............................................. |VDD –V

SS|

Output Short Circuit Current .......................................Continuous

Current at Output and Supply Pins ...................................±30 mA

Storage Temperature ..........................................-65°C to +150°C

Max. Junction Temperature .............................................. +150°C

ESD protection on all pins (HBM, CDM, MM) ≥4 kV,1.5 kV, 300V

†Notice: Stresses above those listed under “Absolute

Maximum Ratings” may cause permanent damage to

the device. This is a stress rating only and functional

operation of the device at those or any other

conditions above those indicated in the operational

listings of this specification is not implied. Exposure to

maximum rating conditions for extended periods may

affect device reliability.

†† See Section 4.2.1, Rail-to-Rail Inputs.

1.2 Specifications

TABLE 1-1: DC ELECTRICAL SPECIFICATIONS

Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.3V to +5.5V, VSS = GND,

VCM = VDD/3, VOUT =V

DD/2, VL=V

DD/2, RL = 10 kΩ to VL and CS = GND (refer to Figure 1-5 and Figure 1-6).

Parameters Sym Min Typ Max Units Conditions

Input Offset

Input Offset Voltage VOS -2 — +2 µV TA = +25°C (Note 1)

Input Offset Voltage Drift

with Temperature (linear Temp. Co.)

TC1-50 — +50 nV/°C TA = -40 to +125°C

(Note 1)

Input Offset Voltage Quadratic

Temperature Coefficient

TC2—±0.2 —nV/°C

2TA = -40 to +125°C

Power Supply Rejection PSRR 125 142 — dB (Note 1)

Input Bias Current and Impedance

Input Bias Current IB—+7 —pA

Input Bias Current across

Temperature

IB—+110 — pAT

A = +85°C

IB—+1.2 +5 nAT

A = +125°C

Input Offset Current IOS —±70 — pA

Input Offset Current across

Temperature

IOS —±50 — pAT

A = +85°C

IOS —±60 — pAT

A = +125°C

Common Mode Input Impedance ZCM —10

13||12 — Ω||pF

Differential Input Impedance ZDIFF —10

13||12 — Ω||pF

Note 1: Set by design and characterization. Due to thermal junction and other effects in the production

environment, these parts can only be screened in production (except TC1; see Appendix B: “Offset

Related Test Screens”).

2: Figure 2-18 shows how VCML and VCMH changed across temperature for the first production lot.

MCP6V26/7/8

Common Mode

Common-Mode Input

Voltage Range Low

VCML ——V

SS −0.15 V (Note 2)

Common-Mode Input

Voltage Range High

VCMH VDD +0.2 — — V (Note 2)

Common-Mode Rejection CMRR 120 136 — dB VDD = 2.3V,

VCM = -0.15V to 2.5V

(Note 1, Note 2)

CMRR 125 142 — dB VDD = 5.5V,

VCM = -0.15V to 5.7V

(Note 1, Note 2)

Open-Loop Gain

DC Open-Loop Gain (large signal) AOL 125 147 — dB VDD =2.3V,

VOUT = 0.2V to 2.1V

(Note 1)

AOL 133 155 — dB VDD =5.5V,

VOUT = 0.2V to 5.3V

(Note 1)

Output

Minimum Output Voltage Swing VOL —V

SS +5 V

SS +15 mV G = +2, 0.5V

input overdrive

Maximum Output Voltage Swing VOH VDD –15 V

DD −5 — mV G = +2, 0.5V

input overdrive

Output Short Circuit Current ISC —±12 — mAV

DD =2.3V

ISC —±22 — mAV

DD =5.5V

Power Supply

Supply Voltage VDD 2.3 — 5.5 V

Quiescent Current per amplifier IQ450 620 800 µA IO = 0

POR Trip Voltage VPOR 1.15 — 1.65 V

TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED)

Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.3V to +5.5V, VSS = GND,

VCM = VDD/3, VOUT =V

DD/2, VL=V

DD/2, RL = 10 kΩ to VL and CS = GND (refer to Figure 1-5 and Figure 1-6).

Parameters Sym Min Typ Max Units Conditions

Note 1: Set by design and characterization. Due to thermal junction and other effects in the production

environment, these parts can only be screened in production (except TC1; see Appendix B: “Offset

Related Test Screens”).

2: Figure 2-18 shows how VCML and VCMH changed across temperature for the first production lot.

MCP6V26/7/8

TABLE 1-2: AC ELECTRICAL SPECIFICATIONS

Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.3V to +5.5V, VSS = GND,

VCM = VDD/3, VOUT =V

DD/2, VL=V

DD/2, RL = 10 kΩ to VL, CL = 60 pF and CS = GND (refer to Figure 1-5 and

Figure 1-6).

Parameters Sym Min Typ Max Units Conditions

Amplifier AC Response

Gain Bandwidth Product GBWP — 2.0 — MHz

Slew Rate SR — 1.0 — V/µs

Phase Margin PM — 65 — ° G = +1

Amplifier Noise Response

Input Noise Voltage Eni —0.32 — µV

P-P f = 0.01 Hz to 1 Hz

Eni —1.0 — µV

P-P f = 0.1 Hz to 10 Hz

Input Noise Voltage Density eni —50 — nV/√Hz f < 5 kHz

eni —29 — nV/√Hz f = 100 kHz

Input Noise Current Density ini —0.6 — fA/√Hz

Amplifier Distortion (Note 1)

Intermodulation Distortion (AC) IMD — 40 — µVPK VCM tone = 50 mVPK at 1 kHz,

GN = 1

Amplifier Step Response

Start Up Time tSTR — 75 — µs G = +1, VOS within 50 µV of its final value

(Note 2)

Offset Correction Settling Time tSTL — 150 — µs G = +1, VIN step of 2V,

VOS within 50 µV of its final value

Output Overdrive Recovery Time tODR — 45 — µs G = -100, ±0.5V input overdrive to VDD/2,

VIN 50% point to VOUT 90% point

(Note 3)

Note 1: These parameters were characterized using the circuit in Figure 1-7. In Figure 2-37 and Figure 2-38, there

is an IMD tone at DC, a residual tone at 1 kHz, other IMD tones and clock tones.

2: High gains behave differently; see Section 4.3.3, Offset at Power Up.

3: tODR includes some uncertainty due to clock edge timing.

MCP6V26/7/8

TABLE 1-4: TEMPERATURE SPECIFICATIONS

TABLE 1-3: DIGITAL ELECTRICAL SPECIFICATIONS

Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.3V to +5.5V, VSS = GND,

VCM = VDD/3, VOUT =V

DD/2, VL=V

DD/2, RL = 10 kW to VL, CL = 60 pF, and CS = GND (refer to Figure 1-5 and

Figure 1-6).

Parameters Sym Min Typ Max Units Conditions

CS Pull-Down Resistor (MCP6V28)

CS Pull-Down Resistor RPD 35—MΩ

CS Low Specifications (MCP6V28)

CS Logic Threshold, Low VIL VSS —0.3V

DD V

CS Input Current, Low ICSL —5—pA

CS = VSS

CS High Specifications (MCP6V28)

CS Logic Threshold, High VIH 0.7VDD —V

DD V

CS Input Current, High ICSH —V

DD/RPD —pA

CS = VDD

CS Input High,

GND Current per amplifier

ISS —-0.4—µA

CS = VDD, VDD = 2.3V

ISS —-1—µA

CS = VDD, VDD = 5.5V

Amplifier Output Leakage,

CS High

IO_LEAK —20—pA

CS = VDD

CS Dynamic Specifications (MCP6V28)

CS Low to Amplifier Output On

Turn-on Time

tON —450µs

CS Low = VSS+0.3 V, G = +1 V/V,

VOUT = 0.9 VDD/2

CS High to Amplifier Output

High-Z

tOFF —1—µs

CS High = VDD – 0.3 V, G = +1 V/V,

VOUT = 0.1 VDD/2

Internal Hysteresis VHYST —0.2—V

Electrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = +2.3V to +5.5V, VSS = GND.

Parameters Sym Min Typ Max Units Conditions

Temperature Ranges

Specified Temperature Range TA-40 — +125 °C

Operating Temperature Range TA-40 — +125 °C (Note 1)

Storage Temperature Range TA-65 — +150 °C

Thermal Package Resistances

Thermal Resistance, 8L-4x4 DFN θJA —48—°C/W(Note 2)

Thermal Resistance, 8L-MSOP θJA —211—°C/W

Thermal Resistance, 8L-SOIC θJA — 150 — °C/W

Thermal Resistance, 8L-2x3 TDFN θJA —53—°C/W(Note 2)

Note 1: Operation must not cause TJ to exceed Maximum Junction Temperature specification (+150°C).

2: Measured on a standard JC51-7, four layer printed circuit board with ground plane and vias.

MCP6V26/7/8

1.3 Timing Diagrams

FIGURE 1-1: Amplifier Start Up.

FIGURE 1-2: Offset Correction Settling

Time.

FIGURE 1-3: Output Overdrive Recove ry.

FIGURE 1-4: Chip Select (MCP6V28).

1.4 Test Circuits

The circuits used for the DC and AC tests are shown in

Figure 1-5 and Figure 1-6. Lay the bypass capacitors

out as discussed in Section 4.3.10, Supply Bypass-

ing and Filtering. RN is equal to the parallel combina-

tion of RF and RG to minimize bias current effects.

FIGURE 1-5: AC and DC Test Circuit for

Most Non-Inv erti ng Ga in Cond iti ons .

FIGURE 1-6: AC and DC Test Circuit for

Most Inverting Gain Conditions.

The circuit in Figure 1-7 tests the op amp input’s

dynamic behavior (i.e., IMD, tSTR, tSTL and tODR). The

potentiometer balances the resistor network (VOUT

should equal VREF at DC). The op amp’s common

mode input voltage is VCM =V

IN/2. The error at the

input (VERR) appears at VOUT with a noise gain of

10 V/V.

FIGURE 1-7: Test Circuit for Dynamic

Input Behavior.

VDD

VOS

VOS +50µV

VOS –50µV

tSTR

2.3V to 5.5V

2.3V

VIN

VOS

VOS +50µV

tSTL

VIN

VOUT

VDD

VSS

tODR

VDD/2

VIL

High-Z

tON

VIH

tOFF

VOUT

-2 µA

High-Z

ISS -2 µA

300 µA

1µA

IDD

1µA

300 µA

VDD/5 MΩ

ICS VDD/5 MΩ

5pA

(typical) (typical)

(typical)

VDD

RGRF

VOUT

VIN

VDD/3

1µF

CLRL

100 nF

RISO

MCP6V2X

VDD

RGRF

VOUT

VDD/3

VIN

1µF

CLRL

100 nF

RISO

MCP6V2X

VDD

VOUT

1µF

CLRL

100 nF

RISO

20.0 kΩ24.9 Ω

20.0 kΩ50Ω

VIN

VREF

0.1%

0.1% 25 turn

20.0 kΩ

0.1%

2.49 kΩ2.49 kΩ

MCP6V2X

MCP6V26/7/8

2.0 TYPICAL PERFORMANCE CURVES

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

2.1 DC Input Precision

FIGURE 2-1: Input Offset Voltage.

FIGURE 2-2: Input Offset Voltage Drift.

FIGURE 2-3: Input Offset Voltage

Quadratic Temperature Coefficient.

FIGURE 2-4: Input Offset Voltage vs.

Power Supply Voltage with VCM =V

CML.

FIGURE 2-5: Input Offset Voltage vs.

Power Supply Voltage with VCM =V

CMH.

FIGURE 2-6: Input Offset Voltage vs.

Output Voltage.

Note: The graphs and tables provided following this note are a statistical summary based on a limited number of

samples and are provided for informational purposes only. The performance characteristics listed herein

are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified

operating range (e.g., outside specified power supply range) and therefore outside the warranted range.

10%

15%

20%

25%

30%

35%

40%

-2.0

-1.0

0.0

1.0

2.0

Input Offset Voltage (µV)

Percentage of Occurrences

20 Samples

TA = +25°C

VDD = 2.3V and 5.5V

10%

15%

20%

25%

30%

-50

-40

-30

-20

-10

Input Offset Voltage Drift; TC1 (nV/°C)

Percentage of Occurrences

20 Samples

VDD = 2.3V and 5.5

-5

-4

-3

-2

-1

0.00.51.01.52.02.53.03.54.04.55.05.5

Output Voltage (V)

Input Offset Voltage (µV)

VDD = 2.3

VDD = 5.5V

Representative Par

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

FIGURE 2-7: Input Offset Voltage vs.

Common Mode Voltage with VDD =2.3V.

FIGURE 2-8: Input Offset Voltage vs.

Common Mode Voltage with VDD =5.5V.

FIGURE 2-9: CMRR.

FIGURE 2-10: PSRR.

FIGURE 2-11: DC Open-Loop Gain.

FIGURE 2-12: CMRR and PSRR vs.

Ambient Temper atu re .

-5

-4

-3

-2

-1

-0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0

Input Common Mode Voltage (V)

Input Offset Voltage (µV)

VDD = 2.3V

Representative Part

-40°C

+25°C

+85°C

+125°C

-5

-4

-3

-2

-1

-0.5

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Input Common Mode Voltage (V)

Input Offset Voltage (µV)

VDD = 5.5V

Representative Part

-40°C

+25°C

+85°C

+125°C

10%

15%

20%

25%

30%

35%

-0.5

-0.3

0.0

0.3

0.5

1/CMRR (µV/V)

Percentage of Occurrences

20 Samples

TA = +25°C

VDD = 2.3V

VDD = 5.5V

10%

15%

20%

25%

30%

-0.4

-0.3

-0.2

-0.1

0.0

0.1

0.2

0.3

0.4

1/PSRR (µV/V)

Percentage of Occurrences

20 Samples

TA = +25°C

10%

20%

30%

40%

50%

60%

70%

80%

90%

100%

-0.4

-0.3

-0.2

-0.1

0.0

0.1

0.2

0.3

0.4

1/AOL (µV/V)

Percentage of Occurrences

20 Samples

TA = +25°C

VDD = 2.3

VDD = 5.5V

120

125

130

135

140

145

150

155

160

-50 -25 0 25 50 75 100 125

Ambient Temperature (°C)

CMRR, PSRR (dB)

PSRR

CMRR

VDD = 5.5

VDD = 2.3

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

FIGURE 2-13: DC Open-Loop Gain vs.

Ambient Temperature.

FIGURE 2-14: Input Bias and Offset

Currents vs. Common Mode Input Voltage with

TA= +85°C.

FIGURE 2-15: Input Bias and Offset

Currents vs. Common Mode Input Voltage with

TA= +125°C.

FIGURE 2-16: Input Bias and Offset

Currents vs. Ambient Temperature with

VDD =+5.5V.

FIGURE 2-17: Input Bias Current vs. Input

Voltage (below VSS).

120

125

130

135

140

145

150

155

160

-50-25 0 255075100125

Ambient Temperature (°C)

DC Open-Loop Gain (dB)

VDD = 5.5V

VDD = 2.3V

-100

-50

100

150

200

-0.5

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Common Mode Input Voltage (V)

Input Bias, Offset Currents (pA)

TA = +85°C

VDD = 5.5V

IOS

-400

-200

200

400

600

800

1000

1200

1400

1600

1800

2000

-0.5

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Common Mode Input Voltage (V)

Input Bias, Offset Currents (pA)

TA = +125°C

VDD = 5.5V

IOS

100

1,000

10,000

25 35 45 55 65 75 85 95 105 115 125

Ambient Temperature (°C)

Input Bias, Offset Currents (A)

VDD = 5.5

-IOS

10p

100p

10n

1.E-11

1.E-10

1.E-09

1.E-08

1.E-07

1.E-06

1.E-05

1.E-04

1.E-03

1.E-02

-1.0 -0.9 -0.8 -0.7 -0.6 -0.5 -0.4 -0.3 -0.2 -0.1 0.0

Input Voltage (V)

Input Current Magnitude (A)

+125°C

+85°C

+25°C

-40°C

10m

100µ

10µ

1µ

100n

10n

100p

10p

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

2.2 Other DC Voltages and Currents

FIGURE 2-18: Input Common Mode

Voltage Headroom (Range) vs. Ambient

Temperature.

FIGURE 2-19: Output Voltage Headroom

vs. Output Current.

FIGURE 2-20: Output Voltage Headroom

vs. Ambient Temperature.

FIGURE 2-21: Output Short Circuit Current

vs. Power Supply Voltage.

FIGURE 2-22: Supply Current vs. Power

Supply Voltage.

FIGURE 2-23: Power On Reset Trip

Voltage.

-0.4

-0.3

-0.2

-0.1

0.0

0.1

0.2

0.3

0.4

-50 -25 0 25 50 75 100 125

Ambient Temperature (°C)

Input Common Mode Voltage

Headroom (V)

Lower (VCML – VSS)

Upper ( VCMH – VDD)

1 Wafer Lo

100

1000

0.1 1 10

Output Current Magnitude (mA)

Output Voltage Headroom (mV)

VDD – VOH

VDD = 5.5V

VDD = 2.3V

VOL – VSS

-50-25 0 255075100125

Ambient Temperature (°C)

Output Headroom (mV)

VDD – VOH

VDD = 5.5

VOL – VSS

VDD = 2.3V

RL = 10 kΩ

-40

-30

-20

-10

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Power Supply Voltage (V)

Output Short Circuit Current

(mA)

-40°C

+25°C

+85°C

+125°C

+85°C

+25°C

-40°C

100

200

300

400

500

600

700

800

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Power Supply Voltage (V)

Supply Current (µA/amplifier)

+125°C

+85°C

+25°C

-40°C

10%

15%

20%

25%

30%

35%

40%

1.25

1.26

1.27

1.28

1.29

1.30

1.31

1.32

1.33

1.34

1.35

POR Trip Voltage (V)

Percentage of Occurrences

820 Samples

1 Wafer Lot

TA = +25°C

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

FIGURE 2-24: Power On Reset V oltage vs.

Ambient Temperature.

0.0

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

1.8

-50 -25 0 25 50 75 100 125

Ambient Temperature (°C)

POR Trip Voltage (V)

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

2.3 Frequency Response

FIGURE 2-25: CMRR and PSRR vs.

Frequency.

FIGURE 2-26: Open-Loop Gain vs.

Frequency with VDD =2.3V.

FIGURE 2-27: Open-Loop Gain vs.

Frequency with VDD =5.5V.

FIGURE 2-28: Gain Bandwidth Product

and Phase Margin vs. Ambient Temperature.

FIGURE 2-29: Gain Bandwidth Product

and Phase Margin vs. Common Mode Input

Voltage.

FIGURE 2-30: Gain Bandwidth Product

and Phase Margin vs. Output Voltage.

100

110

1.E+02 1.E+03 1.E+04 1.E+05 1.E+06

Frequency (Hz)

CMRR, PSRR (dB)

CMR

PSRR+

PSRR-

100 100k

1k 1M10k

-20

-10

1.E+03 1.E+04 1.E+05 1.E+06 1.E+07

Frequency (Hz)

Open-Loop Gain (dB)

-270

-240

-210

-180

-150

-120

-90

-60

-30

Open-Loop Phase (°)

| AOL |

∠AOL

1k 10k 100k 1M 10M

VDD = 2.3V

CL = 60 pF

-20

-10

1.E+03 1.E+04 1.E+05 1.E+06 1.E+07

Frequency (Hz)

Open-Loop Gain (dB)

-270

-240

-210

-180

-150

-120

-90

-60

-30

Open-Loop Phase (°)

| AOL |

∠AOL

1k 10k 100k 1M 10M

VDD = 5.5V

CL = 60 pF

1003.0

902.5

)

ct (M

= 5.5V

GBWP

2.0

rgin (

Prod

1.0

ase M

width

= 2.3V

500.5

in Ban

400.0

100

125

100

125

Ambient Temperature (°C)

1204.0

100

110

3.0

3.5

°)

ct (M

2.5

argin (

Prod

VDD = 5.5V

VDD = 2.3V GBWP

1.5

ase M

dwidt

0.5

1.0

in Ba

400.0

Common Mode Input Voltage (V)

1204.0

100

110

3.0

3.5

)

ct (M

2.5

rgin (

Prod

VDD = 5.5V

VDD = 2.3V PM

1.5

ase M

width

0.5

1.0

n Ban

GBWP

0.0

0.5

Output Voltage (V)

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

FIGURE 2-31: Closed-Loop Output

Impedance vs. Frequency with VDD =2.3V.

FIGURE 2-32: Closed-Loop Output

Impedance vs. Frequency with VDD =5.5V.

FIGURE 2-33: Channel-to-Channel

Separation vs . Frequ enc y.

FIGURE 2-34: Maximum Output Voltage

Swing vs. Frequency.

1.E+00

1.E+01

1.E+02

1.E+03

1.E+04

1.0E+05 1.0E+06 1.0E+07 1.0E+08

Frequency (Hz)

Closed-Loop

Output Impedance (Ω)

VDD = 2.3

100k 1M 10M 100M

100

10k

G = 1 V/V

G = 11 V/V

G = 101 V/V

1.E+00

1.E+01

1.E+02

1.E+03

1.E+04

1.0E+05 1.0E+06 1.0E+07 1.0E+08

Frequency (Hz)

VDD = 2.3V

100k 1M 10M 100M

100

10k

G = 1 V/V

G = 11 V/V

G = 101 V/V 0.1

1.E+03 1.E+04 1.E+05 1.E+06

Frequency (Hz)

Maximum Output Voltage

Swing (VP-P)

VDD = 5.5V

VDD = 2.3V

1k 10k 100k 1M

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

2.4 Input Noise and Distortion

FIGURE 2-35: Input Noise Voltage Density

and Integrated Input Noise Voltage vs.

Frequency.

FIGURE 2-36: Input Noise Voltage Density

vs. Input Common Mode Voltage.

FIGURE 2-37: Int ermodulation Distortion

vs. Frequency with VCM Dis tur ba nce (se e

Figure 1-7).

FIGURE 2-38: Intermodulation Distortion

vs. Frequency with VDD Disturbance (see

Figure 1-7).

FIGURE 2-39: Input Noise vs. Time with

1 Hz and 10 Hz Filters and VDD =2.3V.

FIGURE 2-40: Input Noise vs. Time with

1 Hz and 10 Hz Filters and VDD =5.5V.

1,00010,000

;

ltage;

ensity

;

= 5.5V

= 2.3V

1001,000

ise V

-P

)

tage

Hz)

100

put N

(μV

ise Vo

(nV

100

ated I

ut No

110

1E 01

1E 02

1E 03

1E 04

1E 05

Integ

10k

100k

(0 Hz to f)

100

Fre

uenc

(

)

10k

100k

100

-0.5

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Common Mode Input Voltage (V)

VDD = 5.5V

VDD = 2.3V

Input Noise Voltage Density

(nV/Hz)

f < 5 kHz

0.1

100

1.E+02 1.E+03 1.E+04 1.E+05

Frequency (Hz)

IMD Spectrum, RTI (µVPK)

GDM = 1 V/V

VCM tone = 50 mVPK, f = 1 kHz

100 1k 10k 100k

IMD tone at DC

residual 1 kHz tone

VDD = 2.3

VDD = 5.5

0.1

100

1.E+02 1.E+03 1.E+04 1.E+05

Frequency (Hz)

IMD Spectrum, RTI (µVPK)

100 1k 10k 100k

GDM = 1 V/V

VDD tone = 50 mVP-P, f = 1 kHz

IMD tone at DC

1 kHz tone

VDD = 5.5V

VDD = 2.3V

0 102030405060708090100

t (s)

Input Noise Voltage; eni(t)

(0.2 µV/div)

VDD = 2.3V

NPBW = 10 Hz

NPBW = 1 Hz

0 102030405060708090100

t (s)

Input Noise Voltage; eni (t)

(0.2 µV/div)

VDD = 5.5

NPBW = 10 Hz

NPBW = 1 Hz

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

2.5 Time Response

FIGURE 2-41: Input Offset Voltage vs.

Time with Temperature Change.

FIGURE 2-42: Input Offset Voltage vs.

Time at Power Up.

FIGURE 2-43: The MCP6V26/7/8 Device

Shows No Input Phase Reversal with Overdrive.

FIGURE 2-44: Non-inverting Small Signal

Step Response.

FIGURE 2-45: Non-inverting Large Signal

Step Response.

FIGURE 2-46: Inverting Small Signal Step

Response.

-14

-12

-10

-8

-6

-4

-2

0 20 40 60 80 100 120 140 160 180

Time (s)

Input Offset Voltage (µV)

100

PCB Temperature (°C)

TPCB

VOS

Temperature increased by

using heat gun for 10 seconds.

-10

0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0

Time (ms)

Input Offset Voltage (µV)

-4

-3

-2

-1

Power Supply Voltage (V)

POR Tri

Point

VOS

VDD

G = 1

-1

012345678910

Time (ms)

Input, Output Voltages (V)

VDD = 5.5V

G = 1

VOUT

VIN

012345678910

Time (µs)

Output Voltage (10 mV/div)

VDD = 5.5V

G = 1

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

0 5 10 15 20 25 30 35 40 45 50

Time (µs)

Output Voltage (V)

VDD = 5.5V

G = 1

012345678910

Time (µs)

Output Voltage (10 mV/div)

VDD = 5.5V

G = -1

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF and CS = GND.

FIGURE 2-47: Inverting Large Signal Step

Response.

FIGURE 2-48: Slew Rate vs. Ambient

Temperature.

FIGURE 2-49: Output Overdrive Recovery

vs. Time with G = -100 V/V.

FIGURE 2-50: Output Overdrive Recovery

Time vs. Inverting Gain.

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

0 5 10 15 20 25 30 35 40 45 50

Time (µs)

Output Voltage (V)

VDD = 5.5V

G = -1

0.0

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

-50-25 0 255075100125

Ambient Temperature (°C)

Slew Rate (V/µs)

Falling Edge

VDD = 5.5V

VDD = 2.3V

Rising Edge

-1.0

0.0

1.0

2.0

3.0

4.0

5.0

6.0

Time (50 µs/div)

Output Voltage (V)

-1

Input Voltage × G (V/V)

VDD = 5.5V

G = -100 V/V

0.5V Overdrive

VOU

G VIN

VOU

G VIN

100

1000

1 10 100 1000

Inverting Gain Magnitude (V/V)

Overdrive Recovery Time (µs)

0.5V Output Overdrive

tODR, lo

tODR, high

VDD = 2.3V

VDD = 5.5V

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF, and CS = GND.

2.6 Chip Select Response (MCP6V28 only)

FIGURE 2-51: Chip Select Current vs.

Power Supply Voltage.

FIGURE 2-52: Power Supply Current vs.

Chip Select Voltage with VDD =2.3V.

FIGURE 2-53: Power Supply Current vs.

Chip Select Voltage with VDD =5.5V.

FIGURE 2-54: Chip Select Current vs. Chip

Select V oltage.

FIGURE 2-55: Chip Select Voltage, Output

Voltage vs. Time with VDD =2.3V.

FIGURE 2-56: Chip Select Voltage, Output

Voltage vs. Time with VDD =5.5V.

0.0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.0

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Chip Select Current (μA)

Power Supply Voltage (V)

CS = VDD

100

200

300

400

500

600

700

0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4

Power Supply Current (μA)

Chip Select Voltage (V)

VDD = 2.3V

G= 1

VIN = 1.15V

VL= 0V

Op Amp

turns on

here

Op Amp

turns off

here

Hysteresis

100

200

300

400

500

600

700

800

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Power Supply Current (μA)

Chip Select Voltage (V)

VDD = 5.5V

G= 1

VIN = 2.75V

VL= 0V

Op Amp

turns on

here

Op Amp

turns off

here

Hysteresis

0.0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.0

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Chip Select Current (μA)

Chip Select Voltage (V)

VDD = 5.5V

1.0

1.5

2.0

2.5

put Votage (V)

VOUT On

VOUT Off

0.0

0.5

0 5 10 15 20 25 30 35 40 45 50

Time (5 μs/div)

VDD = 2.3V

G = +1 V/V

VIN = VDD

RL= 10 k tied to VDD/2

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

tput Votage (V)

VOUT On

VOUT Off

55V

0.0

0.5

1.0

1.5

0 5 10 15 20 25 30 35 40 45 50

Time (5 μs/div)

DD =

G = +1 V/V

VIN = VDD

RL= 10 k tied to VDD/2

MCP6V26/7/8

Note: Unless otherwise indicated, TA=+25°C, V

DD = +2.3V to 5.5V, VSS = GND, VCM =V

DD/3, VOUT =V

DD/2,

VL=V

DD/2, RL=10kΩ to VL, CL = 60 pF, and CS = GND.

FIGURE 2-57: Chip Select Relative Logic

Thresholds vs. Ambient Temperature.

FIGURE 2-58: Chip Select Hysteresis.

FIGURE 2-59: Chip Select Turn On Time

vs. Ambient Temperature.

FIGURE 2-60: Chip Select’s Pull-down

Resistor (RPD) vs. Ambient Temperature.

FIGURE 2-61: Quiescent Current in

Shutdown vs. Power Supply Voltage.

30%

35%

40%

45%

50%

55%

60%

65%

70%

-50 -25 0 25 50 75 100 125

Relative Chip Select Logic Levels;

Low and High (V/V)

Ambient Temperature (°C)

VIH/VDD

VIL/VDD

VDD = 5.5V

VDD = 2.3V

0.00

0.05

0.10

0.15

0.20

0.25

0.30

0.35

0.40

-50 -25 0 25 50 75 100 125

Chip Select Hysteresis (V)

Ambient Temperature (°C)

VDD = 5.5V

VDD = 2.3V

-50 -25 0 25 50 75 100 125

Chip Select Turn On Time (μs)

Ambient Temperature (°C)

VDD = 5.5V

VDD = 2.3V

-50 -25 0 25 50 75 100 125

Pull-down Resistor (M)

Ambient Temperature (°C)

0.0

0.2

0.4

0.6

0.8

1.0

1.2

1.4

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Power Supply Current (μA)

Power Supply Voltage (V)

CS = VDD

Representative Part

+125°C

+85°C

+25°C

-40°C

MCP6V26/7/8

3.0 PIN DESCRIPTIONS

Descriptions of the pins are listed in Table 3-1.

TABLE 3-1: PIN FUNCTION TABLE

3.1 Analog Outputs

The analog output pins (VOUT) are low-impedance

voltage sources.

3.2 Analog Inputs

The non-inverting and inverting inputs (VIN+, VIN–, …)

are high-impedance CMOS inputs with low bias

currents.

3.3 Power Supply Pins

The positive power supply (VDD) is 2.3V to 5.5V higher

than the negative power supply (VSS). For normal

operation, the other pins are between VSS and VDD.

Typically, these parts are used in a single (positive)

supply configuration. In this case, VSS is connected to

ground and VDD is connected to the supply. VDD will

need bypass capacitors.

3.4 Chip Select (CS) Digital Input

This pin (CS) is a CMOS, Schmitt-triggered input that

places the MCP6V28 op amp into a low power mode of

operation.

3.5 Exposed Thermal Pad (EP)

There is an internal connection between the Exposed

Thermal Pad (EP) and the VSS pin; they must be

connected to the same potential on the Printed Circuit

Board (PCB).

This pad can be connected to a PCB ground plane to

provide a larger heat sink. This improves the package

thermal resistance (θJA).

MCP6V26 MCP6V27 MCP6V28

Symbol Description

TDFN MSOP, SOIC DFN MSOP, SOIC TDFN MSOP, SOIC

661166V

OUT

, VOUTA Output (op amp A)

222222V

IN–, VINA– Inverting Input (op amp A)

333333V

IN+, VINA+ Non-inverting Input (op amp A)

444444 V

SS Negative Power Supply

—— 5 5 —— V

INB+ Non-inverting Input (op amp B)

—— 6 6 —— V

INB– Inverting Input (op amp B)

—— 7 7 —— V

OUTB Output (op amp B)

778877 V

DD Positive Power Supply

———— 8 8 CS Chip Select (op amp A)

1, 5, 8 1, 5, 8 — — 1, 5 1, 5 NC No Internal Connection

9 — 9 — 9 — EP Exposed Thermal Pad (EP);

must be connected to VSS

MCP6V26/7/8

4.0 APPLICATIONS

The MCP6V26/7/8 family of auto-zeroed op amps are

manufactured using Microchip’s state-of-the-art CMOS

process. This family is designed for low cost, low power

and high precision applications. Its low supply voltage,

low quiescent current and wide bandwidth make the

MCP6V26/7/8 devices ideal for battery-powered appli-

cations.

4.1 Overview of Auto-Zeroing

Operation

Figure 4-1 shows a simplified diagram of the

MCP6V26/7/8 auto-zeroed op amps. This will be used

to explain how the DC voltage errors are reduced in this

architecture.

FIGURE 4-1: Simplified Auto-Zeroed Op Amp Functional Diagram.

4.1.1 BUILDING BLOCKS

The Null Amplifier and Main Amplifier are designed for

high gain and accuracy using a differential topology.

They have a main input pair (+ and - pins at their top

left) used for the signal. They have an auxiliary input

pair (+ and - pins at their bottom left) used for correcting

the offset voltages. Both input pairs are added together

internally. The capacitors at the auxiliary inputs (CFW

and CH) hold the corrected values during normal

operation.

The Output Buffer is designed to drive external loads at

the VOUT pin. It also produces a single-ended output

voltage (VREF is an internal reference voltage).

All of these switches are make-before-break in order to

minimize glitch-induced errors. They are driven by two

clock phases (φ1 and φ2) that select between normal

mode and auto-zeroing mode.

The clock is derived from an internal R-C oscillator

running at a rate of fOSC1 = 850 kHz. The oscillator’s

output is divided down to the desired rate.

The internal POR ensures the part starts up in a known

good state. It also provides protection against power

supply brown-out events.

The Digital Control circuitry takes care of all of the

housekeeping details of the switching operation. It also

takes care of POR events.

VIN+

VIN–Main

Output VOUT

VREF

Amp.

Buffer

Null

Amp.

Null

Input φ1

Switches

Null

Correct

φ2

Switches

Null

Output

Switches

CFW

POR

Digital

Control Oscillator

φ1

φ2

MCP6V26/7/8

4.1.2 AUTO-ZEROING ACTION

Figure 4-2 shows the connections between amplifiers

during the Normal Mode of operation (φ1). The hold

capacitor (CH) corrects the Null Amplifier’s input offset.

Since the Null Amplifier has very high gain, it

dominates the signal seen by the Main Amplifier. This

greatly reduces the impact of the Main Amplifier’s input

offset voltage on overall performance. Essentially, the

Null Amplifier and Main Amplifier behave as a regular

op amp with very high gain (AOL) and very low offset

voltage (VOS).

FIGURE 4-2: Normal Mode of Operation (

1); Equivalent Amplifier Diagram.

Figure 4-3 shows the connections between amplifiers

during the Auto-zeroing Mode of operation (φ2). The

signal goes directly through the Main Amplifier, and the

flywheel capacitor (CFW) maintains a constant correc-

tion on the Main Amplifier’s offset.

The Null Amplifier uses its own high open loop gain to

drive the voltage across CH to the point where its input

offset voltage is almost zero. Because the signal input

pair is connected to VIN+, the auto-zeroing action

corrects the offset at the current common mode input

voltage (VCM) and supply voltage (VDD). This makes

the DC CMRR and PSRR very high also.

Since these corrections happen every 40 µs, or so, we

also minimize slow errors, including offset drift with

temperature (ΔVOS/ΔTA), 1/f noise, and input offset

aging.

FIGURE 4-3: Auto-zeroing Mode of Operation (

2); Equivalent Diagram.

4.1.3 INTERMODULATION DISTORTION

(IMD)

The MCP6V26/7/8 op amps will show intermodulation

distortion (IMD), products when an AC signal is

present.

The signal and clock can be decomposed into sine

wave tones (Fourier series components). These tones

interact with the auto-zeroing circuitry’s non-linear

response to produce IMD tones at sum and difference

frequencies. Each of the square wave clock’s

harmonics has a series of IMD tones centered on it.

See Figure 2-37 and Figure 2-38.

VIN+

VIN–Main

Output VOUT

VREF

Amp.

Buffer

Null

Amp.

CFW

VIN+

VIN–Main

Output VOUT

VREF

Amp.

Buffer

Null

Amp.

CFW

MCP6V26/7/8

4.2 Other Functional Blocks

4.2.1 RAIL-TO-RAIL INPUTS

The input stage of the MCP6V26/7/8 op amps use two

differential CMOS input stages in parallel. One

operates at low common mode input voltage (VCM,

which is approximately equal to VIN+ and VIN– in

normal operation) and the other at high VCM. With this

topology, the input operates with VCM up to VDD +0.2V,

and down to VSS – 0.15V, at +25°C (see Figure 2-18).

The input offset voltage (VOS) is measured at

VCM =V

SS – 0.15V and VDD + 0.2V to ensure proper

operation.

The transition between the input stages occurs when

VCM ≈VDD –1.2V (see Figure 2-7 and Figure 2-8). For

the best distortion and gain linearity, with non-inverting

gains, avoid this region of operation.

4.2.1.1 Phase Reversal

The input devices are designed to not exhibit phase

inversion when the input pins exceed the supply

voltages. Figure 2-43 shows an input voltage

exceeding both supplies with no phase inversion.

4.2.1.2 Input Voltage Limits

In order to prevent damage and/or improper operation

of these amplifiers, the circuit must limit the voltages at

the input pins (see Section 1.1, Absolute Maximum

Ratings †). This requirement is independent of the

current limits discussed later on.

The ESD protection on the inputs can be depicted as

shown in Figure 4-4. This structure was chosen to

protect the input transistors against many (but not all)

over-voltage conditions, and to minimize input bias

current (IB).

FIGURE 4-4: Simplified Analog Input ESD

Structures.

The input ESD diodes clamp the inputs when they try

to go more than one diode drop below VSS. They also

clamp any voltages that are well above VDD; their

breakdown voltage is high enough to allow normal

operation, but not low enough to protect against slow

over-voltage (beyond VDD) events. Very fast ESD

events (that meet the spec) are limited so that damage

does not occur.

In some applications, it may be necessary to prevent

excessive voltages from reaching the op amp inputs;

Figure 4-5 shows one approach to protecting these

inputs. D1 and D2 may be small signal silicon diodes,

Schottky diodes for lower clamping voltages or diode-

connected FETs for low leakage.

FIGURE 4-5: Protecting the Analog Inputs

Against High Voltages.

4.2.1.3 Input Current Limits

In order to prevent damage and/or improper operation

of these amplifiers, the circuit must limit the currents

into the input pins (see Section 1.1, Absolute

Maximum Ratings †). This requirement is

independent of the voltage limits previously discussed.

Figure 4-6 shows one approach to protecting these

inputs. The resistors R1 and R2 limit the possible

current in or out of the input pins (and into D1 and D2).

The diode currents will dump onto VDD.

FIGURE 4-6: Protecting the Analog Inputs

Against High Currents.

Bond

Pad

Bond

Pad

Bond

Pad

VDD

VIN+

VSS

Input

Stage

Bond

Pad VIN–

VDD

VOUT

MCP6V2X

VDD

min(R1,R

2)>VSS –min(V

1,V

2mA

VOUT

min(R1,R

2)>max(V1,V

2)–V

2mA

MCP6V2X

MCP6V26/7/8

It is also possible to connect the diodes to the left of

resistors R1 and R2. In this case, the currents through

diodes D1 and D2 need to be limited by some other

mechanism. The resistors then serve as in-rush current

limiters; the DC current into the input pins (VIN+ and

VIN–) should be very small.

A significant amount of current can flow out of the

inputs (through the ESD diodes) when the common

mode voltage (VCM) is below ground (VSS); see

Figure 2-17.

4.2.2 RAIL-TO-RAIL OUTPUT

The output voltage range of the MCP6V26/7/8

zero-drift op amps is VDD – 15 mV (minimum) and

VSS + 15 mV (maximum) when RL=10kΩ is

connected to VDD/2 and VDD = 5.5V. Refer to

Figure 2-19 and Figure 2-20.

This op amp is designed to drive light loads; use

another amplifier to buffer the output from heavy loads.

4.2.3 CHIP SELECT (CS)

The single MCP6V28 has a Chip Select (CS) pin.

When CS is pulled high, the supply current for the

corresponding op amp drops to about 1 µA (typical),

and is pulled through the CS pin to VSS. When this

happens, the amplifier is put into a high impedance

state. By pulling CS low, the amplifier is enabled. If the

CS pin is left floating, the internal pull-down resistor

(about 5 MΩ) will keep the part on. Figure 1-4 shows

the output voltage and supply current response to a CS

pulse.

4.3 Application Tips

4.3.1 INPUT OFFSET VOLTAGE OVER

TEMPERATURE

Table 1-1 gives both the linear and quadratic

temperature coefficients (TC1 and TC2) of input offset

voltage. The input offset voltage, at any temperature in

the specified range, can be calculated as follows:

EQUATION 4-1:

4.3.2 DC GAIN PLOTS

Figure 2-9, Figure 2-10 and Figure 2-11 are histograms

of the reciprocals (in units of µV/V) of CMRR, PSRR

and AOL, respectively. They represent the change in

input offset voltage (VOS) with a change in common

mode input voltage (VCM), power supply voltage (VDD)

and output voltage (VOUT).

The 1/AOL histogram is centered near 0 µV/V because

the measurements are dominated by the op amp’s

input noise. The negative values shown represent

noise, not unstable behavior. We validate the op amps’

stability by making multiple measurements of VOS; an

unstable part would fail, because it would show either

greater variability in VOS, or the output stuck at one of

the rails.

4.3.3 OFFSET AT POWER UP

When these parts power up, the input offset (VOS)

starts at its uncorrected value (usually less than

±5 mV). Circuits with high DC gain can cause the

output to reach one of the two rails. In this case, the

time to a valid output is delayed by an output overdrive

time (like tODR), in addition to the startup time (like

tSTR).

It can be simple to avoid this extra startup time.

Reducing the gain is one method. Adding a capacitor

across the feedback resistor (RF) is another method.

4.3.4 SOURCE RESISTANCES

The input bias currents have two significant

components; switching glitches that dominate at room

temperature and below, and input ESD diode leakage

currents that dominate at +85°C and above.

Make the resistances seen by the inputs small and

equal. This minimizes the output offset caused by the

input bias currents.

The inputs should see a resistance on the order of 10Ω

to 1 kΩ at high frequencies (i.e., above 1 MHz). This

helps minimize the impact of switching glitches, which

are very fast, on overall performance. In some cases, it

may be necessary to add resistors in series with the

inputs to achieve this improvement in performance.

Small input resistances are needed for high gains.

Without them, parasitic capacitances can cause

positive feedback and instability.

4.3.5 SOURCE CAPACITANCE

The capacitances seen by the two inputs should be

small and matched. The internal switches connected to

the inputs dump charges on these capacitors; an offset

can be created if the capacitances do not match. Large

input capacitances and source resistances, together

with high gain, can lead to positive feedback and

instability.

VOS TA

() VOS TC1ΔTTC

2ΔT2

++=

Where:

ΔT=T

A–25°C

VOS(TA) = input offset voltage at TA

VOS = input offset voltage at +25°C

TC1= linear temperature coefficient

TC2= quadratic temperature

coefficient

MCP6V26/7/8

4.3.6 CAPACITIVE LOADS

Driving large capacitive loads can cause stability

problems for voltage feedback op amps. As the load

capacitance increases, the feedback loop’s phase

margin decreases and the closed-loop bandwidth is

reduced. This produces gain peaking in the frequency

response, with overshoot and ringing in the step

response. These auto-zeroed op amps have a different

output impedance than most op amps, due to their

unique topology.

When driving a capacitive load with these op amps, a

series resistor at the output (RISO in Figure 4-7)

improves the feedback loop’s phase margin (stability)

by making the output load resistive at higher

frequencies. The bandwidth will be generally lower

than the bandwidth with no capacitive load.

FIGURE 4-7: Output Resistor, RISO,

Stabi li ze s Capaci tiv e Load s.

Figure 4-8 gives recommended RISO values for

different capacitive loads and gains. The x-axis is the

normalized load capacitance (CL/GN2). The y-axis is

the normalized resistance (GNRISO).

GN is the circuit’s noise gain. For non-inverting gains,

GN and the Signal Gain are equal. For inverting gains,

GN is 1+|Signal Gain| (e.g., -1 V/V gives GN = +2 V/V).

FIGURE 4-8: Recommended RISO values

for Capacitive Loads.

After selecting RISO for your circuit, double check the

resulting frequency response peaking and step

response overshoot. Modify RISO's value until the

response is reasonable. Bench evaluation and

simulations with the MCP6V26/7/8 SPICE macro

model are helpful.

4.3.7 STABILIZING OUTPUT LOADS

This family of auto-zeroed op amps has an output

impedance (Figure 2-31 and Figure 2-32) that has a

double zero when the gain is low. This can cause a

large phase shift in feedback networks that have low

resistance near the part’s bandwidth. This large phase

shift can cause stability problems.

Figure 4-9 shows that the load on the output is

(RL+R

ISO)||(RF+R

G), where RISO is before the load

(like Figure 4-7). This load needs to be large enough to

maintain stability; it should be at least (2 kΩ)/GN.

FIGURE 4-9: Output Load.

4.3.8 GAIN PEAKING

Figure 4-10 shows an op amp circuit that represents

non-inverting amplifiers (VM is a DC voltage and VP is

the input) or inverting amplifiers (VP is a DC voltage

and VM is the input). The capacitances CN and CG rep-

resent the total capacitance at the input pins; they

include the op amp’s common mode input capacitance

(CCM), board parasitic capacitance and any capacitor

placed in parallel. The capacitance CFP represents the

parasitic capacitance coupling the output and

non-inverting input pins.

FIGURE 4-10: Amplifier with Parasitic

Capacitance.

CG acts in parallel with RG (except for a gain of +1 V/V),

which causes an increase in gain at high frequencies.

CG also reduces the phase margin of the feedback

loop, which becomes less stable. This effect can be

reduced by either reducing CG or RF||RG

CN and RN form a low-pass filter that affects the signal

at VP

. This filter has a single real pole at 1/(2πRNCN).

RISO

VOUT

MCP6V2X

100

1000

1.E-10 1.E-09 1.E-08 1.E-07 1.E-06

CL/GN2 (F)

Recommended GNRISO ()

100p 1n 10n 100n 1μ

100

GN = 1

GN = 2

GN = 5

GN  10

RGRF

VOUT

RLCL

MCP6V2X

VOUT

MCP6V2X

CFP

MCP6V26/7/8

The largest value of RF that should be used depends

on noise gain (see GN in Section 4.3.6, Capacitive

Loads), CG and the open-loop gain’s phase shift. An

approximate limit for RF is:

EQUATION 4-2:

Some applications may modify these values to reduce

either output loading or gain peaking (step response

overshoot).

At high gains, RG and CG need to be small in order to

prevent positive feedback and oscillations.

4.3.9 REDUCING UNDESIRED NOISE

AND SIGNALS

Reduce undesired noise and signals with:

• Low bandwidth signal filters:

- Minimizes random analog noise

- Reduces interfering signals

• Good PCB layout techniques:

- Minimizes crosstalk

- Minimizes parasitic capacitances and

inductances that interact with fast switching

edges

• Good power supply design:

- Provides isolation from other parts

- Filters interference on supply line(s)

4.3.10 SUPPLY BYPASSING AND

FILTERING

With this family of op amps, the power supply pin (VDD

for single supply) should have a local bypass capacitor

(i.e., 0.01 µF to 0.1 µF) within 2 mm of the pin for good

high-frequency performance.

These parts also need a bulk capacitor (i.e., 1 µF or

larger) within 100 mm to provide large, slow currents.

This bulk capacitor can be shared with other low noise,

analog parts.

In some cases, high-frequency power supply noise

(e.g., switched mode power supplies) may cause

undue intermodulation distortion, with a DC offset shift;

this noise needs to be filtered. Adding a resistor into the

supply connection can be helpful. This resistor needs

to be small enough to prevent a large drop in VDD for

the op amp, which would cause a reduced output range

and possible load-induced power supply noise. It also

needs to be large enough to dissipate little power when

VDD is turned on and off quickly. Figure 4-11 shows a

circuit with resistors in the supply connections. It gives

good rejection out to 1 MHz for switched mode power

supplies. Smaller resistors and capacitors are a better

choice for designs where the power supply is not as

noisy.

FIGURE 4-11: Additional Supply Filtering.

4.3.11 PCB DESIGN FOR DC PRECISION

In order to achieve DC precision on the order of ±1 µV,

many physical errors need to be minimized. The design

of the Printed Circuit Board (PCB), the wiring and the

thermal environment has a strong impact on the

precision achieved. A poor PCB design can easily be

more than 100 times worse than the MCP6V26/7/8 op

amps minimum and maximum specifications.

4.3.11.1 PCB Layout

Any time two dissimilar metals are joined together, a

temperature dependent voltage appears across the

junction (the Seebeck or thermo-junction effect). This

effect is used in thermocouples to measure tempera-

ture. The following are examples of thermo-junctions

on a PCB:

• Components (resistors, op amps, …) soldered to

a copper pad

• Wires mechanically attached to the PCB

• Jumpers

• Solder joints

•PCB vias

Typical thermo-junctions have temperature to voltage

conversion coefficients of 10 to 100 µV/°C (sometimes

higher).

Microchip’s AN1258 (“Op Amp Precision Design: PCB

Layout Techniques”) contains in depth information on

PCB layout techniques that minimize thermo-junction

effects. It also discusses other effects, such as

crosstalk, impedances, mechanical stresses and

humidity.

RF2 k

12 p F

---------------

GN2

×≤

VS_ANA

50Ω50Ω

100 µF 100 µF

0.1 µF

1/4W 1/10W

to other analog parts

MCP6V2X

MCP6V26/7/8

4.3.11.2 Crosstalk

DC crosstalk causes offsets that appear as a larger

input offset voltage. Common causes include:

• Common mode noise (remote sensors)

• Ground loops (current return paths)

• Power supply coupling

Interference from the mains (usually 50 Hz or 60 Hz),

and other AC sources, can also affect the DC perfor-

mance. Non-linear distortion can convert these signals

to multiple tones, including a DC shift in voltage. When

the signal is sampled by an ADC, these AC signals can

also be aliased to DC, causing an apparent shift in

offset.

To reduce interference:

- Keep traces and wires as short as possible

- Use shielding (e.g., encapsulant)

- Use ground plane (at least a star ground)

- Place the input signal source near to the DUT

- Use good PCB layout techniques

- Use a separate power supply filter (bypass

capacitors) for these auto-zeroed op amps

4.3.11.3 Miscellaneous Effects

Keep the resistances seen by the input pins as small

and as near to equal as possible, to minimize bias

current-related offsets.

Make the (trace) capacitances seen by the input pins

small and equal. This is helpful in minimizing switching

glitch-induced offset voltages.

Bending a coax cable with a radius that is too small

causes a small voltage drop to appear on the center

conductor (the tribo-electric effect). Make sure the

bending radius is large enough to keep the conductors

and insulation in full contact.

Mechanical stresses can make some capacitor types

(such as ceramic) to output small voltages. Use more

appropriate capacitor types in the signal path and

minimize mechanical stresses and vibration.

Humidity can cause electro-chemical potential voltages

to appear in a circuit. Proper PCB cleaning helps, as

does the use of encapsulants.

4.4 Typical Applications

4.4.1 WHEATSTONE BRIDGE

Many sensors are configured as Wheatstone bridges.

Strain gauges and pressure sensors are two common

examples. These signals can be small and the

common mode noise large. Amplifier designs with high

differential gain are desirable.

Figure 4-12 shows how to interface to a Wheatstone

bridge with a minimum of components. Because the

circuit is not symmetric, the ADC input is single ended,

there is a minimum of filtering, and the CMRR is good

enough for moderate common mode noise.

FIGURE 4-12: Simp le Des ign .

Figure 4-13 shows a higher performance circuit for

Wheatstone bridges. This circuit is symmetric and has

high CMRR. Using a differential input to the ADC helps

with the CMRR.

FIGURE 4-13: High Performance Design.

VDD

100R

0.01C

ADC

VDD

0.2R

3kΩ

MCP6V26

20 kΩ

1µF

200Ω

20 kΩ

1µF

ADC

VDD

200 Ω

3kΩ

1µF

VDD

10 nF

200Ω

U1A

½ MCP6V27

U1B

½ MCP6V27

MCP6V26/7/8

4.4.2 RTD SENSOR

The ratiometric circuit in Figure 4-14 conditions a three

wire RTD. It corrects for the sensor’s wiring resistance

by subtracting the voltage across the middle RW. The

top R1 does not change the output voltage; it balances

the op amp inputs. Failure (open) of the RTD is

detected by an out-of-range voltage.

FIGURE 4-14: RTD Sensor.

The voltages at the input of the ADC can be calculated

with the following:

4.4.3 THERMOCOUPLE SENSOR

Figure 4-15 shows a simplified diagram of an amplifier

and temperature sensor used in a thermocouple

application. The type K thermocouple senses the

temperature at the hot junction (THJ), and produces a

voltage at V1 proportional to THJ (in °C). The amplifier’s

gain is set so that V4/THJ is 10 mV/°C. V3 represents

the output of a temperature sensor, which produces a

voltage proportional to the temperature (in °C) at the

cold junction (TCJ), and with a 0.50V offset. V2 is set so

that V4 is 0.50V when THJ –T

CJ is 0°C.

EQUATION 4-3:

FIGURE 4-15: Thermocouple Sensor;

Simplified Circuit.

Figure 4-16 shows a more complete implementation of

this circuit. The dashed red arrow indicates a thermally

conductive connection between the thermocouple and

the MCP9700A; it needs to be very short and have low

thermal resistance.

FIGURE 4-16: Thermocouple Sensor.

100 nF

10 nF

100 nF

ADC

VDD

2.49 kΩ

10 nF

VDD

RRTD

1µF

100Ω

3kΩ

20 kΩ

100 kΩ

2.49 kΩ

2.55 kΩ

U1A

½ MCP6V27

U1B

½ MCP6V27

VDM GRTD VTVB

–()GWVW

VCM VTVBGRTD 1G

–+()VW

------------------------------------------------------------------------------=

GRTD 12R

3R2

⁄⋅

GWGRTD R3R1

⁄

–=

Where:

VT= Voltage at the top of RRTD

VB= Voltage at the bottom of RRTD

VW= Voltage across top and middle RW’s

VCM = ADC’s common mode input

VDM = ADC’s differential mode input

V1≈THJ(40 µV/°C)

V2= (1.00V)

V3=T

CJ(10 mV/°C) + (0.50V)

V4=250V

1+(V

2–V

≈(10 mV/°C) (THJ –T

CJ)+(0.50V)

(RTH)/250

(RTH)

(RTH)/250

(RTH)

Type K

40 µV/°C

(RTH)

(hot junction

(cold junction

Thermocouple

at THJ)

at TCJ)

RTH = Thevenin Equivalent Resistance

MCP6V26

(RTH)/250

0.5696(RTH)

(RTH)/250

(RTH)

CV4

Type K

(RTH)

4.100(RTH)

Temp. Se nsor

VDD

VREF

VDD

3kΩ

RTH = Thevenin Equivalent Resistance (e.g., 10 kΩ)

MCP6V26

MCP1541

MCP9700A

MCP6V26/7/8

The MCP9700A senses the temperature at its physical

location. It needs to be at the same temperature as the

cold junction (TCJ), and produces V3 (Figure 4-15).

The MCP1541 produces a 4.10V output, assuming

VDD is at 5.0V. This voltage, tied to a resistor ladder of

4.100(RTH) and 1.3224(RTH), would produce a

Thevenin equivalent of 1.00V and 250(RTH). The

1.3224(RTH) resistor is combined in parallel with the

top right RTH resistor (in Figure 4-15), producing the

0.5696(RTH) resistor.

V4 should be converted to digital, then corrected for the

thermocouple’s non-linearity. The ADC can use the

MCP1541 as its voltage reference. Alternately, an

absolute reference inside a PICmicro® device can be

used instead of the MCP1541.

4.4.4 OFFSET VOLTAGE CORRECTION

Figure 4-17 shows an MCP6V27 correcting the input

offset voltage of another op amp. R2 and C2 integrate

the offset error seen at the other op amp’s input; the

integration needs to be slow enough to be stable (with

the feedback provided by R1 and R3).

FIGURE 4-17: Offset Correction.

4.4.5 PRECISION COMPARATOR

Use high gain before a comparator to improve the

latter’s performance. Do not use MCP6V26/7/8 as a

comparator by itself; the VOS correction circuitry does

not operate properly without a feedback loop.

FIGURE 4-18: Precision Comparator.

R1R3

VDD/2

VIN VOUT

MCP6V26

VDD/2

MCP661

VIN

VDD/2

VOUT

1kΩ

MCP6V26

MCP6541

MCP6V26/7/8

NOTES:

MCP6V26/7/8

5.0 DESIGN AIDS

Microchip provides the basic design aids needed for

the MCP6V26/7/8 family of op amps.

5.1 SPICE Macro Model

The latest SPICE macro model for the MCP6V26/7/8

family of op amps is available on the Microchip web site

at www.microchip.com. This model is intended to be an

initial design tool that works well in the op amp’s linear

region of operation over the temperature range. See

the model file for information on its capabilities.

Bench testing is a very important part of any design and

cannot be replaced with simulations. Also, simulation

results using this macro model need to be validated by

comparing them to the data sheet specifications and

characteristic curves.

5.2 FilterLab® Software

Microchip’s FilterLab® software is an innovative

software tool that simplifies analog active filter (using

op amps) design. Available at no cost from the

Microchip web site at www.microchip.com/filterlab, the

Filter-Lab design tool provides full schematic diagrams

of the filter circuit with component values. It also

outputs the filter circuit in SPICE format, which can be

used with the macro model to simulate actual filter

performance.

5.3 Microchip Advanced Part Selector

(MAPS)

MAPS is a software tool that helps efficiently identify

Microchip devices that fit a particular design require-

ment. Available at no cost from the Microchip website

at www.microchip.com/maps, the MAPS is an overall

selection tool for Microchip’s product portfolio that

includes Analog, Memory, MCUs and DSCs. Using this

tool, a customer can define a filter to sort features for a

parametric search of devices and export side-by-side

technical comparison reports. Helpful links are also

provided for Data sheets, Purchase and Sampling of

Microchip parts.

5.4 Analog Demonstration and

Evaluation Boards

Microchip offers a broad spectrum of Analog Demon-

stration and Evaluation Boards that are designed to

help customers achieve faster time to market. For a

complete listing of these boards and their correspond-

ing user’s guides and technical information, visit the

Microchip web site at www.microchip.com/analogtools.

Some boards that are especially useful are:

• MCP6V01 Thermocouple Auto-Zeroed Reference

Design

• MCP6XXX Amplifier Evaluation Board 1

• MCP6XXX Amplifier Evaluation Board 2

• MCP6XXX Amplifier Evaluation Board 3

• MCP6XXX Amplifier Evaluation Board 4

• Active Filter Demo Board Kit

• P/N SOIC8EV: 8-Pin SOIC/MSOP/TSSOP/DIP

Evaluation Board

• P/N SOIC14EV: 14-Pin SOIC/TSSOP/DIP

Evaluation Board

5.5 Application Notes

The following Microchip Application Notes are

available on the Microchip web site at www.microchip.

com/appnotes and are recommended as supplemental

reference resources.

ADN003: “Select the Right Operational Amplifier for

your Filteri ng Circuits”, DS21821

AN722: “Operational Amplifier Topologies and DC

Specifications”, DS00722

AN723: “Operational Amplifier AC Specifications and

Applications”, DS00723

AN884: “Driving Capacitive Loads With Op Amps”,

DS00884

AN990: “Analog Sensor Conditioning Circuits – An

Overview”, DS00990

AN1177: “Op Amp Precision Design: DC Errors”,

DS01177

AN1228: “Op Amp Precision Design: Random Noise”,

DS01228

AN1258: “Op Amp Precision Design: PCB Layout

Techniques”, DS01258

These application notes and others are listed in the

design guide:

“Signal Chain Design Guide”, DS21825

MCP6V26/7/8

NOTES:

MCP6V26/7/8

6.0 PACKAGING INFORMATION

6.1 Package Marking Information

8-Lead TDFN (2x3x0.75 mm)

(MCP6V26, MCP6V28)

Example

Legend: XX...X Customer-specific information

Y Year code (last digit of calendar year)

YY Year code (last 2 digits of calendar year)

WW Week code (week of January 1 is week ‘01’)

NNN Alphanumeric traceability code

Pb-free JEDEC designator for Matte Tin (Sn)

*This package is Pb-free. The Pb-free JEDEC designator ( )

can be found on the outer packaging for this package.

Note: In the event the full Microchip part number cannot be marked on one line, it will

be carried over to the next line, thus limiting the number of available

characters for customer-specific information.

8-Lead DFN (4x4x0.9 mm) (MCP6V27)Example

YYWW

NNN

XXXXXX

PIN 1 PIN 1

6V27

E/MD

1129

256

8-Lead MSOP (3x3 mm) Example

6V27E

129256

8-Lead SOIC (3.90 mm) Example

NNN

MCP6V27E

SN^^ 1129

256

Device Code

MCP6V26T-E/MNY ABA

MCP6V28T-E/MNY ABB

ABA

129

MCP6V26/7/8

8-Lead Plastic Dual Flat, No Lead Package (MD) – 4x4x0.9 mm Body [DFN]

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

Microchip Technology Drawing C04-131E Sheet 1 of 2

MCP6V26/7/8

8-Lead Plastic Dual Flat, No Lead Package (MD) – 4x4x0.9 mm Body [DFN]

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

Microchip Technology Drawing C04-131E Sheet 2 of 2

MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

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  

  

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  

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NOTE 1

A2 c

L1 L

   

MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

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MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

Note: For the most current package drawings, please see the Microchip Packaging Specification located at

http://www.microchip.com/packaging

MCP6V26/7/8

+," -.#.&$/0'(1+,

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MCP6V26/7/8

APPENDIX A: REVISION HISTORY

Revision B (August 2011)

The following is the list of modifications:

1. Added the MCP6V26 and MCP6V28 single op

amps.

a) Updated package drawings on page 1.

b) Updated the pinout table (Ta b l e 3 - 1 ).

c) Added 8-lead, 2×3 TDFN package to the

Thermal Characteristics Table (Tab le 1 -4 ).

d) Added 8-lead, 2×3 TDFN package to

Section 6.0 “Packaging Information”.

e) Added parts numbers to Product Identifica-

tion System.

2. Added Chip Select (CS) information.

a) Added Digital Electrical Specifications table

(Ta b l e 1 - 3 ).

b) Added Timing Diagram (Figure 1-4).

c) Added Section 2.6 “Chip Select

Response (MCP6V28 only)” to the Typical

Performance Curves.

d) Added Section 4.2.3 “Chip Select (CS)”

to the applications write up.

3. Added information on positive feedback and

parasitic feedback capacitance.

a) Added to Section 4.3.4 “Source

Resistances”.

b) Added to Section 4.3.5 “Source

Capacitance”.

c) Modified Figure 4-10.

d) Added to Section 4.3.8 “Gain Peaking”.

4. Other minor typographical corrections.

Revision A (March 2011)

• Original data sheet for the MCP6V27 dual op

amps.

MCP6V26/7/8

APPENDIX B: OFFSET RELATED

TEST SCREENS

Input offset voltage-related specifications in the DC

spec table (Ta b l e 1 - 1 ) are based on bench

measurements (see Section 2.1 “DC Input

Precision”). These measurements are much more

accurate because:

• More compact circuit

• Soldered parts on the PCB (to validate other

measurements)

• More time spent averaging (reduces noise)

• Better temperature control

- Reduced temperature gradients

- Greater accuracy

We use production screens to ensure the quality of our

outgoing products. These screens are set at wider

limits to eliminate any fliers; see Table B-1.

TABLE B-1: OFFSET RELATED TEST SCREENS

Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.3V to +5.5V, VSS = GND,

VCM = VDD/3, VOUT =V

DD/2, VL=V

DD/2, RL = 10 kΩ to VL and CS = GND (refer to Figure 1-5 and Figure 1-6).

Parameters Sym Min Max Units Conditions

Input Offset

Input Offset Voltage VOS -10 +10 µV TA = +25°C (Note 1, Note 2)

Input Offset Voltage Drift with Temperature

(linear Temp. Co.)

TC1——nV/°CT

A = -40 to +125°C (Note 3)

Power Supply Rejection PSRR 115 — dB (Note 1)

Common Mode

Common Mode Rejection CMRR 106 — dB VDD = 2.3V, VCM = -0.15V to 2.5V (Note 1)

CMRR 116 — dB VDD = 5.5V, VCM = -0.15V to 5.7V (Note 1)

Open-Loop Gain

DC Open-Loop Gain (large signal) AOL 114 — dB VDD = 2.3V, VOUT = 0.2V to 2.1V (Note 1)

AOL 122 — dB VDD = 5.5V, VOUT = 0.2V to 5.3V (Note 1)

Note 1: Due to thermal junctions and other errors in the production environment, these specifications are only

screened in production.

2: VOS is also sample screened at +125°C.

3: TC1 is not measured in production.

MCP6V26/7/8

PRODUCT IDENTIFICATION SYSTEM

To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office.

Device: MCP6V26 Single Op Amp

MCP6V26T Single Op Amp (Tape and Reel)

MCP6V27 Dual Op Amp

MCP6V27T Dual Op Amp (Tape and Reel)

MCP6V28 Single Op Amp with Chip Select

MCP6V28T Single Op Amp with Chip Select

(Tape and Reel)

Temperature Range: E = -40°C to +125°C

Package: MD = Plastic Dual Flat, No-Lead (4×4x0.9), 8-lead

MNY * = Plastic Dual Flat, No-Lead (2×3x0.75), 8-lead

MS = Plastic Micro Small Outline Package, 8-lead

SN = Plastic SOIC (150mil Body), 8-lead

* Y = Nickel Palladium gold manufacturing designator. Only

available on the TDFN package.

PART NO. –X /XX

PackageTemperature

Range

Device

Examples:

a) MCP6V26T-E/MNY: Extended temperature,

8LD 2×3 TDFN

package

b) MCP6V26-E/MS: Extended temperature,

8LD MSOP package

a) MCP6V26T-E/SN: Tape and Reel,

Extended temperature,

8LD SOIC package

a) MCP6V27-E/MD: Extended temperature,

8LD 4x4 DFN package

b) MCP6V27-E/MS: Extended temperature,

8LD MSOP package

c) MCP6V27-E/SN: Extended temperature,

8LD SOIC package

a) MCP6V28T-E/MNY: Extended temperature,

8LD 2×3 TDFN

package

b) MCP6V28-E/MS: Extended temperature,

8LD MSOP package

c) MCP6V28T-E/SN: Tape and Reel,

Extended temperature,

8LD SOIC package

MCP6V26/7/8

NOTES:

Information contained in this publication regarding device

applications and the like is provided only for your convenience

and may be superseded by updates. It is your responsibility to

ensure that your application meets with your specifications.

MICROCHIP MAKES NO REPRESENTATIONS OR

WARRANTIES OF ANY KIND WHETHER EXPRESS OR

IMPLIED, WRITTEN OR ORAL, STATUTORY OR

OTHERWISE, RELATED TO THE INFORMATION,

INCLUDING BUT NOT LIMITED TO ITS CONDITION,

QUALITY, PERFORMANCE, MERCHANTABILITY OR

FITNESS FOR PURPOSE. Microchip disclaims all liability

arising from this information and its use. Use of Microchip

devices in life support and/or safety applications is entirely at

the buyer’s risk, and the buyer agrees to defend, indemnify and

hold harmless Microchip from any and all damages, claims,

suits, or expenses resulting from such use. No licenses are

conveyed, implicitly or otherwise, under any Microchip

intellectual property rights.

Trademarks

The Microchip name and logo, the Microchip logo, dsPIC,

KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART,

PIC32 logo, rfPIC and UNI/O are registered trademarks of

Microchip Technology Incorporated in the U.S.A. and other

countries.

FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor,

MXDEV, MXLAB, SEEVAL and The Embedded Control

Solutions Company are registered trademarks of Microchip

Technology Incorporated in the U.S.A.

Analog-for-the-Digital Age, Application Maestro, chipKIT,

chipKIT logo, CodeGuard, dsPICDEM, dsPICDEM.net,

dsPICworks, dsSPEAK, ECAN, ECONOMONITOR,

FanSense, HI-TIDE, In-Circuit Serial Programming, ICSP,

Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB,

MPLINK, mTouch, Omniscient Code Generation, PICC,

PICC-18, PICDEM, PICDEM.net, PICkit, PICtail, REAL ICE,

rfLAB, Select Mode, Total Endurance, TSHARC,

UniWinDriver, WiperLock and ZENA are trademarks of

Microchip Technology Incorporated in the U.S.A. and other

countries.

SQTP is a service mark of Microchip Technology Incorporated

in the U.S.A.

All other trademarks mentioned herein are property of their

respective companies.

Printed on recycled paper.

ISBN: 978-1-61341-503-0

Note the following details of the code protection feature on Microchip devices:

• Microchip products meet the specification contained in their particular Microchip Data Sheet.

• Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the

intended manner and under normal conditions.

• There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our

knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data

Sheets. Most likely, the person doing so is engaged in theft of intellectual property.

• Microchip is willing to work with the customer who is concerned about the integrity of their code.

• Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not

mean that we are guaranteeing the product as “unbreakable.”

Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our

products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts

allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.

Microchip received ISO/TS-16949:2009 certification for its worldwide

headquarters, design and wafer fabrication facilities in Chandler and

Tempe, Arizona; Gresham, Oregon and design centers in California

and India. The Company’s quality system processes and procedures

are for its PIC® MCUs and d sPIC® DSCs, KEELOQ® code hopping

devices, Serial EEPROMs, microperiph erals, nonvolatile memory and

analog products. In addition, Microchip’s quality system for the design

and manufacture of development systems is ISO 9001:2000 certified.

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