REV. B
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Tel: 781/329-4700 www.analog.com
Fax: 781/326-8703 © 2003 Analog Devices, Inc. All rights reserved.
AD8041
160 MHz Rail-to-Rail
Amplifier with Disable
FEATURES
Fully Specified for +3 V, +5 V, and ⴞ5 V Supplies
Output Swings Rail to Rail
Input Voltage Range Extends 200 mV Below Ground
No Phase Reversal with Inputs 1 V Beyond Supplies
Disable/Power-Down Capability
Low Power of 5.2 mA (26 mW on 5 V)
High Speed and Fast Settling on 5 V:
160 MHz –3 dB Bandwidth (G = +1)
160 V/␮s Slew Rate
30 ns Settling Time to 0.1%
Good Video Specifications (RL = 150 ⍀, G = +2)
Gain Flatness of 0.1 dB to 30 MHz
0.03% Differential Gain Error
0.03ⴗ Differential Phase Error
Low Distortion
–69 dBc Worst Harmonic @ 10 MHz
Outstanding Load Drive Capability
Drives 50 mA 0.5 V from Supply Rails
Cap Load Drive of 45 pF
APPLICATIONS
Power Sensitive High Speed Systems
Video Switchers
Distribution Amplifiers
A/D Drivers
Professional Cameras
CCD Imaging Systems
Ultrasound Equipment (Multichannel)
Single-Supply Multiplexer
CONNECTION DIAGRAM
8-Lead PDIP, CERDIP and SOIC
ⴙVS
DISABLE
1
2
3
4
8
7
6
5
NC = NO CONNECT
NC
NC
OUTPUT
–INPUT
ⴙINPUT
–VSAD8041
(Top View)
PRODUCT DESCRIPTION
The AD8041 is a low power voltage feedback, high speed ampli-
fier designed to operate on +3 V, +5 V, or ±5 V supplies. It has
true single-supply capability with an input voltage range extending
200 mV below the negative rail and within 1 V of the positive rail.
5V
2.5V
0V 200ns
1V
Figure 1. Output Swing: G = –1, V
S
= 5 V
The output voltage swing extends to within 50 mV of each rail,
providing the maximum output dynamic range. Additionally, it
features gain flatness of 0.1 dB to 30 MHz while offering differ-
ential gain and phase error of 0.03% and 0.03° on a single 5 V
supply. This makes the AD8041 ideal for professional video
electronics such as cameras, video switchers, or any high speed
portable equipment. The AD8041’s low distortion and fast settling
make it ideal for buffering high speed A-to-D converters.
The AD8041 has a high speed disable feature useful for mul-
tiplexing or for reducing power consumption (1.5 mA). The
disable logic interface is compatible with CMOS or open-
collector logic. The AD8041 offers a low power supply current
of 5.8 mA maximum and can run on a single 3 V power supply.
These features are ideally suited for portable and battery-
powered applications where size and power are critical.
The wide bandwidth of 160 MHz along with 160 V/µs of slew
rate on a single 5 V supply make the AD8041 useful in many
general-purpose high speed applications where dual power
supplies of up to ±6 V and single supplies from 3 V to 12 V are
needed. The AD8041 is available in 8-lead PDIP and SOIC
over the industrial temperature range of –40°C to +85°C.
FREQUENCY (MHz)
V
S
= 5V
G = +2
R
F
= 400⍀
0
100
NORMALIZED GAIN (dB)
80
60
40
20
0
2
1
–2
–1
–8
–7
–6
–5
–4
–3
Figure 2. Frequency Response: G = +2, V
S
= 5 V
REV. B–2–
AD8041–SPECIFICATIONS
(@ T
A
= 25ⴗC, V
S
= 5 V, R
L
= 2 k⍀ to 2.5 V, unless otherwise noted.)
AD8041A
Parameter Conditions Min Typ Max Unit
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, V
O
< 0.5 V p-p G = +1 130 160 MHz
Bandwidth for 0.1 dB Flatness G = +2, R
L
= 150 Ω30 MHz
Slew Rate G = –1, V
O
= 2 V Step 130 160 V/µs
Full Power Response V
O
= 2 V p-p 24 MHz
Settling Time to 0.1% G = –1, V
O
= 2 V Step 35 ns
Settling Time to 0.01% 55 ns
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion f
C
= 5 MHz, V
O
= 2 V p-p, G = +2, R
L
= 1 kΩ–72 dB
Input Voltage Noise f = 10 kHz 16 nV/√Hz
Input Current Noise f = 10 kHz 600 fA/√Hz
Differential Gain Error (NTSC) G = +2, R
L
= 150 Ω to 2.5 V 0.03 %
G = +2, R
L
= 75 Ω to 2.5 V 0.01 %
Differential Phase Error (NTSC) G = +2, R
L
= 150 Ω to 2.5 V 0.03 Degrees
G = +2, R
L
= 75 Ω to 2.5 V 0.19 Degrees
DC PERFORMANCE
Input Offset Voltage 27mV
T
MIN
to T
MAX
8mV
Offset Drift 10 µV/°C
Input Bias Current 1.2 3.2 µA
T
MIN
to T
MAX
3.5 µA
Input Offset Current 0.2 0.5 µA
Open-Loop Gain R
L
= 1 kΩ86 95 dB
T
MIN
to T
MAX
90 dB
INPUT CHARACTERISTICS
Input Resistance 160 kΩ
Input Capacitance 1.8 pF
Input Common-Mode Voltage Range –0.2 to +4 V
Common-Mode Rejection Ratio V
CM
= 0 V to 3.5 V 74 80 dB
OUTPUT CHARACTERISTICS
Output Voltage Swing: R
L
= 10 kΩ0.05 to 4.95 V
Output Voltage Swing: R
L
= 1 kΩ0.35 to 4.75 0.1 to 4.9 V
Output Voltage Swing: R
L
= 50 Ω0.4 to 4.4 0.3 to 4.5 V
Output Current V
OUT
= 0.5 V to 4.5 V 50 mA
Short-Circuit Current Sourcing 90 mA
Sinking 150 mA
Capacitive Load Drive G = +1 45 pF
POWER SUPPLY
Operating Range 312V
Quiescent Current 5.2 5.8 mA
Quiescent Current (Disabled) 1.4 1.7 mA
Power Supply Rejection Ratio V
S
= 0, +5 V, ±1 V 72 80 dB
DISABLE CHARACTERISTICS V
O
= 2 V p-p @ 10 MHz, G = +2
Turn-Off Time R
F
= R
L
= 2 kΩ120 ns
Turn-On Time R
F
= R
L
= 2 kΩ230 ns
Off Isolation (Pin 8 Tied to –V
S
)R
L
= 100 Ω, f = 5 MHz, G = +2, R
F
= 1 kΩ70 dB
Off Voltage (Device Disabled) <V
S
– 2.5 V
On Voltage (Device Enabled) Open or +V
S
V
Specifications subject to change without notice.
REV. B
AD8041
–3–
SPECIFICATIONS
(@ T
A
= 25ⴗC, V
S
= 3 V, R
L
= 2 k⍀ to 1.5 V, unless otherwise noted.)
AD8041A
Parameter Conditions Min Typ Max Unit
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, V
O
< 0.5 V p-p G = +1 120 150 MHz
Bandwidth for 0.1 dB Flatness G = +2, R
L
= 150 Ω25 MHz
Slew Rate G = –1, V
O
= 2 V Step 120 150 V/µs
Full Power Response V
O
= 2 V p-p 20 MHz
Settling Time to 0.1% G = –1, V
O
= 2 V Step 40 ns
Settling Time to 0.01% 55 ns
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion f
C
= 5 MHz, V
O
= 2 V p-p, G = –1, R
L
= 100 Ω–55 dB
Input Voltage Noise f = 10 kHz 16 nV/√Hz
Input Current Noise f = 10 kHz 600 fA/√Hz
Differential Gain Error (NTSC) G = +2, R
L
= 150 Ω to 1.5 V, Input V
CM
= 1 V 0.07 %
Differential Phase Error (NTSC) G = +2, R
L
= 150 Ω to 1.5 V, Input V
CM
= 1 V 0.05 Degrees
DC PERFORMANCE
Input Offset Voltage 27mV
T
MIN
to T
MAX
8mV
Offset Drift 10 µV/°C
Input Bias Current 1.2 3.2 µA
T
MIN
to T
MAX
3.5 µA
Input Offset Current 0.2 0.6 µA
Open-Loop Gain R
L
= 1 kΩ85 94 dB
T
MIN
to T
MAX
89 dB
INPUT CHARACTERISTICS
Input Resistance 160 kΩ
Input Capacitance 1.8 pF
Input Common-Mode Voltage Range –0.2 to +2 V
Common-Mode Rejection Ratio V
CM
= 0 V to 1.5 V 72 80 dB
OUTPUT CHARACTERISTICS
Output Voltage Swing: R
L
= 10 kΩ0.05 to 2.95 V
Output Voltage Swing: R
L
= 1 kΩ0.45 to 2.7 0.1 to 2.9 V
Output Voltage Swing: R
L
= 50 Ω0.5 to 2.6 0.25 to 2.75 V
Output Current V
OUT
= 0.5 V to 2.5 V 50 mA
Short-Circuit Current Sourcing 70 mA
Sinking 120 mA
Capacitive Load Drive G = +1 40 pF
POWER SUPPLY
Operating Range 312V
Quiescent Current 5.0 5.6 mA
Quiescent Current (Disabled) 1.3 1.5 mA
Power Supply Rejection Ratio V
S
= 0, +3 V, ±0.5 V 68 80 dB
DISABLE CHARACTERISTICS V
O
= 2 V p-p @ 10 MHz, G = +2
Turn-Off Time R
F
= R
L
= 2 kΩ90 ns
Turn-On Time R
F
= R
L
= 2 kΩ170 ns
Off Isolation (Pin 8 Tied to –V
S
)R
L
= 100 Ω, f = 5 MHz, G = +2, R
F
= 1 kΩ70 dB
Off Voltage (Device Disabled) <V
S
– 2.5 V
On Voltage (Device Enabled) Open or +V
S
V
Specifications subject to change without notice.
REV. B–4–
AD8041
SPECIFICATIONS
(@ T
A
= 25ⴗC, V
S
= ⴞ5 V, R
L
= 2 k⍀ to 0 V, unless otherwise noted.)
AD8041A
Parameter Conditions Min Typ Max Unit
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, V
O
< 0.5 V p-p G = +1 140 170 MHz
Bandwidth for 0.1 dB Flatness G = +2, R
L
= 150 Ω32 MHz
Slew Rate G = –1, V
O
= 2 V Step 140 170 V/µs
Full Power Response V
O
= 2 V p-p 26 MHz
Settling Time to 0.1% G = –1, V
O
= 2 V Step 30 ns
Settling Time to 0.01% 50 ns
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion f
C
= 5 MHz, V
O
= 2 V p-p, G = +2, R
L
= 1 kΩ–77 dB
Input Voltage Noise f = 10 kHz 16 nV/√Hz
Input Current Noise f = 10 kHz 600 fA/√Hz
Differential Gain Error (NTSC) G = +2, R
L
= 150 Ω0.02 %
G = +2, R
L
= 75 Ω0.02 %
Differential Phase Error (NTSC) G = +2, R
L
= 150 Ω0.03 Degrees
G = +2, R
L
= 75 Ω0.10 Degrees
DC PERFORMANCE
Input Offset Voltage 27mV
T
MIN
to T
MAX
8mV
Offset Drift 10 µV/°C
Input Bias Current 1.2 3.2 µA
T
MIN
to T
MAX
3.5 µA
Input Offset Current 0.2 0.6 µA
Open-Loop Gain R
L
= 1 kΩ90 99 dB
T
MIN
to T
MAX
95 dB
INPUT CHARACTERISTICS
Input Resistance 160 kΩ
Input Capacitance 1.8 pF
Input Common-Mode Voltage Range –5.2 to +4 V
Common-Mode Rejection Ratio V
CM
= –5 V to +3.5 V 72 80 dB
OUTPUT CHARACTERISTICS
Output Voltage Swing: R
L
= 10 kΩ–4.95 to +4.95 V
Output Voltage Swing: R
L
= 1 kΩ–4.45 to +4.6 –4.8 to +4.8 V
Output Voltage Swing: R
L
= 50 Ω–4.3 to +3.2 –4.5 to +3.8 V
Output Current V
OUT
= –4.5 V to +4.5 V 50 mA
Short-Circuit Current Sourcing 100 mA
Sinking 160 mA
Capacitive Load Drive G = +1 50 pF
POWER SUPPLY
Operating Range 312V
Quiescent Current 5.8 6.5 mA
Quiescent Current (Disabled) 1.6 2.2 mA
Power Supply Rejection Ratio V
S
= –5 V, +5 V, ±1 V 68 80 dB
DISABLE CHARACTERISTICS V
O
= 2 V p-p @ 10 MHz, G = +2
Turn-Off Time R
F
= 2 kΩ120 ns
Turn-On Time R
F
= 2 kΩ320 ns
Off Isolation (Pin 8 Tied to –V
S
)R
L
= 100 Ω, f = 5 MHz, G = +2, R
F
= 1 kΩ70 dB
Off Voltage (Device Disabled) <V
S
– 2.5 V
On Voltage (Device Enabled) Open or +V
S
V
Specifications subject to change without notice.
REV. B
AD8041
–5–
ABSOLUTE MAXIMUM RATINGS
1
Supply Voltage ............................................................ 12.6 V
Internal Power Dissipation
2
PDIP Package (N) .................................................... 1.3 W
SOIC Package (R) .................................................... 0.9 W
Input Voltage (Common Mode) ...................................... ±V
S
Differential Input Voltage ........................................... ±3.4 V
Output Short-Circuit Duration
.......................................... Observe Power Derating Curves
Storage Temperature Range N, R .............. –65°C to +125°C
Operating Temperature Range (A Grade) ... –40°C to +85°C
Lead Temperature Range (Soldering 10 sec) ............... 300°C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for the device in free air:
8-Lead PDIP Package: θ
JA
= 90°C/W.
8-Lead SOIC Package: θ
JA
= 155°C/W.
MAXIMUM POWER DISSIPATION
The maximum power that can be safely dissipated by the
AD8041 is limited by the associated rise in junction temperature.
The maximum safe junction temperature for plastic encapsulated
devices is determined by the glass transition temperature of the
plastic, approximately 150°C. Exceeding this limit temporarily
may cause a shift in parametric performance due to a change in
the stresses exerted on the die by the package. Exceeding a
junction temperature of 175°C for an extended period can result
in device failure.
While the AD8041 is internally short-circuit protected, this may
not be sufficient to guarantee that the maximum junction tem-
perature (150°C) is not exceeded under all conditions. To
ensure proper operation, it is necessary to observe the maximum
power derating curves.
MAXIMUM POWER DISSIPATION (W)
AMBIENT TEMPERATURE (ⴗC)
2.0
1.5
0
–50 90–40 –30 –20 –10 0 10 20 30 50 60 70 8040
1.0
0.5
8-LEAD PDIP PACKAGE
8-LEAD SOIC PACKAGE
TJ = 150ⴗC
Figure 3. Maximum Power Dissipation vs. Temperature
ORDERING GUIDE
Temperature Package Package
Model Range Description Options
AD8041AN –40°C to +85°C8-Lead PDIP N-8
AD8041AR –40°C to +85°C8-Lead Plastic SOIC R-8
AD8041AR-REEL –40°C to +85°C13" Tape and Reel R-8
AD8041AR-REEL7 –40°C to +85°C7" Tape and Reel R-8
AD8041ARZ-REEL
1
–40°C to +85°C13" Tape and Reel R-8
5962-9683901MPA
2
–55°C to +125°C8-Lead CERDIP Q-8
NOTES
1
The Z indicates a lead-free product.
2
Refer to official DSCC drawing for tested specifications.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD8041 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
REV. B–6–
AD8041–Typical Performance Characteristics
VOS (mV)
30
15
0
25
20
10
5
–6 6–5 –4 –3 –2 –1 0 1 2 3 4 5
NUMBER OF PARTS IN BIN
VS = ⴞ2.5V
TA = 25ⴗC
91 PARTS
MEAN = +0.21
STD DEVIATION = 1.47
TPC 1. Typical Distribution of V
OS
VOS DRIFT (␮V/ⴗC)
0.20
0.15
0
–10 10–7.5
PROBABILITY DENSITY
–5 –2.5 0 2.5 5 7.5
0.10
0.05
MEAN = 0.02␮V/ⴗC
STD DEV = 2.87␮V/ⴗC
SAMPLE SIZE = 45
TPC 2. V
OS
Drift Over –40
°
C to +85
°
C
TEMPERATURE (ⴗC)
2
1.5
0
–45 85–35 –25 –15 –5 5 15 25 35 45 55 65 75
1
0.5
INPUT BIAS CURRENT (␮A)
VS = 5V
VCM = 0V
TPC 3. I
B
vs. Temperature
LOAD RESISTANCE (⍀)
100
70
95
90
85
80
75
0 2000250 500 750 1000 1250 1500 1750
VS = 5V
TA = 25ⴗC
OPEN-LOOP GAIN (dB)
TPC 4. Open-Loop Gain vs. R
L
to 25
°
C
TEMPERATURE (ⴗC)
100
97
85
–60 –40 –20 0 20 40 60 80 100 120
94
91
88
OPEN-LOOP GAIN (dB)
VS = 5V
RL = 1k⍀ TO 2.5V
TPC 5. Open-Loop Gain vs. Temperature
OUTPUT VOLTAGE (V)
100
70
40
90
80
60
50
050.5 1 1.5 2 2.5 3 3.5 4 4.5
RL = 500⍀ TO 2.5V VS = 5V
RL = 50⍀ TO 2.5V
OPEN-LOOP GAIN (dB)
TPC 6. Open-Loop Gain vs. Output Voltage
REV. B
AD8041
–7–
FREQUENCY (Hz)
200
150
0
100
50
10 100k100
INPUT VOLTAAGE NOISE (nV/
Hz)
1k 10k
TPC 7. Input Voltage Noise vs. Frequency
FUNDAMENTAL FREQUENCY (MHz)
–30
–40
–1001102
TOTAL HARMONIC DISTORTION (dBc)
–60
–70
–80
–90
–50
3456789
VS = 3V, AV = –1,
RL = 100⍀ TO 1.5V
VS = 5V, AV = 1,
RL = 1k⍀ TO 2.5V
VS = 5V, AV = 2,
RL = 100⍀ TO 2.5V
VS = 5V, AV = 2,
RL = 1k⍀ TO 2.5V
VS = 5V, AV = 1,
RL = 100⍀ TO 2.5V
TPC 8. Total Harmonic Distortion
OUTPUT VOLTAGE (V
P-P
)
01.50.5 1 2 2.5 33.5 4 4.5 5
WORST HARMONIC (dBc)
–140
–30
–40
–50
–70
–100
–80
–90
–60
–110
–120
–130
10MHz
5MHz
1MHz
V
S
= 5V
R
L
= 2k⍀ TO 2.5V
G = +2
TPC 9. Worst Harmonic vs. Output Voltage
DIFF PHASE (Degrees) DIFF GAIN (%)
11th1st 6th2nd 3rd 4th 5th 7th 8th 9th 10th
–0.005
0.015
0.005
0.025
0.020
0.010
0.000
0.030
–0.010
0.035 VS = 5V
G = +2
RL = 150⍀ TO 2.5V
11th1st 6th2nd 3rd 4th 5th 7th 8th 9th 10th
VS = 5V
G = +2
RL = 150⍀
–0.005
0.015
0.005
0.025
0.020
0.010
0.000
0.030
–0.010
0.035 VS = 5V
G = +2
RL = 150⍀ TO 2.5V
VS = 5V
G = +2
RL = 150⍀
DC OUTPUT LEVEL (100 IRE MAX)
TPC 10. Differential Gain and Phase Errors
FREQUENCY (MHz)
6.5
6.4
5.5
6.3
6.2
6.1
6.0
5.9
5.8
5.7
5.6
1 50010 100
32.4MHz
VS = 5V
G = +2
RL = 150⍀ TO 2.5V
RF = 402⍀
CLOSED-LOOP GAIN (dB)
TPC 11. 0.1 dB Gain Flatness
–10
40
30
60
0
50
10
20
80
90
70
OPEN-LOOP GAIN (dB)
–180
0
–90
90
180
270
PHASE (ⴗC)
360
450
–270
–360
–450
5000.1
0.01 10010
FREQUENCY (MHz)
VS = 5V
RL = 2k⍀ TO 2.5V
CL = 5pF TO 2.5V
PHASE
GAIN
TPC 12. Open-Loop Gain and Phase vs. Frequency
REV. B–8–
AD8041
FREQUENCY (MHz)
5
4
–5
3
2
1
0
–1
–2
–3
–4
1 50010 100
VS = 5V
RL = 2k⍀ TO 2.5V
CL = 5pF
G = +1
T = +125ⴗC
T = +25ⴗC
T = –55ⴗC
CLOSED-LOOP GAIN (dB)
TPC 13. Closed-Loop Frequency Response
vs. Temperature
FREQUENCY
(
MHz
)
5
4
–5
3
2
1
0
–1
–2
–3
–4
1 50010 100
G = +1
R
L
= 2k⍀
C
L
= 5pF
V
S
= 3V
R
L
AND C
L
TO 1.5V
V
S
= ⴞ5V
CLOSED-LOOP GAIN (dB)
V
S
= 5V
R
L
AND C
L
TO 2.5V
TPC 14. Closed-Loop Frequency Response vs. Supply
FREQUENCY (MHz)
100
10
1
0.1
0.01
1 50010 1000.1
0.01
G = +1
VS = 5V
OUTPUT RESISTANCE (⍀)
TPC 15. Output Resistance vs. Frequency
INPUT STEP (V p-p)
TIME (ns)
50
40
100.5 21 1.5
30
20
VS = 3V, 0.1%
VS = ⴞ5V, 0.1%
VS = 3V, 1%
VS = ⴞ5V, 1%
G = –1
TPC 16. Settling Time vs. Input Step
FREQUENCY (MHz)
–10
–40
–60
–80
–100
–20
–30
–50
–70
–90
–110 1 50010 1000.1
0.01
CMRR (dB)
VS = +3V AND ⴞ5V
TPC 17. CMRR vs. Frequency
LOAD CURRENT (mA)
OUTPUT SATURATION VOLTAGE (mV)
1000
10
0
0.001 0.01
100
0.1 1 10 10
0
V
OL
, –55ⴗC
+5V – V
OH
,
+125ⴗC
V
OL
, +125ⴗC
V
S
=
5V
+5V – V
OH
,
–55ⴗC
TPC 18. Output Saturation Voltage vs. Load Current
REV. B
AD8041
–9–
TEMPERATURE (
ⴗ
C)
8
5
2
–60 –40 –20 0 20 40 60 80 100 120
SUPPLY CURRENT (mA)
7
6
4
3
V
S
=
5V
V
S
=
ⴞ5V
V
S
=
3V
TPC 19. Supply Current vs. Temperature
FREQUENCY (MHz)
40
–20
–60
–100
–140
20
0
–40
–80
–120
–160 1 50010 1000.1
0.01
PSRR (dB)
VS = 5V
–PSRR
+PSRR
TPC 20. PSRR vs. Frequency
10
9
0
6
3
2
1
8
7
4
5
V
OUT
p-p (V)
FREQUENCY (MHz)
V
S
= ⴞ5V
R
L
= 2k⍀
0.1 1000110100
TPC 21. Output Voltage Swing vs. Frequency
SERIES RESISTANCE (⍀)
90
10
80
50
40
30
20
70
60
006010 20 30 40 50
VIN
100k⍀
RSERIES
CLOAD
1k⍀
20
ⴗ
PHASE
MARGIN 45
ⴗ
PHASE
MARGIN
VS = 5V
CAPACITIVE LOAD (pF)
TPC 22. Capacitive Load vs. Series Resistance
FREQUENCY (MHz)
5
4
–5
NORMALIZED OUTPUT (dB)
3
2
1
0
–1
–2
–3
–4
1 50010 100
G = +2
G = +10
G = +5
G = +2,
R
F
= 402⍀
V
S
= 5V
R
L
= 5k⍀ TO 2.5V
R
F
= 2k⍀
TPC 23. Frequency Response vs. Closed-Loop Gain
1.600V
1.550V
1.500V
1.450V
1.400V
1.425V
1.475V
1.525V
1.575V
10ns
50mV
VIN = 0.1V p-p
RL = 2k⍀
VS = 3V
G = +1
TPC 24. Pulse Response, V
S
= 3 V
REV. B–10–
AD8041
5V
4V
3V
2V
1V
0V 200
␮
s
1V 0.111V MIN
R
L
= 150⍀ TO 2.5V
4.840V MAX
a.
5V
4V
3V
2V
1V
0V 200
␮
s
1V
R
L
= 150⍀ TO GND
4.741V MAX
0.043V MIN
b.
TPC 25. Output Swing vs. Load Reference Voltage,
V
S
= 5 V, G = –1
4.5V
3.5V
2.5V
1.5V
0.5V 40ns1V
VS = 5V
G = +2
RL = 2k⍀
VIN = 1V p-p
TPC 26. One Volt Step Response, V
S
= 5 V, G = +2
2.60V
2.55V
2.50V
2.45V
2.40V
V
S
= 5V
G = +1
R
L
= 2k⍀
V
L
= 5pF
40ns
50mV
TPC 27. 100 mV Step Response, V
S
= 5 V, G = +1
3.0V
2.5V
2.0V
1.5V
1.0V
0.5V
0V 2␮s
500mV
VIN = 3V p-p
f = 0.1MHz
RL = 2k⍀
VS = 3V
G = –1
TPC 28. Output Swing, V
S
= 3 V, V
IN
= 3 V p-p
3.0V
2.5V
2.0V
1.5V
1.0V
0.5V
0V 2␮s
500mV
VIN = 2.8V p-p
f = 0.8MHz
RL = 2k⍀
VS = 3V
G = –1
TPC 29. Output Swing, V
S
= 3 V, V
IN
= 2.8 V p-p
REV. B
AD8041
–11–
Overdrive Recovery
Overdrive of an amplifier occurs when the output and/or input
range are exceeded. The amplifier must recover from this over-
drive condition. As shown in Figure 4, the AD8041 recovers
within 50 ns from negative overdrive and within 25 ns from
positive overdrive.
5.0V
2.5V
0V 40ns50mV
OUTPUT
INPUT
G = +2
V
S
= 5V
Figure 4. Overdrive Recovery
Circuit Description
The AD8041 is fabricated on Analog Devices’ proprietary
eXtra-Fast Complementary Bipolar (XFCB) process, which
enables the construction of PNP and NPN transistors with similar
f
T
in the 2 GHz to 4 GHz region. The process is dielectrically
isolated to eliminate the parasitic and latch-up problems caused
by junction isolation. These features allow the construction of
high frequency, low distortion amplifiers with low supply currents.
This design uses a differential output input stage to maximize
bandwidth and headroom (see Figure 5). The smaller signal
swings required on the first stage outputs (nodes S1P, S1N) reduce
the effect of nonlinear currents due to junction capacitances and
improve the distortion performance. With this design harmonic
distortion of better than –85 dB @ 1 MHz into 100 Ω with V
OUT
=
2 V p-p (Gain = +2) on a single 5 V supply is achieved.
The complementary common-emitter design of the output stage
provides excellent load drive without the need for emitter follow-
ers, thereby improving the output range of the device consider-
ably with respect to conventional op amps. High output drive
capability is provided by injecting all output stage predriver
currents directly into the bases of the output devices Q8 and
Q36. Biasing of Q8 and Q36 is accomplished by I8 and I5,
along with a common-mode feedback loop (not shown). This
circuit topology allows the AD8041 to drive 50 mA of output
current with the outputs within 0.5 V of the supply rails.
On the input side, the device can handle voltages from –0.2 V
below the negative rail to within 1.2 V of the positive rail. Exceed-
ing these values will not cause phase reversal; however, the
input ESD devices will begin to conduct if the input voltages
exceed the rails by greater than 0.5 V.
A “Nested Integrator” topology is used in the AD8041 (see
the small-signal schematic in Figure 6). The output stage can
be modeled as an ideal op amp with a single-pole response and
a unity-gain frequency set by transconductance g
m2
and
Capacitor C9. R1 is the output resistance of the input stage; g
m
is the input transconductance. C7 and C9 provide Miller com-
pensation for the overall op amp. The unity gain frequency will
occur at g
m
/C9. Solving the node equations for this circuit yields:
V
Vi
A
sR C A s g
C
OUT
m
=
++×
+
0
19 21 1 31
2
([( )])
where A0 = g
m
g
m2
R2 R1 (Open-Loop Gain of Op Amp)
A2 = g
m2
R2 (Open-Loop Gain of Output Stage)
The first pole in the denominator is the dominant pole of the
amplifier and occurs at about 180 Hz. This equals the input
stage output impedance R1 multiplied by the Miller-multiplied
value of C9. The second pole occurs at the unity-gain bandwidth
of the output stage, which is 250 MHz. This type of architecture
allows more open-loop gain and output drive to be obtained
than a standard two-stage architecture would allow.
Output Impedance
The low frequency open-loop output impedance of the common
emitter output stage used in this design is approximately 6.5 kΩ.
While this is significantly higher than a typical emitter follower
output stage, when connected with feedback, the output imped-
ance is reduced by the open-loop gain of the op amp. With
110 dB of open-loop gain, the output impedance is reduced
to less than 0.1 Ω. At higher frequencies, the output impedance
will rise as the open-loop gain of the op amp drops; however, the
output also becomes capacitive due to the integrator capacitors
C9 and C3. This prevents the output impedance from ever becom-
ing excessively high (see TPC 15), which can cause stability
problems when driving capacitive loads. In fact, the AD8041
has excellent cap-load drive capability for a high frequency op
amp. TPC 22 demonstrates that the AD8041exhibits a 45°
margin while driving a 20 pF direct capacitive load. In addition,
running the part at higher gains will also improve the capacitive
load drive capability of the op amp.
S1N
R21 R3
V
EE
Q11
Q3
I10
R26 R39
Q5
Q4
Q40
I7
R2R15
Q13 Q17
R5
C7
Q2
S1P
Q22Q7
Q21
Q24
R23 R27
I2 I3
I1
Q51
Q25 Q50
Q39
Q47
Q27
Q31
Q23
I9
I5
V
EE
V
CC
I8
Q36
Q8
V
OUT
C3
C9
V
CC
V
IN
P
V
IN
N
V
EE
Figure 5. AD8041 Simplified Schematic
REV. B–12–
AD8041
R2
C3
g
m2
V
OUT
R1
C9
g
m
Vi
S1N
S1P
C7
R1
g
m
Vi
Figure 6. Small Signal Schematic
Disable Operation
The AD8041 has an active-low disable pin, which can be used
to three-state the output of the part and also lower its supply
current. If the disable pin is left floating, the part is enabled and
will perform normally. If the disable pin is pulled to 2.5 V (min)
below the positive supply, output of the AD8041 will be disabled
and the nominal supply current will drop to less than 1.6 mA.
For best isolation, the disable pin should be pulled to as low a
voltage as possible; ideally, the negative supply rail.
The disable pin on the AD8041 allows it to be configured as a 2:1
mux as shown in Figure 7 and can be used to switch many types of
high speed signals. Higher order multiplexers can also be built.
The break-before-make switching time is approximately 50 ns to
disable the output and 300 ns to enable the output.
6
4
7
3
2
AD8041
330⍀
50⍀
10␮F
5V
330⍀
8
6
4
7
3
2
AD8041
330⍀
50⍀
10␮F
5V
330⍀
8
13 12 11 10
74HC04
50⍀
G = +2
G = +2
CH0
5MHz
CH1
10MHz
Figure 7. 2:1 Multiplexer
10
0%
100
90
200ns
1V
V
S
= 5V
Figure 8. 2:1 Multiplexer Performance
Single-Supply A/D Conversion
Figure 9 shows the AD8041 driving the analog inputs of the
AD9050 in a dc-coupled system with single-ended signals. All
components are powered from a single 5 V supply. The AD820
is used to offset the ground referenced input signal to the level
required by the AD9050. The AD8041 is used to add in the offset
with the ground referenced input signal and buffer the input to
AD9050. The nominal input range of the AD9050 is 2.8 V
and 3.8 V (1 V p-p centered at 3.3 V). This circuit provides
40 MSPS analog-to-digital conversion on just 330 mW of power
while delivering 10-bit performance.
0.1␮F
5V
AD8041 2.8V – 3.8V
3.3V
5V
AD9050
10
9
1k⍀
V
IN
–0.5V TO +0.5V
1k⍀1k⍀
0.1␮F5V
AD820
1k⍀
Figure 9. 10-Bit, 40 MSPS A/D Conversion
0
–10
–100
–60
–70
–80
–90
–40
–50
–30
–20
F1 = 4.9MHz
FUNDAMENTAL = 0.6dB
SECOND HARMONIC = 66.9dB
THIRD HARMONIC = 74.7dB
SNR = 55.2dB
NOISE FLOOR = – 86.1dB
ENCODE FREQUENCY = 40MHz
Figure 10. FFT Output of Circuit in Figure 9
REV. B
AD8041
–13–
APPLICATIONS
RGB Buffer
The AD8041 can provide buffering of RGB signals that include
ground while operating from a single 3 V or 5 V supply.
The signals that drive an RGB monitor are usually supplied by
current output DACs that operate from a 5 V only supply. These
can triple DACs like the ADV7120 and ADV7122 from Analog
Devices or integrate into the graphics controller IC as in most
PCs these days.
During the horizontal blanking interval, the currents output
from the DACs go to zero and the RGB signals are pulled to
ground via the termination resistors. If more than one RGB
monitor is desired, it cannot simply be connected in parallel
because it will provide an additional termination. Therefore,
buffering must be provided before connecting a second monitor.
Since the RGB signals include ground as part of their dynamic
output range, it has previously been required to use a dual-
supply op amp to provide this buffering. In some systems, this is
the only component that requires a negative supply, so it can be
quite inconvenient to incorporate this multiple monitor feature.
Figure 11 shows a schematic of one channel of a single-supply,
gain-of-two buffer for driving a second RGB monitor. No cur-
rent is required when the amplifier output is at ground. The
termination resistor at the monitor helps pull the output down
at low voltage levels.
6
4
7
3
2
AD8041
1k⍀
75⍀
10␮F
875⍀
R, G OR B NC
0.1␮F
1k⍀
PRIMARY RGB
MONITOR
75⍀
SECOND RGB
MONITOR
3V OR 5V
Figure 11. Single-Supply RGB Buffer
Figure 12 is an oscilloscope photo of the circuit in Figure 11
operating from a 3 V supply and driven by the blue signal of a
color bar pattern. Note that the input and output are at ground
during the horizontal blanking interval. The RGB signals are
specified to output a maximum of 700 mV peak. The output of
the AD8041 is 1.4 V with the termination resistors providing a
divide-by-two. The red and green signals can be buffered in the
same manner with duplication of this circuit.
VIN
GND
GND
VOUT
10
0%
100
90
5
␮
s500mV
500mV
Figure 12. 3 V, RGB Buffer
Single-Supply Composite Video Line Driver
Figure 13 shows a schematic of a single-supply gain-of-two
composite video line driver. Since the sync tips of a composite
video signal extend below ground, the input must be ac-coupled
and shifted positively to provide signal swing during these nega-
tive excursions in a single-supply configuration.
The input is terminated in 75 Ω and ac-coupled via C
IN
to a
voltage divider that provides the dc bias point to the input.
Setting the optimal bias point requires some understanding of
the nature of composite video signals and the video performance
of the AD8041.
Signals of bounded peak-to-peak amplitude that vary in duty
cycle require larger dynamic swing capability than their peak-to-
peak amplitude after ac coupling. As a worst case, the dynamic
signal swing required will approach twice the peak-to-peak value.
The two bounding cases are for a duty cycle that is mostly low,
but occasionally goes high at a fraction of a percent duty cycle
and vice versa.
Composite video is not quite this demanding. One bounding
extreme is for a signal that is mostly black for an entire frame
but has a white (full intensity), minimum width spike at least
once per frame.
The other extreme is for a video signal that is full white every-
where. The blanking intervals and sync tips of such a signal will
have negative going excursions in compliance with composite
video specifications. The combination of horizontal and vertical
blanking intervals limit such a signal to being at its highest level
(white) for only about 75% of the time.
As a result of the duty cycle variations between the two extremes
presented above, a 1 V p-p composite video signal that is multi-
plied by a gain of two requires about 3.2 V p-p of dynamic voltage
swing at the output for an op amp to pass a composite video
signal of arbitrary duty cycle without distortion.
Some circuits use a sync tip clamp along with ac coupling to
hold the sync tips at a relatively constant level in order to lower
the amount of dynamic signal swing required. However, these
circuits can have artifacts like sync tip compression unless they
are driven by sources with very low output impedance.
6
4
7
3
2
AD8041
R
F
1k⍀
10k⍀
10␮F
5V
75⍀
COMPOSITE
VIDEO IN
NC
0.1␮F
R
T
75⍀
8
1000␮F
0.1␮F
4.99k⍀
10␮F
4.99k⍀
47␮F
R
G
1k⍀
220␮F
75⍀
COAX
R
L
75⍀
V
OUT
Figure 13. Single-Supply Composite Video Line Driver
The AD8041 not only has ample signal swing capability to
handle the dynamic range required without using a sync tip
clamp but also has good video specifications like differential
gain and differential phase when buffering these signals in an ac-
coupled configuration.
REV. B–14–
AD8041
To test this, the differential gain and differential phase were
measured for the AD8041 while the supplies were varied. As the
lower supply is raised to approach the video signal, the first effect
to be observed is that the sync tips become compressed before
the differential gain and differential phase are adversely affected.
Thus, there must be adequate swing in the negative direction to
pass the sync tips without compression.
As the upper supply is lowered to approach the video, the differ-
ential gain and differential phase were not significantly adversely
affected until the difference between the peak video output and
the supply reached 0.6 V. Thus, the highest video level should
be kept at least 0.6 V below the positive supply rail.
Taking the above into account, it was found that the optimal
point to bias the noninverting input is at 2.2 V dc. Operating at
this point, the worst-case differential gain is measured at 0.06%
and the worst-case differential phase is 0.06°.
The ac coupling capacitors used in the circuit at first glance
appear quite large. A composite video signal has a lower fre-
quency band edge of 30 Hz. The resistances at the various ac
coupling points—especially at the output—are quite small. In
order to minimize phase shifts and baseline tilt, the large value
capacitors are required. For video system performance that is
not to be of the highest quality, the value of these capacitors can
be reduced by a factor of up to five with only a slightly observ-
able change in the picture quality.
Sync Stripper
Some RGB monitor systems use only three cables total and
carry the synchronizing signals along with the green (G) signal
on the same cable. The sync signals are pulses that go in the
negative direction from the blanking level of the G signal.
In some applications like prior to digitizing component video
signals with A/D converters, it is desirable to remove or strip the
sync portion from the G signal. Figure 14 is a schematic of a
circuit using the AD8041 running on a single 5 V supply that
performs this function.
AD8041
R2
1k⍀
10␮F
0.1␮F
0.8V
(2X V
BLANK
)
5V
75⍀
V
IN
75⍀
75⍀
(MONITOR)
R1
1k⍀
7
6
3
24
GREEN W/SYNC
V
BLANK
+0.4
GROUND
GREEN W/OUT SYNC
GROUND
Figure 14. Single-Supply Sync Stripper
Referring to Figure 15, the green plus sync signal is output
from an ADV7120, a single-supply triple video DAC. Because
the DAC is single supply, the lowest level of the sync tip is at
ground or slightly above. The AD8041 is set for a gain of two to
compensate for the divide by two of the output terminations.
10
0%
100
90
10
␮
s
500mV
500mV
Figure 15. Single-Supply Sync Stripper
The reference voltage for R1 should be twice the dc blanking
level of the G signal. If the blanking level is at ground and the
sync tip is negative as in some dual-supply systems, then R1 can
be tied to ground. In either case, the output will have the sync
removed and have the blanking level at ground.
Layout Considerations
The specified high speed performance of the AD8041 requires
careful attention to board layout and component selection.
Proper RF design techniques and low-pass parasitic component
selection are necessary.
The PCB should have a ground plane covering all unused portions
of the component side of the board to provide a low impedance
path. The ground plane should be removed from the area near
the input pins to reduce the stray capacitance.
Chip capacitors should be used for the supply bypassing.
One end should be connected to the ground plane and the other
within 1/8 inch of each power pin. An additional large (0.47 µF
to 10 µF) tantalum electrolytic capacitor should be connected in
parallel, but not necessarily so close, to supply current for fast,
large signal changes at the output.
The feedback resistor should be located close to the inverting
input pin in order to keep the stray capacitance at this node to a
minimum. Capacitance variations of less than 1 pF at the inverting
input will significantly affect high speed performance.
Stripline design techniques should be used for long signal traces
(greater than about 1 inch). These should be designed with a
characteristic impedance of 50 Ω or 75 Ω and be properly termi-
nated at each end.
REV. B
AD8041
–15–
OUTLINE DIMENSIONS
8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
Dimensions shown in inches and (millimeters)
SEATING
PLANE
0.180
(4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79) 0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
8
14
5
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.100 (2.54)
BSC
0.375 (9.53)
0.365 (9.27)
0.355 (9.02)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-095AA
0.015
(0.38)
MIN
8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
0.50 (0.0196)
0.25 (0.0099) ⴛ 45ⴗ
8ⴗ
0ⴗ
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
85
41
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
6.20 (0.2440)
5.80 (0.2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012AA
8-Lead Ceramic Dual In-Line Package [CERDIP]
(Q-8)
Dimensions shown in inches and (millimeters)
14
85
0.310 (7.87)
0.220 (5.59)
PIN 1
0.005 (0.13)
MIN 0.055 (1.40)
MAX
0.100 (2.54) BSC
15
0
0.320 (8.13)
0.290 (7.37)
0.015 (0.38)
0.008 (0.20)
SEATING
PLANE
0.200 (5.08)
MAX
0.405 (10.29) MAX
0.150 (3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36) 0.070 (1.78)
0.030 (0.76)
0.060 (1.52)
0.015 (0.38)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETERS DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
REV. B
C01058–0–6/03(B)
–16–
AD8041
Revision History
Location Page
5/03—Data Sheet changed from REV. A to REV. B.
Deleted all references to evaluation board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .Universal
Updated OUTLINES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
4/01—Data Sheet changed from REV. 0 to REV. A.
Specifications changed DISABLE CHARACTERISTICS, Off Voltage (Device Disabled) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2