MIC2169B
500kHz PWM Synchronous Buck
Control IC
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
April 2010 M9999-041210-B
General Description
The MIC2169B is a high-efficiency, simple to use 500kHz
PWM synchronous buck control IC housed in small
MSOP-10 and MSOP-10 ePad packages. The MIC2169B
allows compact DC/DC solutions with a minimal external
component count and cost. The device features high-
output driver capability to drive loads up to 30A.
The MIC2169B operates from a 3V to 14.5V input, without
the need of any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range within the smallest possible
printed circuit board space area.
The MIC2169B senses current across the high-side N-
Channel MOSFET, eliminating the need for an expensive
and lossy current-sense resistor. Current-limit accuracy is
maintained by a positive temperature coefficient that tracks
the increasing RDS(ON) of the external MOSFET. Further
cost and space are saved by the internal in-rush current-
limiting digital soft-start. The MIC2169B is identical to the
MIC2169A with the exception that the MIC2169B supports
pre-bias loads and has a lower impedance gate-drive
circuit. Internal pre-bias circuit prevents output voltage
drooping and excessive reverse inductor current when
powering up with a pre-bias voltage at the output.
The MIC2169B is available in a 10-pin MSOP and a
thermally-capable 10-pin ePad MSOP package, with a
wide junction operating range of -40°C to +125°C.
All support documentation can be found on Micrel’s web
site at www.micrel.com.
Features
• 3V to 14.5V input voltage range
• Adjustable output voltage down to 0.8V
• 500kHz PWM operation
• Up to 95% efficiency
• Output Pre-biased Protection
• Build-in 2.2 drivers to drive two n-channel MOSFETs
• Adaptive gate drive increases efficiency
• Simple, externally-compensated voltage-mode PWM
control
• Short minimum ON time of 30ns allowing very-low duty
cycle
• Fast transient response
• Adjustable current limit senses high-side N-Channel
MOSFET current
• Hiccup mode short-circuit protection
• No external current-sense resistor
• Internal soft-start current source
• Dual function COMP and EN pin allows low-power
shutdown
• Available in small-size 10-pin MSOP and 10-pin MSOP
ePad packages
Applications
• Point-of-load DC/DC conversion
• High-Current Power Supplies
• Telecom/Datacom and Networking Power Supplies
• Servers and Workstations
• Graphic cards and other PC Peripherals
• Set-top boxes
• LCD power supplies
___________________________________________________________________________________________________________
Micrel, Inc. MIC2169B
April 2010 2 M9999-041210-B
Typical Application
1.0µH 3.3V
VIN = 5V to 12V
VDD
COMP/EN
VIN
CS
FB
GND EP
LSD
BST
10µF
100µF
0.1µF
0.1µF
100nF IRF7821
SD103BWS
IRF7821
150pF
HSD
VSW
MIC2169B
1µF 1000pF
330µF x 2
10µF
50
55
60
65
70
75
80
85
90
95
100
0246810121416
EFFICIENCY (%)
IL
O
AD (A)
MIC2169B Efficienc
VIN =5V
VOUT =3.3V
MIC2169B Adjustable Output 500kHz Converter
Micrel, Inc. MIC2169B
April 2010 3 M9999-041210-B
Ordering Information
Part Number Frequency Junction Temperature Range(1) Package Lead Finish
MIC2169BYMME 500kHz -40° to +125°C 10-Lead ePad MSOP Pb-Free
MIC2169BYMM 500kHz -40° to +125°C 10-Lead MSOP Pb-Free
Pin Configur ation
10-Pin ePad MSOP (MME) 10-Pin MSOP
Pin Description
Pin Number Pin Name Pin Function
1 VIN Supply Voltage (Input): +3V to +14.5V.
2 VDD
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate-drive
supply voltage and an internal supply bus for the IC. When VIN is <5V, short
VDD to the input supply through a 10 resistor.
3 CS
Current Sense (Input): Current-limit comparator noninverting input. The current
limit is sensed across the MOSFET during the ON time. The current can be set
by the resistor in series with the CS pin.
4 COMP/EN
Compensation / Enable (Input): Dual function pin. Pin for external compensation.
If this pin is pulled below 0.25V, with the reference fully up the device shuts
down (50A typical current draw).
5 FB Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
6 GND Ground (Return).
7 LSD
Low-Side Drive (Output): High-current driver output for external synchronous
MOSFET.
8 VSW Switch (Return): High-side MOSFET driver return.
9 HSD
High-Side Drive (Output): High-current output-driver for the high-side MOSFET.
When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be used. At
VIN > 5V, 4.5V threshold MOSFETs should be used.
10 BST
Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The
gate-drive voltage is higher than the source voltage by VDD minus a diode drop.
ePad EP Connect to Ground.
Micrel, Inc. MIC2169B
April 2010 4 M9999-041210-B
Absolute Maximum Ratings(1)
Supply Voltage (VIN)...................................... -0.3V to 15.5V
Booststrapped Voltage (VBST) .................... -0.3V to VIN +6V
VSW .............................................................. -0.3V to 15.5V
CS ............................................................................15.25V
LSD,FB............................................................... -0.3V to 6V
Storage Temperature (TS)..........................-65°C to +150°C
Peak Reflow Temperature (10 to 20 sec) ................ +260°C
ESD (HBM) (3)................................................................. 2kV
ESD (MM).....................................................................200V
Operating Ratings(2)
Supply Voltage (VIN)...................................... +3V to +14.5V
Ambient Temperature (TA) ...........................-40°C to +85°C
Junction Temperature (TJ) ..........................-40°C to+125°C
Junction Thermal Resistance
ePad MSOP (JA)............................................76.7°C/W
ePad MSOP (JC) .............................................9.6°C/W
MSOP (JA) ......................................................130°C/W
MSOP (JC).....................................................42.6°C/W
Output Voltage Range............................. 0.8V to VIN × DMAX
Electrical Characteristics(4)
TJ = 25°C, VIN = 5V; Bold values indicate –40°C TJ +125°C; unless otherwise specified.
Parameter Condition Min Typ Max Units
Feedback Voltage Reference (±1%) 0.792 0.8 0.808 V
Feedback Voltage Reference (±2% over temp) 0.784 0.8 0.816 V
Feedback Bias Current 150 350 nA
Output Voltage Line
Regulation
0.03 % / V
Output Voltage Load
Regulation
0.5 %
Output Voltage Total
Regulation
3V VIN 14.5V; 1A IOUT 10A; (VOUT = 2.5V)(4) 0.6
1.5 %
Oscillator Section
Oscillator Frequency 450 500 550 kHz
Maximum Duty Cycle 92 %
Minimum On-Time(5) 30
60 ns
Input and VDD Supply
PWM Mode Supply Current VCS = VIN –0.25V; VFB = 0.7V (output switching but
excluding external MOSFET gate current.)
1.5 3 mA
Shutdown Quiescent Current VCOMP/EN = 0V 50
150 µA
VCOMP Shutdown Threshold 0.1 0.25 0.35 V
VCOMP Shutdown Blanking
Period
CCOMP = 100nF 675 μs
Digital Supply Voltage (VDD) VIN 6V 4.7 5 5.3 V
Micrel, Inc. MIC2169B
April 2010 5 M9999-041210-B
Electrical Characteristics(4) (continued)
TJ = 25°C, VIN = 5V; Bold values indicate –40°C TJ +125°C; unless otherwise specified.
Parameter Condition Min Typ Max Units
Error Amplifier
DC Gain(5) 70 dB
Transconductance 1.1 mΩ–1
Soft-Start
Soft-Start Current After time out of internal timer. VCOMP = 0.8V 4 8.5 13 µA
Current Sense
CS Over Current Trip Point VCS = VIN –0.25V 160 200 240 µA
Temperature Coefficient 1800 ppm/°C
Gate Drivers
Rise/Fall Time Into 3000pF at VIN > 5V 15 ns
Source, VIN = 4.5V 2.2 3
Sink, VIN = 4.5V 1.3 3
Source, VIN = 3V 2.7 4
Output Driver Impedance
Sink, VIN = 3V 1.7 4
Driver Non-Overlap Time(5) 50 ns
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max),
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive, handling precautions required.
4. Specification for packaged product only.
5. Guaranteed by design.
Micrel, Inc. MIC2169B
April 2010 6 M9999-041210-B
Typical Characteristics
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
2.1
2.3
2.5
2.7
2.9
IDD
(
mA
)
TEMPERATURE (°C)
PWM Mode Supply Current
vs. Temperature
-40 -20 0 20 40 60 80 10012014
0
0.5
1.0
1.5
2.0
0 5 10 15
QUIESCENT CURRENT (mA)
SUPPLY VOLTAGE (V)
PWM Mode Suppl
y
Current
vs. Suppl Voltage
0.7980
0.7985
0.7990
0.7995
0.8000
0.8005
0.8010
0 5 10 15
VFB (V)
VIN (V)
V
FB Line Regulation
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
-60 -30 0 30 60 90 120 150
VFB (V)
TEMPERATURE (°C)
V
FB vs. Temperature
0
1
2
3
4
5
6
0 5 10 15
VDD (V)
VIN (V)
V
DD Line Regulation
4.90
4.92
4.94
4.96
4.98
5.00
5.02
0 5 10 15 20 25 30
VDD REGULATOR VOLTAGE (V)
LOAD CURRENT (mA)
V
DD Load Regulation
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
-60 -30 0 30 60 90 120 150
VDD LINE REGULATION (%)
TEMPERATURE
(
°C
)
DD L
i
ne
R
egu
l
at
i
on
vs. Temperature
Oscillator Frequency
vs. Temperature
450
460
470
480
490
500
510
520
530
540
550
-60 -30 0 30 60 90 120 150
TEM PERATURE (°C)
FREQ UE NCY ( k Hz )
-1.5
-1.0
-0.5
0
0.5
1.0
1.5
0 5 10 15
FREQUENCY VARIATION (%)
VIN (V)
Oscillator Frequency
vs. Suppl Voltage
Micrel, Inc. MIC2169B
April 2010 7 M9999-041210-B
Typical Characteristics (continued)
100
120
140
160
180
200
220
240
260
ICS(μA)
TEMPERATURE (°C)
O
v
e
r
current T
r
ip Point
vs. Temperature
-60 -30 0 30 60 90 120 150
Functional Diagram
MIC2169B Block Diagram
Micrel, Inc. MIC2169B
April 2010 8 M9999-041210-B
Functional Description
The MIC2169B is a voltage-mode, synchronous step-
down switching regulator controller designed for high
power. Current limit is implemented without the use of an
external sense resistor. It includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time, a PWM generator, a reference voltage, two
MOSFET drivers, and short-circuit current limiting
circuitry to form a complete 500kHz switching regulator.
MIC2169B is identical to the MIC2169A except it
supports pre-bias loads and has a lower impedance
gate-drive circuit.
Theory of Operation
The MIC2169B is a voltage mode step-down regulator.
The figure above illustrates the block diagram for the
voltage control loop. The output voltage variation due to
load or line changes will be sensed by the inverting input
of the transconductance error amplifier via the feedback
resistors R3, and R2 and compared to a reference
voltage at the non-inverting input. This will cause a small
change in the DC voltage level at the output of the error
amplifier which is the input to the PWM comparator. The
other input to the comparator is a 0.95V to 1.45V
triangular waveform. The comparator generates a
rectangular waveform whose width tON is equal to the
time from the start of the clock cycle t0 until t1, the time
the triangle crosses the output waveform of the error
amplifier. To illustrate the control loop, let us assume the
output voltage drops due to sudden load turn-on, this
would cause the inverting input of the error amplifier,
which is divided down version of VOUT, to be slightly less
than the reference voltage, causing the output voltage of
the error amplifier to go high. This will cause the PWM
comparator to increase tON time of the top side
MOSFET, causing the output voltage to go up and
bringing VOUT back in regulation.
Soft-Start
The COMP/EN pin on the MIC2169B is used for the
following three functions:
1. Disables the part by grounding this pin
2. External compensation to stabilize the voltage
control loop
3. Soft-start
For better understanding of the soft-start feature,
assume VIN = 12V, and the MIC2169B is allowed to
power-up by un-grounding the COMP/EN pin. The
COMP pin has an internal 8.5µA current source that
charges the external compensation capacitor. As soon
as this voltage rises to 250mV (t = Cap_COMP ×
0.25V/8.5µA) and VIN crosses the 2.6V UVLO threshold,
the MIC2169B allows the internal VDD linear regulator to
power up, and the chip’s internal oscillator starts
switching. At this point in time, the COMP pin current
source increases to 40µA and an internal 12-bit counter
starts counting which takes approximately 2ms to
complete. During counting, the COMP voltage is
clamped at 0.65V. After this counting cycle the COMP
current source is reduced to 8.5µA and the COMP pin
voltage rises from 0.65V to 0.95V, the bottom edge of
the saw-tooth oscillator. This is the beginning of 0% duty
cycle and it increases slowly causing the output voltage
to rise slowly. The MIC2169B has one hysteretic
comparator whose output is asserted high when VOUT is
within -3% of steady state. When the output voltage
reaches 97% of programmed output voltage then the gm
error amplifier is enabled along with the hysteretic
comparator output is asserted high. This point onwards,
the voltage control loop (gm error amplifier) is fully in
control and will regulate the output voltage.
Soft-start time can be calculated approximately by
adding the following four time frames:
t1 = Cap_COMP × 0.25V/8.5µA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5µA
µA5.8
COMP_Cap
5.0
V
V
4t
IN
OUT ××
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=
Soft-Start Time(Cap_COMP=100nF) = t1 + t2
+ t3 + t4 = 2.9ms + 2ms + 3.5ms + 1.6ms =
10ms
Current Limit
The MIC2169B uses the RDS(ON) of the top power
MOSFET to measure output current. Since it uses the
drain to source resistance of the power MOSFET, it is
not very accurate. This scheme is adequate to protect
the power supply and external components during a fault
condition by cutting back the time the top MOSFET is on
if the feedback voltage is greater than 0.67V. In case of
a hard short when feedback voltage is less than 0.67V,
the MIC2169B discharges the COMP capacitor to 0.65V,
resets the digital counter and automatically shuts off the
top gate drive, the gm error amplifier is completely
disabled, the –3% hysteretic comparators is asserted
low, and the soft-start cycles restart from t2 to t4. This
mode of operation is called the “hiccup mode” and its
purpose is to protect the down stream load in case of a
hard short. The circuit in Figure 1 illustrates the
MIC2169B current limiting circuit.
Micrel, Inc. MIC2169B
April 2010 9 M9999-041210-B
L1 Inductor
VIN
VOUT
HSD
LSD
RCS
CSVSW
200 A
C2
CIN
C1
COUT
Q1
MOSFET N
Q2
MOSFET N
0.1µF
1000pF
Figure 1. The MIC2169B Current Limiting Circuit
The current limiting resistor RCS is calculated by the
following equation:
µA200
IR
RL1Q)ON(DS
CS
×
=
where:
2
Current Ripple Inductor
II LOADL +=
Inductor Ripple Current =
()
LFV
VV
V
SIN
OUTIN
OUT ××
−
×
FS = 500kHz
200µA is the internal sink current to program the
MIC2169B current limit.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to the load current (ILOAD) in the above equation
to avoid false current limiting due to increased MOSFET
junction temperature rise. It is also recommended to
connect RCS resistor directly to the drain of the top
MOSFET Q1, and the RSW resistor to the source of Q1 to
accurately sense the MOSFETs RDS(ON). To make the
MIC2169B insensitive to board layout and noise
generated by the switch node, a 1.4 resistor and a
1000pF capacitor is recommended between the switch
node and GND.
Internal VDD Supply
The MIC2169B controller internally generates VDD for
self biasing and to provide power to the gate drives. This
VDD supply is generated through a low-dropout regulator
and generates 5V from VIN supply greater than 5V. For
supply voltage less than 5V, the VDD linear regulator is
approximately 200mV in dropout. Therefore, it is
recommended to short the VDD supply to the input supply
through a 10 resistor for input supplies between 3.0V
to 5V.
MOSFET Gate Drive
The MIC2169B high-side drive circuit is designed to
switch an N-Channel MOSFET. The Functional Block
Diagram shows a bootstrap circuit, consisting of D1 and
CBST, supplies energy to the high-side drive circuit.
Capacitor CBST is charged while the low-side MOSFET is
on and the voltage on the VSW pin is approximately 0V.
When the high-side MOSFET driver is turned on, energy
from CBST is used to turn the MOSFET on. As the
MOSFET turns on, the voltage on the VSW pin
increases to approximately VIN. Diode D1 is reversed
biased and CBST floats high while continuing to keep the
high-side MOSFET on. When the low-side switch is
turned back on, CBST is recharged through D1. The drive
voltage is derived from the internal 5V VDD bias supply.
The nominal low-side gate drive voltage is 5V and the
nominal high-side gate drive voltage is approximately
4.5V due the voltage drop across D1. An approximate
50ns delay between the high-side and low-side driver
transitions is used to prevent current from
simultaneously flowing unimpeded through both
MOSFETs (shoot-through).
Adaptive gate drive is implemented on the high-side (off)
to low-side (on) driver transition to reduce losses in the
flywheel diode and to prevent shoot-through. This is
operated by detecting the VSW pin; once this pin is
detected to reach 1.5V, the high-side MOSFET can be
assumed to be off and the low side driver is enabled.
Total Power Dissipation and Thermal Considerations
Total power dissipation in the MIC2169B equals the
power dissipation caused by driving the external
MOSFETs plus the quiescent supply current:
PdissTOTAL = PdissSUPPLY + PdissDRIVE
where:
PdissSUPPLY = VDD × IDD
IDD is shown in the “PWM Mode Supply Current” graph in
the Typical Characteristics section of the specification.
PdissDRIVE calculations are shown in the Applications
section of the specification.
The die temperature may be calculated once the total
power dissipation is known:
TJ = TA + PdissTOTAL × θJA
where:
TA is the maximum ambient temperature (°C)
TJ is the junction temperature (°C)
PdissTOTAL is the power dissipation of the
MIC2169B (W)
JC is the thermal resistance from junction-to-
ambient air (°C/W)
Micrel, Inc. MIC2169B
April 2010 10 M9999-041210-B
The following graphs are used to determine the
maximum gate charge that can be driven with respect to
supply voltage and ambient temperature. Figure 2 shows
the power dissipation in the driver for different values of
gate charge.
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0 20406080100
GATE CHARGE (nC)
P OWER DISSI P AT IO N (W)
5Vin
12Vin
Figure 2. Power Dissipation vs. Total Gate Charge
Figure 3 shows the maximum allowable power
dissipation vs ambient temperature. For a given total
gate charge, the maximum operating ambient
temperature can be found by using the two graphs.
0
20
40
60
80
100
120
140
0.0 0.2 0.4 0.6 0.8 1.0 1.2
PO WER DISSIPATION (W)
MAXIMUM AMBIE NT TEMPERATURE (°C
)
MSOP
ePAD
MSOP
Figure 3. Maximum Ambient Temp erature vs.
Power Dissipation
Figures 4 and 5 show the increase in junction and case
temperature for a given power dissipation.
0
10
20
30
40
50
60
0.00.20.40.60.81.01.2
P OWER DI S SI PAT IO N (W)
CAS E TEM PE RATURE RISE (°C
)
MSOP
ePAD
MSOP
Figure 4. Case Temperature Rise vs.
Power Dissipation
0
20
40
60
80
100
120
0.0 0.2 0.4 0.6 0.8 1.0 1.2
POWER DIS SIPATION (W)
JUNCTION TEMP ERATURE RISE (°C
)
MSOP
ePAD
MSOP
Figure 5. Junction Temperature Rise vs.
Power Dissipation
Micrel, Inc. MIC2169B
April 2010 11 M9999-041210-B
Application Information
MOSFET Selection
The MIC2169B controller works from input voltages of
3V to 14.5V and has an internal 5V regulator to provide
power to turn the external N-Channel power MOSFETs
for high- and low-side switches. For applications where
VIN < 5V, the internal VDD regulator operates in dropout
mode, and it is necessary that the power MOSFETs
used are sub-logic level and are in full conduction mode
for VGS of 2.5V. For applications when VIN > 5V; logic-
level MOSFETs, whose operation is specified at VGS =
4.5V must be used. For the lower (<5V) applications, the
VDD supply can be connected directly to VIN to help
increase the driver voltage to the MOSFET.
It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in
junction temperature will increase the channel resistance
of the MOSFET by 50% to 75% of the resistance
specified at 25°C. This change in resistance must be
accounted for when calculating MOSFET power
dissipation and in calculating the value of current-sense
(CS) resistor. Total gate charge is the charge required to
turn the MOSFET on and off under specified operating
conditions (VDS and VGS). The gate charge is supplied by
the MIC2169B gate-drive circuit. At 500kHz switching
frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2169B.
At low output load, this power dissipation is noticeable
as a reduction in efficiency. The average current
required to drive the high-side MOSFET is:
SG)avg](sidehigh[G fQI ×=
−
where:
IG[high-side](avg) = average high-side MOSFET gate
current.
QG = total gate charge for the high-side MOSFET
taken from manufacturer’s data sheet for VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching loss for the low-side MOSFET is
usually negligible. Also, the gate-drive current for the
low-side MOSFET is more accurately calculated using
CISS at VDS = 0 instead of gate charge.
For the low-side MOSFET:
SGSISS)avg](sidelow[G fVCI ××=
−
Since the current from the gate drive comes from the
input voltage, the power dissipated in the MIC2169B due
to gate drive is:
(
)
)avg](sidelow[G)avg](sidehigh[GINGATEDRIVE IIVP −− +×=
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON)×QG.
Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2169B.
Parameters that are important to MOSFET switch
selection are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of
20% should be added to the VDS(max) of the MOSFETs
to account for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the
sum of the conduction losses during the on-time
(PCONDUCTION) and the switching losses that occur during
the period of time when the MOSFETs turn on and off
(PAC).
ACCONDUCTIONSW PPP +
=
where:
)on(AC)off(ACAC
SW
2
)rms(SWCONDUCTION
PPP
RIP
+=
×=
R
SW = on-resistance of the MOSFET switch
D = duty cycle = ⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
IN
O
V
V
Making the assumption the turn-on and turn-off transition
times are equal; the transition times can be
approximated by:
G
INOSSGSISS
TI
VCVC
t×+×
=
where:
C
ISS and COSS are measured at VDS = 0
I
G = gate-drive current (1.4A for the MIC2169B)
The total high-side MOSFET switching loss is:
(
)
STPKDINAC ftIVVP ××
×
+
=
where:
t
T = switching transition time (typically 20ns to
50ns)
V
D = freewheeling diode drop, typically 0.5V
f
S it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible
and can be ignored for these calculations.
Micrel, Inc. MIC2169B
April 2010 12 M9999-041210-B
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below.
(
)
(max)OUTS(max)IN
OUT(max)INOUT
I2.0fV
VVV
L×××
−×
=
where:
f
S = switching frequency, 500kHz
0.2 = ratio of AC ripple current to DC output
current
V
IN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
(
)
LfV
VVV
I
S(max)IN
OUT(max)INOUT
PP ××
−×
=
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor ripple
current.
PP(max)OUTPK I5.0II ×+=
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
12
I
)I(I
2
PP
2
MAX_OUTINDUCTOR +=
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2169B requires the
use of ferrite materials for all but the most cost sensitive
applications.
Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the
power supply. This is especially noticeable at low output
power. The winding resistance decreases efficiency at
the higher output current levels. The winding resistance
must be minimized although this usually comes at the
expense of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
currents, the core losses can be a significant contributor.
Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is
calculated by the equation below:
WINDING
2
ms)INDUCTOR(rINDUCTORC RIP U×=
The resistance of the copper wire, RWINDING, increases
with temperature. The value of the winding resistance
used should be at the operating temperature.
)TT(0042.01(RR C20HOT)C20(WINDING)hot(WINDING °°
−
×
+×
=
where:
THOT = temperature of the wire under operating
load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The output capacitor values are usually determined by
the capacitors ESR (equivalent series resistance).
Voltage and RMS current capability are two other
important factors selecting the output capacitor.
Recommended capacitors are tantalum, low-ESR
aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output
ripple. The output capacitor ESR also affects the overall
voltage feedback loop from stability point of view. See
“Feedback Loop Compensation” section for more
information. The maximum value of ESR is calculated:
PP
OUT
ESR I
V
RΔ
≤
where:
VOUT = peak-to-peak output voltage ripple
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ripple due
to the output capacitors’ ESR and the ripple due to the
output capacitor. The total ripple is calculated below:
()()
2
ESRPP
2
SOUT
PP
OUT RI
fC
D1I
V×+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
×
−×
=Δ
where:
D = duty cycle
COUT = output capacitance value
fS = switching frequency
Micrel, Inc. MIC2169B
April 2010 13 M9999-041210-B
The voltage rating of capacitor should be twice the
voltage for a tantalum and 20% greater for aluminum
electrolytic.
The output capacitor RMS current is calculated below:
12
I
IPP
C)rms(OUT =
The power dissipated in the output capacitor is:
()
)C(ESR
2
C)C(DISS OUT)rms(OUTOUT RIP ×=
Input Capacitor Selection
The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may
fail when subjected to high inrush currents, caused by
turning the input supply on. A tantalum input capacitor’s
voltage rating should be at least 2 times the maximum
input voltage to maximize reliability. Aluminum
electrolytic, OS-CON, and multilayer polymer film
capacitors can handle the higher inrush currents without
voltage derating. The input voltage ripple will primarily
depend on the input capacitor’s ESR. The peak input
current is equal to the peak inductor current, so:
)C(ESR)peak(INDUCTORIN IN
RIV ×=Δ
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor ripple current is low:
()
D1DII (max)OUTC )rms(IN −××≈
The power dissipated in the input capacitor is:
()
)C(ESR
2
)rms(CIN)C(DISS ININ RIP ×=
Voltage Setting Components
The MIC2169B requires two resistors to set the output
voltage as shown in Figure 6.
Error
Amp
5
MIC2169B
FB
VREF
0.8V
R2
R1
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation:
⎟
⎠
⎞
⎜
⎝
⎛+×= 2R
1R
1VV REFO
where
VREF for the MIC2169B is typically 0.8V
A typical value of R1 can be between 3kΩ and 10kΩ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, in value, it
will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
REFO
REF
VV
1RV
2R −
×
=
External Schottky Diode
An external freewheeling diode is used to keep the
inductor current flow continuous while both MOSFETs
are turned off. This dead time prevents current from
flowing unimpeded through both MOSFETs and is
typically 50ns. The diode conducts twice during each
switching cycle. Although the average current through
this diode is small, the diode must be able to handle the
peak current.
ID(avg) = IOUT × 2 × 50ns × fS
The reverse voltage requirement of the diode is:
VDIODE(rrm) = VIN
The power dissipated by the Schottky diode is:
PDIODE = ID(avg) × VF
where:
VF = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for
circuit operation since the low-side MOSFET contains a
parasitic body diode. The external diode will improve
efficiency and decrease high frequency noise. If the
MOSFET body diode is used, it must be rated to handle
the peak and average current. The body diode has a
relatively slow reverse recovery time and a relatively
high forward voltage drop. The power lost in the diode is
proportional to the forward voltage drop of the diode. As
the high-side MOSFET starts to turn on, the body diode
becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts
at a lower forward voltage preventing the body diode in
the MOSFET from turning on. The lower forward voltage
drop dissipates less power than the body diode. The lack
of a reverse recovery mechanism in a Schottky diode
causes less ringing and less power loss.
Micrel, Inc. MIC2169B
April 2010 14 M9999-041210-B
Depending on the circuit components and operating
conditions, an external Schottky diode will give a ½% to
1% improvement in efficiency.
Feedback Loop Compensation
The MIC2169B controller comes with an internal
transconductance error amplifier used for compensating
the voltage feedback loop by placing a capacitor (C1) in
series with a resistor (R1) and another capacitor C2 in
parallel from the COMP pin to ground. See “Functional
Block Diagram.”
Power Stage
The power stage of a voltage mode controller has an
inductor, L1, with its winding resistance (DCR)
connected to the output capacitor, COUT, with its
electrical series resistance (ESR) as shown in Figure 7.
The transfer function G(s), for such a system is:
ESR
COUT
VO
DCRL
Figure 7. The Output LC Filter in a Voltage-Mode Buck
Converter
()
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
××++××+××
××+
=
CsESR1CLsCsDCR
CsESR1
)s(G 2
Plotting this transfer function with the following assumed
values (L=1μH, DCR=0.009Ω, COUT=660μF,
ESR=0.025Ω) gives lot of insight as to why one needs to
compensate the loop by adding resistor and capacitors
on the COMP pin. Figures 8 and 9 show the gain curve
and phase curve for the above transfer function.
Figure 8. The Gain Curve for G(s)
Figure 9. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the
gain curve that the output inductor and capacitor create
a two pole system with a break frequency at:
OUT
LC CL2
1
f×π×
=
Therefore, fLC = 6.2kHz
By looking at the phase curve, it can be seen that the
output capacitor ESR (0.025Ω) cancels one of the two
poles (LCOUT) system by introducing a zero at:
OUT
ZERO CESR2
1
f××π×
=
Therefore, FZERO = 9.6kHz.
From the point of view of compensating the voltage loop,
it is recommended to use higher ESR output capacitors
since they provide a 90° phase gain in the power path.
For comparison purposes, Figure 10, shows the same
phase curve with an ESR value of 0.002Ω.
Figure 10. The Ph ase Curve with ESR = 0.002Ω
Micrel, Inc. MIC2169B
April 2010 15 M9999-041210-B
It can be seen from Figure 9 that at 50kHz, the phase is
approximately –90° versus Figure 10 where the number
is –150°. This means that the transconductance error
amplifier has to provide a phase boost of about 45° to
achieve a closed loop phase margin of 45° at a
crossover frequency of 50kHz for Figure 9, versus 105°
for Figure 10. The simple RC and C2 compensation
scheme allows a maximum error amplifier phase boost
of about 90°. Therefore, it is easier to stabilize the
MIC2169B voltage control loop by using high-ESR value
output capacitors.
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would
be picked up and transmitted at large amplitude to the
output, thus, gain should be permitted to fall off at high
frequencies. At low frequency, it is desired to have high
open-loop gain to attenuate the power line ripple. Thus,
the error amplifier gain should be allowed to increase
rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal
gm error amplifier can be approximated by the following
equation:
()
⎥
⎥
⎥
⎥
⎦
⎤
⎢
⎢
⎢
⎢
⎣
⎡
⎟
⎠
⎞
⎜
⎝
⎛
+
×
××+×+×
××+
×=
C2C1
C2C1
R1s1C2C1s
C1R1s1
gm(z) AmplifierError
The above equation can be simplified by assuming
C2<<C1,
⎥
⎦
⎤
⎢
⎣
⎡
××+××
××+
×= C2)R1s(1C1s
C1R1s1
gm(z) AmplifierError
From the above transfer function, one can see that R1
and C1 introduce a zero and R1 and C2 a pole at the
following frequencies:
FZERO= 1/2 π × R1 × C1
FPOLE = 1/2 π × C2 × R1
FPOLE@origin = 1/2 π × C1
Figures 11 and 12 show the gain and phase curves for
the above transfer function with R1 = 4.02k, C1 = 100nF,
C2 = 150pF, and gm = 1.1mΩ–1.
Figure 11. Error Amplifier Gain Curve
Figure 12. Erro r Amplifier Phase Curve
Total Open-Loop Response
The open-loop response for the MIC2169B controller is
easily obtained by adding the power path and the error
amplifier gains together, since they already are in Log
scale. It is desirable to have the gain curve intersect zero
dB at tens of kilohertz, this is commonly called crossover
frequency; the phase margin at crossover frequency
should be at least 45°. Phase margins of 30° or less
cause the power supply to have substantial ringing when
subjected to transients, and have little tolerance for
component or environmental variations.
Figures 13 and 14 show the open-loop gain and phase
margin for the 5V input and 1.8V output application, and
it can be seen from Figure 13 that the gain curve
intersects the 0dB at approximately 50kHz, and from
Figure 14 that at 50kHz, the phase shows approximately
74° of margin.
Micrel, Inc. MIC2169B
April 2010 16 M9999-041210-B
Figure 13.Open-Loop Gain Margin
Figure 14.Open-Loop Phase Margin
Pre-Biased Loads
The MIC2169B supports pre-biased loads. Some
applications have a pre-existing voltage on the output.
This pre-existing or pre-biased load is generated by an
external supply (other than the MIC2169B). During
startup without pre-bias support, MIC2169A will pull the
output voltage to ground through the inductor and low
side FET (see Figure 15).
The MIC2169B prevents the current sinking of any pre-
existing voltage source at the output (see Figure 16). It
does this by keeping the low-side FET off during the soft
start period. In some applications this pre-bias current
sink is not a problem, and the MIC2169A may be used.
In some applications the pre-bias current sink may
cause a problem, and the MIC2169B should be used.
The MIC2169B can support up to 90% of a pre-bias
condition (up to 90% of the final regulated output
voltage) see Figure 17.
Figure 15. MIC2169A Startup without Pre-Bias Support
Figure 15 shows MIC2169B startup with a pre-bias of 1V
on the output, in which the pre-existing output voltage
discharges during soft start.
Figure 16. MIC2169B Startup with Pre-Bias Support
Figure 16 shows MIC2169B startup with a pre-bias of 1V
on the output, in which the pre-existing output voltage
has no discharge.
Micrel, Inc. MIC2169B
April 2010 17 M9999-041210-B
Figure 17. MIC2169B Startup with Pre-Bias Support,
Pre-Bias At 90% of VOUT_FINAL
Figure 17 shows MIC2169B startup with a pre-bias of
2.2V on the output (90% of VOUT) without the pre-existing
output voltage discharge.
Design and PCB Lay out Guideline
WARNING!!! TO MINIMIZE EMI AND OUTP UT N OISE,
FOLLOW THESE LAYOUT RECOMMENDATIONS:
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2169B converter.
IC
• Place the IC and MOSFETs close to the point of
load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
• Place the VIN input capacitor next.
• Place the VIN input capacitors on the same side of
the board and as close to the MOSFETs as
possible.
• Keep both the VIN and power GND connections
short.
• Place several vias to the ground plane close to the
VIN input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
• An additional Tantalum or Electrolytic bypass input
capacitor of 22uF or higher is required at the input
power connection.
• Use a 5 resistor from the input supply to the VDD
pin on the MIC2169B. Also, place a 1µF ceramic
capacitor from this pin to GND, preferably not
through a via. The capacitor must be located right at
the IC. The Vdd terminal is very noise sensitive and
placement of the capacitor is very critical.
Connections must be made with wide trace.
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close to
the inductor.
• Keep the switch node (SW) away from the feedback
(FB) pin.
• To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
Micrel, Inc. MIC2169B
April 2010 18 M9999-041210-B
MOSFETs
• Low gate charge MOSFETs should be used to
maximize efficiency, such as Si4800, Si4804BDY,
IRF7821, IRF8910, FDS6680A and FDS6912A, etc.
RC Snubber
• Add a RC snubber of 1.4 resistor and a 1000pF
capacitor from the switch node to ground pin. Place
the snubber on the same side of the board and as
close to the MOSFETs as possible. See page 8,
Current Limiting section for more detail.
Schottky Diode (Optional)
• Place the Schottky diode on the same side of the
board as the MOSFETs and VIN input capacitor.
• The connection from the Schottky diode’s Anode to
the input capacitors ground terminal must be as
short as possible.
• The diode’s Cathode connection to the switch node
(SW) must be keep as short as possible.
Others
• Connect the current limiting (R2) resistor directly to
the drain of top MOSFET Q3.
• The feedback resistors R3 and R4/R5/R6 should be
placed close to the FB pin. The top side of R3
should connect directly to the output node. Run this
trace away from the switch node (junction of Q3, Q2,
and L1). The bottom side of R3 should connect to
the GND pin on the MIC2169B.
• The compensation resistor and capacitors should be
placed right next to the COMP pin and the other side
should connect directly to the GND pin on the
MIC2169B rather than going to the plane.
• Add a place holder for a gate resistor on the top
MOSFET gate drive. Do not use a resistor in series
with the low-side MOSFET gate.
Evaluation Board Schematics
MIC2169B Evaluation Board Schematic
Micrel, Inc. MIC2169B
April 2010 19 M9999-041210-B
Bill of Materials
Item Part Number Manufacturer Description Qty.
U1 MIC2169B-YMME Micrel, Inc. Buck Controller 1
Q1, Q2 IRF7821-TR
SI4174DY
IR
Vishay
30V, N-Channel HEXFET, Power MOSFET 2
0
Q3 2N7002E On Semiconductor 60V, N-Channel MOSFET 0
D1 SD103BWS Vishay 30V, Schottky Diode 1
D2 1N5819HW
SL04
CMMSH1-40
Diodes, Inc.
Vishay
Central Semi
40V, Schottky Diode 1
0
0
L1 CDRH127LDNP-1R0NC
HC5-1R0
SER1360-1R0
Sumida
Cooper Electronic
Coilcraft
1.0H, 10A Inductor 1
0
0
C1 C3225X7R1C106M TDK 10F/16V, X7R Ceramic Capacitor 1
C2, C3 TPSD686M020R0070
594D686X0020D2T
AVX
Vishay/Sprague
68F, 20V Tantalum 2
0
C4 C2012X5R0J106M TDK 10F/6.3V, 0805 Ceramic Capacitor 1
C5, C10, C12,
C16
VJ1206Y104KXXAT Vishay Victramon 0.1F/25V Ceramic Capacitor 4
C6, C7 TPSD337M006R0045 AVX 330F/6.3V, Tantalum 2
C8, C11, C17 Open 0
C13 C2012X7R1C105K
GRM21BR71C105KA01B
VJ1206S105KXJAT
TDK
muRata
Vishay Victramon
1F/16V, 0805 Ceramic Capacitor 1
0
0
C15 VJ0603A102KXXAT Vishay Victramon 1000pF/25V, 0603, NPO 1
R2 CRCW06034700JRT1 Vishay 470, 0603, 1/16W, 5% 1
R3 CRCW08051002FRT1 Vishay 10k, 0805, 1/10W, 1% 1
R4 CRCW08053161FRT1 Vishay 3.16k, 0805, 1/10W, 1% 1
R5 CRCW08054641FRT1 Vishay 4.64k, 0805, 1/10W, 1% 1
R6 CRCW08051132FRT1 Vishay 11.3k, 0805, 1/10W, 1% 1
R7 CRCW08051003FRT1 Vishay 100k, 0805, 1/10W, 1% 1
C9 VJ0603A151KXAAT Vishay 150pF/50V, 0603, NPO 1
Notes:
1. Micrel.Inc 408-944-0800
2. Vishay corp 206-452-5664
3. Diodes. Inc 805-446-4800
4. Sumida 408-321-9660
5. TDK 847-803-6100
6. muRata 800-831-9172
7. AVX 843-448-9411
8. International Rectifier 847-803-6100
9. Fairchild Semiconductor 207-775-8100
10. Cooper Electronic 561-752-5000
11. Coilcraft 1-800-322-2645
12. Central Semi 631-435-1110
Micrel, Inc. MIC2169B
April 2010 20 M9999-041210-B
Bill of Materials (continued)
Item Part Number Manufacturer Description Qty.
R8 CRCW06034021FRT1 Vishay 4.02k, 0603, 1/16W, 1% 1
R9 CRCW120610R0FRT1 Vishay 10, 1/8W, 1206, 1% 1
R10 CRCW12062R00FRT1 Vishay 2, 1/8W, 1206, 1% 1
R12 CRCW12061R40FRT1 Vishay 1.4, 1/8W, 1206, 1%
R14 Open 0
J1, J3, J4, J5 2551-2-00-01-00-00-07-0 MillMax Turrent Pins 4
Notes:
13. Micrel.Inc 408-944-0800
14. Vishay corp 206-452-5664
15. Diodes. Inc 805-446-4800
16. Sumida 408-321-9660
17. TDK 847-803-6100
18. muRata 800-831-9172
19. AVX 843-448-9411
20. International Rectifier 847-803-6100
21. Fairchild Semiconductor 207-775-8100
22. Cooper Electronic 561-752-5000
23. Coilcraft 1-800-322-2645
24. Central Semi 631-435-1110
Micrel, Inc. MIC2169B
April 2010 21 M9999-041210-B
MIC2169B PCB Layout
MIC2169B Top Layer MIC2169B Bottom Layer
Micrel, Inc. MIC2169B
April 2010 22 M9999-041210-B
Package Information
10-Pin ePad MSOP (MME)
Micrel, Inc. MIC2169B
April 2010 23 M9999-041210-B
Package Information (continued)
10-Pin MSOP (MM)
Micrel, Inc. MIC2169B
April 2010 24 M9999-041210-B
MIC2169B Land Patterns
Recommended Land Pattern for 10-Pin MSOP
Micrel, Inc. MIC2169B
April 2010 25 M9999-041210-B
MIC2169B Land Patterns (continued)
Recommended Land Pattern for ePad 10-Pin MSOP
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2006 Micrel, Incorporated.