FEBRUARY 1997 VOLUME VII NUMBER 1
, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,
LinearView, Micropower SwitcherCAD, PowerPath, SwitcherCAD and UltraFast are trademarks of Linear Technology
Corporation. Other product names may be trademarks of the companies that manufacture the products.
The New LT1425
Isolated Flyback Controller
by Kurk Matthews
Introduction
Low voltage circuitry, such as local
area networks (LAN), isolation ampli-
fiers and telephone interfaces,
frequently requires isolated power
supplies. The flyback converter is of-
ten the choice for these low power
supplies because of its simplicity, size
and low parts count. Unfortunately,
designers are forced to add opto-
couplers and references in order to
achieve the desired output regulation
and transient response.
The new LT1425 provides a one-
chip solution for these and other
applications. The LT1425 is a 275kHz
current mode controller with an inte-
gral 1.25A switch designed primarily
to provide well regulated, isolated
voltages from 3V–20V sources. The
LT1425 is available in a 16-pin SO.
Features include a new error amplifier
and load compensation circuitry that
eliminate the need for optocouplers
while maintaining output regulation
typically within a few percent.
Figure 1 shows a typical flyback
LAN supply using the LT1425. Figure
1 also includes details on an alter-
nate transformer for a complete
PCMCIA type II height solution. The
output voltage is within 1% of –9V for
load currents of 0mA–250mA. Input
current is limited to 0.35 amps in the
event the output is short circuited.
The output voltage droops only 300mV
during a 50mA to 250mA load tran-
sient (see Figure 2). The off-the-shelf
transformers provide 500V
AC
of isola-
tion. The high switching frequency
allows the use of small case size, low
cost, high value ceramic capacitors
on the input and output of the supply.
Isolated Feedback
The heart of the LT1425 is shown in
Figure 3. During S1’s off-time, the
voltage on the V
SW
pin increases to V
IN
+ (V
OUT
+ V
D
)/n, where n is the trans-
former turns ratio and V
D
is the output
diode voltage. Q1’s collector current
becomes I
CQ1
= (V
OUT
+ V
D
)/(n × R1).
R2 converts I
CQ1
into the input volt-
age for the transconductance feedback
amplifier. C1 on the V
C
pin then inte-
grates the feedback amplifier’s output
current. The voltage on the V
C
pin
sets the current mode trip point.
Although we now have a means for
generating a feedback voltage, a few
problems remain. The feedback volt-
age is not present during S1’s on-time
or when the secondary current de-
cays to zero, which is often the case
with a discontinuous flyback. To make
matters worse, T1’s leakage induc-
tance can cause large voltage spikes
at turn-off.
These issues are taken care of by
the error amplifier enable block, which
incorporates enable-delay, collapse-
detect and minimum-enable-time
circuitry. Enable delay waits approxi-
mately 200ns after the switch turns
off before enabling the feedback
amplifier, thus avoiding the leakage-
inductance spike. The collapse detect
continued on page 3
IN THIS ISSUE . . .
COVER ARTICLE
The New LT®1425
Isolated Flyback Controller ........ 1
Kurk Matthews
Issue Highlights ........................2
LTC in the News .........................2
DESIGN FEATURES
The LT1328: a Low Cost IrDA®
Receiver Solution for Data Rates
up to 4Mbps ...............................6
Alexander Strong
The LTC®1473 Dual PowerPath™
Switch Driver Simplifies Portable
Power Management Design ........8
Jaime Tseng
The LTC1560-1: a 1MHz/500kHz
Continuous-Time, Low Noise,
Elliptic Lowpass Filter ............ 11
Nello Sevastopoulos
The LTC1594 and LTC1598:
Micropower 4- and 8-Channel
12-Bit ADCs ............................. 14
Kevin R. Hoskins and Marco Pan
LTC1474 and LTC1475 High
Efficiency Switching Regulators
Draw Only 10µA Supply Current
................................................17
Greg Dittmer
DESIGN IDEAS .................. 21–32
(Complete list on page 21)
DESIGN INFORMATION
Introducing the LT2078/LT2079
and LT2178/LT2179 Single
Supply, Micropower, Precision
Amplifiers in Surface Mount
Packages
................................................33
Raj Ramchandani
LTC1387 Single 5V
RS232/RS485 Multiprotocol
Transceiver .............................34
Y.K. Sim
New Device Cameos.................. 37
Design Tools ............................39
Sales Offices ............................40
LINEAR TECHNOLOGY
LINEAR TECHNOLOGY
LINEAR TECHNOLOGY
Linear Technology Magazine • February 1997
2
EDITOR’S PAGE
Issue Highlights
To mark the new year, we have a
collection of exciting new parts from
the design gurus at LTC. This issue’s
lead article features the new LT1425
isolated flyback converter, which pro-
vides a one-chip solution for low
voltage circuitry, such as local area
networks, isolation amplifiers and
telephone interfaces. The LT1425 is a
275kHz current mode controller with
an integral 1.25A switch, designed to
provide well regulated, isolated volt-
ages from 3V–20V sources.
Other power products featured in
this issue include the LTC1474 and
LTC1475 ultralow quiescent current,
high efficiency step-down switching
regulators. These regulators draw only
10µA at no load and require only four
external components to make a com-
plete, high efficiency (up to 92%)
step-down regulator. Low component
count and the parts’ tiny MSOP pack-
ages provide a minimum-area solution
to meet the limited space require-
ments of portable applications.
Another boon to the designers of
portable, battery-powered equipment
is the LTC1473 dual PowerPath switch
driver, which simplifies the design of
circuitry for switching between two
batteries or a battery and an AC
adapter. Presently, switching between
power sources is implemented with
discrete components—regulators,
comparators, references, glue logic,
MOSFET switches and drivers. These
solutions are expensive and occupy a
lot of printed circuit board space. The
LTC1473 drives low loss N-channel
MOSFET switches that direct power
in the main power path of a single or
dual rechargeable battery system, the
type found in most notebook comput-
ers and other portable equipment.
In the filter department, this issue
introduces the LTC1560-1, a high
frequency, continuous-time, low noise
filter. This device is a single-ended
input and output, 5th order elliptic
lowpass filter with a pin-selectable
cutoff frequency of 1MHz or 500kHz.
It requires no external components or
clocks and provides better than 60dB
LTC in the News…
Results of LTC’s second fiscal quarter
underscore the company’s standing as
an industry leader. Net sales for the
second fiscal quarter of 1997, ended
December 29, 1996, were $90,080,000.
Although this was a decrease of 6% over
net sales of $96,017,000 for the second
quarter of the previous year, this actu-
ally represented phenomenally good
performance, as we will see in just a
moment.
LTC reported net income for the quar-
ter of $31,631,000 or $0.40 per share,
a decrease of 8% from the second quar-
ter of the previous year. Sequentially,
the results for the second quarter were
essentially flat as compared to net sales
and income for the quarter ended Sep-
tember 29, 1996 of $90,063,000 and
$31,358,000 or $0.40 per share, re-
spectively. A cash dividend of $0.05 will
be paid on February 12, 1997 to share-
holders of record on January 24, 1997.
According to Robert H. Swanson,
president and CEO, “Although we en-
tered the quarter with reduced backlog,
we were able to achieve flat sequential
sales and profits with our return on
sales continuing to lead the industry. In
addition, cash and short-term invest-
ments grew by approximately $20 mil-
lion in the quarter. Customers’ demand
picked up moderately throughout the
quarter and our shorter lead times en-
abled us to ship some of their demand
within the quarter. We believe our mar-
ket is improving and we are optimistic
about the future.”
It’s becoming known around the
nation’s regional stock exchanges that
LTC is a leading pace setter and vital
indicator of the economic condition of
the electronics industry. For many
months, the Bloomberg Silicon Valley
Index and the Dean Witter Silicon Val-
ley Stock Index have included LTC’s
business performance in their assess-
ment of the industry.
Now, the New York-based Reuters
America News Service reports that the
Philadelphia Stock Exchange has added
two chip companies—one of which is
LTC—and an equipment maker to its
Semiconductor Index to replace three
stocks the exchange has removed. Late
last month, the exchange added Linear
Technology Corp. (LLTC), Xilinx Inc.
(XLNX.O) and Lam Research Corp.
(LRCX.O). To us at LTC, this move is
further proof that even in the realm of
high finance, “it’s a linear world.”
of stopband attenuation and 75dB
SNR, with only 0.3dB passband ripple.
In the data conversion area, we
debut the LTC1594 and LTC1598,
micropower 12-bit ADCs, which fea-
ture a 4- or 8-channel multiplexer,
respectively. These devices include
an auto shutdown feature that re-
duces power dissipation when the
converter is inactive. Nominal power
dissipation with the converter clocked
at 320kHz is typically 1.6mW. Each
ADC includes a simple, efficient serial
interface that reduces interconnects
and, thereby, possible noise sources.
Reduced interconnections also reduce
board size and allow the use of pro-
cessors having fewer I/O pins, both of
which help reduce system costs.
For data communications, this is-
sue introduces the low cost LT1328
IrDA data receiver. This device con-
tains all the necessary circuitry to
convert current pulses from an exter-
nal photodiode to a digital TTL output
while rejecting unwanted lower fre-
quency interference. The LT1328 plus
six external components are all that
is required to make an IrDA-compat-
ible receiver. Power requirements for
the LT1328 are minimal: a single 5V
supply and 2mA of quiescent current.
This issue includes a varied selec-
tion of Design Ideas, including three
power supplies, a battery charger that
doubles as the main step-down con-
verter, a voltage controlled limiter for
video, a detector circuit for 470MHz
signals, and an evaluation of battery
life under a variety of load conditions.
Penultimately, we present Design
Information on the LT2078/LT2079
and LT2178/LT2179, improved
single-supply, precision surface
mount op amps, and the LT1387
single 5V multiprotocol transceiver.
We conclude with a selection of New
Device Cameos.
Linear Technology Magazine • February 1997
3
DESIGN FEATURES
+
+
1425_03.eps
V
IN
Q1 Q2 2.6V
R
FB
R1
V
SW
V
OUT
C1
R3
V
C
R
OCMP
R
CMPC
50k
2 A/V
S1
T1
1:N
COMP LOGIC/
DRIVER
R
REF
1.224V ERROR
AMP
ENABLE
R2
disables the feedback amplifier when
the R
REF
voltage falls below 80% of the
1.224V reference. This natural col-
lapse of the feedback voltage occurs
sometime during the off-time in the
discontinuous flyback mode (see Fig-
ure 4, Trace C) or when the switch
turns on in the continuous mode (see
Trace A). Finally, a 200ns minimum
enable time, which follows the enable
delay time, ensures that the error
amplifier can pump up the V
C
node
during start up and other conditions
when V
OUT
is low.
This unique feedback system pro-
duces controlled output voltages while
maintaining fast dynamic response
not found in similar isolated flyback
schemes. 200ns of leading edge, cur-
rent sense blanking is also included
to reject turn-on spikes.
Load Compensation
If the world were a perfect place, with
ideal transformers, diodes and ca-
pacitors, no additional compensation
would be required to maintain perfect
regulation. Unfortunately, as the load
current increases, the additional volt-
age drop due to secondary winding
resistance, the output diode and ca-
pacitor ESR results in decreased
output voltage. To compensate for
this change in output voltage, a cur-
rent is generated in Q2 (see Figure 3),
which is proportional to the average
primary current. Since primary cur-
rent changes with output load, the
effects of nonideal components are
minimized and regulation is possible
over a wide load range. R3 determines
the amount of load compensation.
Connecting R
CMPC
to ground defeats
the load compensation.
Figure 3. LT1425 isolated feedback block diagram
LT1425, continued from page 1
1425_01.eps
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
GND
N/C
R
FB
V
C
R
REF
SYNC
SGND
GND
3.01k
1%
R3
R1 R2
2
4
1
3
7
MBRS130LT3
T1
6
22.1k
1%
0.1µF
LT1425
47pF
C5
C6
C3
10µF
25V
C4
10µF
25V
1.8k
OUT
COM
–9V
1000pF
C1
10µF
25V
5V
INPUT
COM
C2
10µF
25V
0.1µF
100k
GND
SD
R
OCMP
R
CMPC
V
IN
V
SW
PGND
GND
C1, C2, C3, C4 = MARCON THCS50E1E106Z CERAMIC
 CAPACITOR, SIZE 1812. (847) 696-2000

D1
D2
1TREMROFSNART
L
IRP
SNRUT OITARNOITALOSI EZIS
L(×W×)HI
T
U
O
YCNEICIFFE1D2D2R,1R6C,5C3R
ELAD
703A-1484-EPL63µH1:1:1CAV0057.01×5.11×mm3.6Am052%67DESUTONDESUTON74Fp033k3.31
SCINORTLIOC
38431-20XTC72µH1:1CAV00541×41×mm2.2Am002%078425N11LT0450RBM57Fp022k9.5
Figure 1. 5V to –9V/250mA isolated LAN supply
Figure 2. Transient response of LT1425 5V to
–9V converter
200mV/DIV
100mA/DIV
5ms/DIV
Linear Technology Magazine • February 1997
4
DESIGN FEATURES
of 10V–15V). The isolation voltage is
ultimately limited only by bobbin se-
lection and transformer construction.
The schematic shows details on build-
ing the transformer.
Figure 7 implements a 12V to
5V/1A step-down regulator with off-
the-shelf magnetics. The circuit uses
an external, cascoded 100V MOSFET
to extend the LT1425’s 35V maxi-
mum switch voltage limit. D1 and Q1
ensure the LT1425 does not start
until almost 9V, guaranteeing ad-
equate gate voltage for the MOSFET.
The MUR120 prevents the source from
rising above the gate at turn-off.
The circuit in Figure 8 achieves
even higher input voltages, this time
in the form of a –48V to 5V/2A iso-
lated telecom supply. The input
voltage is too high to directly run Q1
or the LT1425, so a bootstrap wind-
ing is used to provide feedback and
power for the IC after start-up. The
voltage to the V
IN
pin is controlled by
D1, D2, Q2, Q3 and associated com-
ponents, which form the necessary
start-up circuitry with hysteresis.
Nothing happens until C1 charges
through R1 to 15V. At that point, Q2
turns on Q3, pulling the shutdown
pin high. Q3, in turn, latches Q2 on,
setting the turn-off voltage to ap-
proximately 11V. Switching begins
1
2
3
4
8
7
6
5
TOP VIEW
LT1424
SD
V
C
SYNC
SGND
R
CMPC
V
IN
V
SW
PGND
1425_05.eps
+
+
+
1425_06.eps
GND
N/C
LT1425
MBRS1100T3
MBRS1100T3
45
6
7
T1*
3
2
18
R
FB
V
C
R
REF
SYNC
SGND
GND
GND
SD
R
OCMP
R
CMPC
PIN 3 TO 4, 7 TURNS BIFILAR 34AWG
*PHILIPS EFD-15-3F3 CORE
GAP FOR PRIMARY
L = 40µH
0.12 INCH MARGIN TAPE
PIN 7 TO 8, 28 TURNS 40AWG
PIN 5 TO 6, 28 TURNS 40AWG
PIN 1 TO 2, 7 TURNS BIFILAR 34AWG
3 LAYERS 2 MIL
POLYESTER FILM
V
IN
V
SW
PGND
GND
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
3.01k
1%
1N759
18.4k
0.1%
3k
+15V
–15V
OUT
COM
7.32k
1%
75
5V TO
15V
INPUT
COM
0.1µF
220pF
1µF
22µF
35V
15µF
35V
3k
15µF
35V
1000pF
0.1µF
130 330pF
9
MBR0540LT1
Figure 5. LT1424 pinout
Figure 6. Fully isolated ±15V, ±60mA supply
D: ISW = 0.2A/DIV
C: VSW = 20V/DIV
B: ISW = 1A/DIV
A: VSW = 20V/DIV
1µs/DIV
Figure 4. Switch voltage and current for
Figure 1’s circuit with outputs of –9V/250mA
and –9V/30mA
The LT1424
The LT1424, devoted to fixed output
voltage applications, is available in
an 8-pin SO package. The LT1424
retains the features of the LT1425
and incorporates the feedback, refer-
ence and load compensation resistors.
Figure 5 shows the LT1424 pinout.
Both the LT1424 and LT1425 include
shutdown and synchronization func-
tions. Consult the factory for further
information on the LT1424.
Typical Applications
Figure 6 shows a ±15V supply with
1.5kV of isolation. Output regulation
remains within ±3% over the entire
5V to 15V input voltage and ±60mA
output current range, even with one
output fully loaded and the other
unloaded (±1.5% with input voltages
and, before C1 has a chance to dis-
charge to 11V, the bootstrap winding
begins to supply power. If the output
is shorted, R2 prevents C1 from being
charged by the transformer’s leakage
energy, causing the supply to con-
tinually attempt to restart. This limits
input and output current during a
short circuit. Feedback voltage is fed
directly through a resistor divider to
the R
REF
pin. The sampling error am-
plifier still works, but the load
compensation circuitry is bypassed.
This results in a ±5% load regulation
over line and load. A dedicated feed-
back winding referencing the feedback
voltage to the V
IN
pin could be used to
include the load compensation func-
tion and improve regulation.
Conclusion
The LT1425 offers high performance
and accuracy without the additional
circuitry traditionally associated with
isolated DC to DC converters.
Linear Technology Magazine • February 1997
5
DESIGN FEATURES
++
+
220µF
10V
1425_07.eps
GND
N/C
LT1425
MBRS340T3
2
5
1
4
6
3
10
7
11
8
12
9
R
FB
V
C
R
REF
SYNC
SGND
GND
GND
SD
R
OCMP
R
CMPC
V
IN
V
SW
PGND
GND
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
3.01k
1%
25.5k
1%
9.3k
1%
MMFT1N10E
2.4k
12V
INPUT
COM
0.1µF
22µF
35V 220µF
10V 200
5V
OUT
COM
COILTRONIX
VP1-0190
TURNS RATIO 1 : 1 : 1 : 1 : 1 : 1
12µH PER WINDING
407-241-7876
1000pF
1000pF
MUR120
Q1
2N3906
0.1µF
100
10
1.8k
330pF
9
D1
1N755
7.5V
+
++
1425_08.eps
GND
N/C
BAV21
BAV21
MUR120
LT1425
5k
18
MBR745
10
47
8
T1*
3
2
1
R
FB
V
C
R
REF
SYNC
SGND
GND
GND
SD
R
OCMP
R
CMPC
PIN 3 TO 4, 15 TURNS BIFILAR 31AWG
*PHILIPS EFD-15-3F3 CORE
GAP FOR PRIMARY
L = 100µH
PIN 7 TO 8, 6 TURNS QUADFILAR 29AWG
PIN 5 TO 6, 15 TURNS BIFILAR 33AWG
PIN 1 TO 2, 15 TURNS BIFILAR 31AWG
1 LAYER 2 MIL 
POLYESTER FILM
2 LAYERS 2 MIL 
POLYESTER FILM
V
IN
V
SW
PGND
GND
1
2
3
4
5
6
7
8
16
T1
6
5
15
14
13
12
11
10
3.16k
1%
Q2
2N3906 Q3
2N3904
Q1
IRF610
D1
7.5V
1N755
D2
7.5V
1N755
30.1k
1%
R2
18
R1
24k 50
1W
510
10k
2.4k
100k
INPUT
COM
–36V TO
–72V
3.3µF
150pF
0.1µF
0.1µFC1
27µF
35V
150µF
6.3V
150µF
6.3V
5V
OUT
COM
1000pF
470pF
9
Figure 7. 5V/1A step-down, isolated supply
Figure 8. 5V/2A telecommunications supply
Linear Technology Magazine • February 1997
6
DESIGN FEATURES
V
CC
(5V)
1328_01.eps
C4
0.1µFC5
10µF
HIGH – SIR
LOW – FIR AND 4PPM
IN
FILT
FILT LO
GND
LT1328
MODE
C3
1000pF
V
CC
DATA
V
BIAS
8
7
6
TTL DATA OUT
C1
330pF
LIGHT IN
D1
BPU22NF
TEMIC
C2
10nF
1328_02.eps
R
D3
10k
R
D1
100
DRIVER
Q3
2N7002
TRANSMIT
INPUT
R
D2
6.8
1/2W
Q4
2N7002
D2
HSDL-4220
V
CC
The LT1328: a Low Cost IrDA Receiver
Solution for Data Rates up to 4Mbps
by Alexander Strong
Introduction
The need for ever increasing data
rates required by a vast array of de-
vices, such as notebook computers,
printers, mobile phones, pagers, elec-
tronic cameras and modems, has been
satisfied by the technology of infrared
data transmission. The Infrared Data
Association (IrDA
®
) standard, which
covers data rates from 2400bps to
4Mbps, is the overwhelming choice
for infrared data transmission. The
LT1328 is a photodiode receiver that
supports IrDA data rates up to 4Mbs,
as well as other modulation methods,
such as Sharp ASK and TV remote
control.
The LT1328, in the MSOP package,
contains all the necessary circuitry to
convert current pulses from an exter-
nal photodiode to a digital TTL output
while rejecting unwanted lower fre-
quency interference. The LT1328 plus
six external components is all that is
required to make the IrDA-compat-
ible receiver shown in Figure 1. An
IrDA- compatible transmitter can also
be implemented with only six compo-
nents, as shown in Figure 2. Power
requirements for the LT1328 are mini-
mal: a single 5V supply and 2mA of
quiescent current.
LT1328
Functional Description
Figure 3 is a block diagram of the
LT1328. Photodiode current from D1
is transformed into a voltage by feed-
back resistor R
FB
. The DC level of the
preamp is held at V
BIAS
by the servo
action of the transconductance
amplifier’s g
m
. The servo action only
suppresses frequencies below the
R
g
m
/C
FILT
pole. This highpass filter-
ing attenuates interfering signals,
such as sunlight or incandescent or
fluorescent lamps, and is selectable
at pin 7 for low or high data rates. For
high data rates, pin 7 should be held
low. The highpass filter breakpoint is
set by the capacitor C1 at f = 25/(2
× R
g
m
× C), where R
g
m
= 60k. The
330pF capacitor (C1) sets a 200kHz
corner frequency and is used for data
rates above 115kbps. For low data
rates (115kbps and below), the ca-
pacitance at pin 2 is increased by
taking pin 7 to a TTL high. This
switches C2 in parallel with C1,
lowering the highpass filter break-
point. A 10nF cap (C2) produces a
6.6kHz corner. Signals processed by
the preamp/g
m
amplifier combina-
tion cause the comparator output to
swing low.
IrDA SIR
The LT1328 circuit in Figure 1 oper-
ates over the full 1cm to 1 meter range
of the IrDA standard at the stipulated
light levels. For IrDA data rates of
115kbs and below, a 1.6µs pulse width
is used for a zero and no pulse for a
one. Light levels are 40mW/sr (Watts
per steradian) to 500mW/sr. Figure 4
shows a scope photo for a transmitter
input (top trace) and the LT1328 out-
put (bottom trace). Note that the input
to the transmitter is inverted; that is,
transmitted light produces a high at
the input, which results in a zero at
the output of the transmitter. The
Mode pin (pin 7) should be high for
these data rates.
IrDA FIR
The second fastest tier of the IrDA
standard addresses 576kbps and
1.152Mbs data rates, with pulse
widths of 1/4 of the bit interval for
zero and no pulse for one. The
1.152Mbs rate, for example, uses a
pulse width of 217ns; the total bit
time is 870ns. Light levels are
100mW/sr to 500mW/sr over the 1cm
to 1 meter range. A photo of a trans-
mitted input and LT1328 output is
Figure 1. LT1328 IrDA receiver—typical application Figure 2. IrDA transmitter
Linear Technology Magazine • February 1997
7
DESIGN FEATURES
+
+
1
1328_03.eps
COMPARATOR
g
m
CELL
BIAS
V
BIAS
FILTER
PHOTODIODE
IN
FILTER LO
DATA OUT
V
CC
MODE
GND
R
GM
R
FB
R
IN
PREAMP
2
3
4
C1
330pF C2
10nF
8
C3
7
6
5
D1
shown in Figure 5. The LT1328 output
pulse width will be less than 800ns
wide over all of the above conditions
at 1.152Mbps. Pin 7 should be held
low for these data rates and above.
4ppm
The last IrDA encoding method is for
4Mbs and uses pulse position modu-
lation, thus its name: 4ppm. Two bits
are encoded by the location of a 125ns
wide pulse at one of the four positions
within a 500ns interval (2 bits ×
1/500ns = 4Mbps). Range and input
levels are the same as for 1.152Mbs.
Figure 6 shows the LT1328 reproduc-
tion of this modulation.
Conclusion
In summary, the LT1328 can be used
to build a low cost receiver compat-
ible with IrDA standards. Its ease of
use and flexibility also allow it to
provide solutions to numerous other
photodiode receiver applications. The
tiny MSOP package saves on PC board
area.
Figure 3. LT1328 block diagram IrDA is a registered trademark of the Infrared Data
Association
Authors can be contacted
at (408) 432-1900
Figure 4. IrDA 115kbs modulation Figure 5. IrDA 1.152Mbs modulation Figure 6. IrDA 4ppm modulation
TRANSMITTER
INPUT
LT1328 OUTPUT
2ns/DIV
TRANSMITTER
INPUT
LT1328 OUTPUT
200ns/DIV
TRANSMITTER
INPUT
LT1328 OUTPUT
200ns/DIV
Linear Technology Magazine • February 1997
8
DESIGN FEATURES
The LTC1473 Dual PowerPath Switch
Driver Simplifies Portable Power
Management Design by Jaime Tseng
Introduction
The LTC1473 is the latest addition to
Linear Technology’s new family of
power management controllers, which
simplify the design of circuitry for
switching between two batteries or a
battery and an AC adapter. Presently,
switching between power sources is
implemented with discrete compo-
nents—a mixture of regulators,
comparators, references, glue logic,
MOSFET switches and drivers. In-
variably, these solutions are expensive
and occupy a considerable amount of
printed circuit board space. Although
these circuits are frequently designed
in a hurry, the problems associated
with power path switching are often
subtle and daunting. For example,
switching from one battery to another
can produce huge inrush currents
between the batteries when their volt-
ages differ. In extreme cases, system
bypass capacitors can be destroyed if
tantalums are used. Slowing the
switch turn-on rate helps reduce the
inrush current, but may cause a pre-
cipitous drop in the system supply
voltage.
Solutions to these “real world” prob-
lems have been designed into our new
power path
switch driver. The
LTC1473 dual PowerPath™ switch
driver drives low loss N-channel MOS-
FET switches that direct power in the
main power path of a single or dual
rechargeable battery system, the type
found in most notebook computers
and other portable equipment.
Overview
The power management system in
Figure 1 shows the LTC1473 driving
two sets of back-to-back N-channel
MOSFET switches connecting the two
batteries to the system DC/DC regu-
lator. Each of the switches is controlled
by a TTL/CMOS compatible input
that interfaces directly with a power
management system microprocessor.
An internal boost regulator provides
the voltage to fully enhance the logic-
level N-channel MOSFET switches.
The LTC1473 uses a current sense
loop to limit current rushing in and
out of the batteries and the system
supply capacitor during switch-over
transitions or during a fault condi-
tion. A user programmable timer
monitors the time during which the
MOSFET switches are in current limit
and latches them off if the pro-
grammed time is exceeded. A unique
“2-diode logic mode” ensures system
start-up, regardless of which input
receives power first.
BAT1
BAT2
SWA1
SWA2
+
HIGH
EFFICIENCY
DC/DC
SWITCHING
REGULATOR
5V
3.3V
12V
C
IN
DCIN
LTC1473
POWER
MANAGEMENT
µP
SWB2
SWB1 R
SENSE
STEP-UP
SWITCHING
REGULATOR
1µF
50V
C1
GATE
DRIVER
GATE
DRIVER
1µF
50V
C2
V
GG
V
+
INRUSH
CURRENT
SENSING
AND LIMITING
C1
4700pF
C
TIMER
TIMER
1473_01.eps
IN1
IN2
DIODE
SW
L1
1mH
+
+
Si9926
Si9926
MB914LT1
MBRD340
Figure 1. Dual-battery PowerPath switch driver: V
GG
regulator, inrush limiting and switch-gate drivers
Linear Technology Magazine • February 1997
9
DESIGN FEATURES
BAT1
BAT2
SWA1
SWA2
+
HIGH
EFFICIENCY
DC/DC
SWITCHING
REGULATOR
5V
3.3V
1473_02.eps
12V
C
IN
DCIN
LTC1473
POWER
MANAGEMENT
µP
SWB2
SWB1
ON OFF
ON OFF
R
SENSE
MBRD340
Si9926
Si9926
Back-to-Back Switches
The back-to-back topology eliminates
the problems associated with the in-
herent body diodes in power MOSFET
switches and allows each switch pair
to block current flow in either direc-
tion when the two switches are turned
off. The low loss, N-channel switch
pairs are housed in 8-pin SO and
SSOP packaging and are available
from a number of manufacturers. The
Si9926DY, for example, houses two
20V MOSFETs rated at 0.03 with
V
GS
= 4.5V.
Inrush Current Limiting
The back-to-back topology also al-
lows for independent control of each
half of the switch pair, facilitating
bidirectional inrush current limiting.
The voltage across a single low value
resistor, R
SENSE
, is measured to de-
termine the instantaneous current
flowing through the two main switch
pairs, SWA1/B1 and SWA2/B2. The
inrush current is then controlled by
the gate drivers until the transition
from one power source to the other
has been completed. The current flow-
ing in and out of the two main power
sources and the DC/DC converter
input capacitor is dramatically
reduced.
Tantalum Capacitors
Tantalum capacitors, with their high
volumetric efficiency and low ESR,
are the dielectric of choice for low
impedance applications, such as fil-
tering the input of a switching
regulator. However, because these ca-
pacitors are exposed to uncontrolled
energy supplies, they are subject to
failures caused by high inrush cur-
rents unless current surges are
restricted. The inrush-current limit-
ing feature of the LTC1473 makes it
feasible to use low profile tantalum
surface mount capacitors in place of
bulkier electrolytic capacitors.
Built-In Step-Up Regulator
The gate drive for the two low loss N-
channel switches is supplied by a
micropower step-up regulator that
continuously generates 8.5V above
V+, up to 37V maximum. The V
GG
supply provides sufficient headroom
to ensure that the logic-level MOS-
FET switches are fully enhanced by
the gate drivers which, supply a regu-
lated 5.7V gate-to-source voltage, V
GS
,
when turned on.
The power for the micropower boost
regulator is taken from external di-
odes connected to each power source.
The highest voltage potential is di-
rected to V
+
, where L1, an inexpensive
1mH surface mount inductor, is con-
nected. An internal diode directs the
current from L1 to the V
GG
output
capacitor, C2.
Programmable Fault Timer
A fault-timer capacitor, C
TIMER
, is used
to program the time during which the
MOSFET switches are allowed to be
in current limit continuously. In the
event of a fault condition, the MOS-
FET current is limited by the inrush
current-limit loop. A MOSFET switch
operating in current limit is in a high
dissipation mode and can fail cata-
strophically if this condition is not
promptly terminated.
The fault-time delay is programmed
with an external capacitor connected
between the TIMER pin and ground.
At the instant the MOSFET switch
enters current limit, a 5µA current
source starts charging C
TIMER
through
the TIMER pin. When the voltage
across C
TIMER
reaches 1.2V, an inter-
nal latch is set and the MOSFET
switch is turned off. To reset the
latch, the gate-drive input of the
MOSFET switch is deselected.
The “2-Diode Mode”
Under normal operating conditions,
both halves of each switch pair are
turned on and off simultaneously.
For example, when the input power
source is switched from BAT1 to BAT2,
both gates of switch pair SWA1/B1
are turned off and both gates of switch
pair SWA2/B2 are turned on. The
back-to-back body diodes in switch
pair SWA1/B1 block current flow into
or out of the BAT1 input connector.
In the “2-diode mode,” only the
first half of each power path switch
pair, for example, SWA1 and SWA2, is
turned on, and the second half, that
is, SWB1 and SWB2, is turned off.
These two switch pairs now simply
act as two diodes connected to the
two main input power sources, as
illustrated in Figure 2. The power
path diode with the highest input
voltage passes current to the input of
the DC/DC converter to ensure that
the power management microproces-
sor is powered, even under start-up
Figure 2. LTC1473 dual PowerPath switch driver in “2-diode mode”
Linear Technology Magazine • February 1997
10
DESIGN FEATURES
or abnormal operating conditions.
After “good” power is reconnected to
one of the main inputs, the LTC1473
can be instructed to drive the appro-
priate switch pair on fully as the other
switch is turned off, restoring normal
operation.
Typical Application
A typical dual-battery system is shown
in Figure 3. The LTC1473 accepts
commands from a power manage-
ment microprocessor to select the
SAB1
GB1
SENSE+
SENSE-
GA2
SAB2
GB2
IN1
IN2
DIODE
TIMER
V
+
V
GG
SW
GND
LTC1473GA1
R
SENSE
0.04
+
POWER
MANAGEMENT
µP
SUPPLY
MONITOR
BAT1
DCIN
BAT2
C
OUT
INPUT OF SYSTEM
HIGH EFFICIENCY DC/DC
SWITCHING REGULATOR
(LTC1435,ETC)
C
TIMER
4700PF 1µF
1µF
1mH
Si9926
Si9926
1473_03.eps
MBRD340
MMBD2823LT1
MMBD2823LT1
MMBD914LT1
+
+
appropriate battery. The micropro-
cessor monitors the presence of
batteries and the AC adapter through
a supply monitor block, or, in the case
of some battery packs, through a
thermistor sensor. This block com-
prises a resistor divider and a
comparator for each supply. If the AC
adapter is present, the two switches
are turned off by the microprocessor
and the power is delivered to the
input of the system DC/DC switching
regulator via a Schottky diode.
Conclusion
The LTC1473 dual PowerPath switch
driver eases the design of the front
end of the power management sys-
tem. Designed to drive low cost
N-channel MOSFET switches and
packed with numerous protection fea-
tures in a narrow, 16-lead SSOP
package, the LTC1473 solves the
problems of cost, space and reliability
for power management system
designers.
Figure 3. Dual-battery power-management system
Authors can be contacted
at (408) 432-1900
Linear Technology Magazine • February 1997
11
DESIGN FEATURES
The LTC1560-1: a 1MHz/500kHz
Continuous-Time, Low Noise,
Elliptic Lowpass Filter by Nello Sevastopoulos
Introduction
The LTC1560-1 is a high frequency,
continuous-time, low noise filter. It is
a single-ended input, single-ended
output, 5th order elliptic lowpass fil-
ter with a pin-selectable cutoff
frequency (f
C
) of 1MHz or 500kHz.
Several features distinguish the
LTC1560-1 from other commercially
available high frequency, continuous-
time monolithic filters:
Fifth order 1MHz or 500kHz
elliptic response in an SO-8
package
No external components or
clocks required
Better than 60dB stopband
attenuation
75dB signal-to-noise ratio (SNR)
0.3dB passband ripple
The LTC1560-1 delivers accurate
fixed cutoff frequencies of 500kHz
and 1MHz without the need for inter-
nal or external clocks. Through a
simple mask change, other fre-
quencies from 450kHz to 1.3MHz can
be produced upon demand. The
LTC1560-1’s extremely small size
makes it suitable for compact designs
(see Figure1) and for a variety of ap-
plications, including communication
filters, antialiasing filters and smooth-
ing or reconstruction filters.
DC Performance and
Power Shutdown
The LTC1560-1 operates with ±5V
supplies and has a power shutdown
mode. The typical DC output swing of
the filter is from –3V to 3.5V. The
output DC offset of the filter is typi-
cally ±200mV. The operating power
supply range is ±4V to ±6V.
AC Performance
Frequency Response
The LTC1560-1 offers a pin-select-
able cutoff frequency of either 500kHz
(Figure 2; pin 5 tied to V
+
) or 1MHz
(pin 5 tied to V
). The detailed pass-
band frequency response of the 1MHz
filter is shown in Figure 3. In the
1MHz mode, the passband is flat up
to 0.55 × f
C
with a typical ripple of
±0.2dB, increasing to ±0.3dB up to
0.9 × f
C
. The typical gain at f
C
is
–0.6dB. Referring to Figure 4, note
that the transition band has a gain of
–22dB at 1.44 × f
C
rolling off to –47dB
1
2
3
4
8
7
6
5
VOUT
SD
V+
0.5FC/FC
LTC1560-1
GND
VIN
GND
V
1560_01.eps
1
2
3
4
8
7
6
5
VOUT
5V
5V, 1MHz
5V, 500kHz
0.1µF
(OR 5V)
0.01µF
0.01µF0.1µF
LTC1560-1
VIN
–5V
1560_02.eps
FREQUENCY (MHz)
0.6
0
0.1
0.2
0.3
0.4
0.5
0.6
0.5
0.4
0.3
0.2
0.1
GAIN (dB)
1.00.90.80.70.6
1560_03.eps
0.1 0.50.40.30.2
FREQUENCY (MHz)
–90
–40
–50
–60
–70
–80
10
0
–10
–20
–30
GAIN (dB)
10
1560_04.eps
0.1 1
f
C
= 500kHz
f
C
= 1MHz
Figure 1. LTC1560-1 in an SO-8 package Figure 2. 1MHz or 500kHz elliptic lowpass
filter with no external components Figure 4. Gain vs frequency of the 1MHz and
500kHz filters
Figure 3. Expanded passband ripple for the
1MHz filter
The LTC1560-1’s extremely
small size makes it well
suited for compact designs
and a variety of applications,
including communication
filters, antialiasing filters
and smoothing or
reconstruction filters.
The LTC1560-1 provides a power
shutdown option that significantly
reduces current consumption when
the device is not being used. The filter
operation could be controlled by a
TTL input together with an inverter
applied to the SD pin (pin 7). A logic
high input turns the device on for
normal operation, whereas a logic low
puts the filter into its sleep mode, in
which it dissipates only 5mW of power.
(Leaving pin 7 open yields the default
mode of normal operation.)
Linear Technology Magazine • February 1997
12
DESIGN FEATURES
at 2 × f
C
. The stopband attenuation is
63dB starting from 2.43 × f
C
and
remains at least 60dB for input
frequencies up to 10MHz. When pro-
grammed for f
C
= 500kHz, the
frequency response remains the same,
with the exception of the gain at f
C
,
which is typically –1.3dB. Figure 4
compares the gain responses of the
1MHz and 500kHz filters.
Noise and
Distortion Performance
The LTC1560-1 architecture offers
not only low wide band noise but also
low total harmonic distortion (THD).
The combination of low noise and low
distortion means a wide dynamic
range. With a 1V
RMS
input signal, the
signal-to-noise ratio (SNR) is 69dB
and the THD + Noise is –63dB (0.07%).
The maximum SNR of 75dB is
achieved with a 2.1V
RMS
input signal.
This results in –46dB (0.5%) THD.
For the 500kHz device, the noise per-
formance is even better, with 77dB
SNR at a 1V
RMS
input.
+
1
2
3
4
8
7
6
5
5V
3
1k
2
4
V
OUT
7
LT1360
15V
15V
8
5V, 1MHz
5V, 500kHz
0.1µF
(OR 5V)
0.01µF
0.01µF
0.1µF
0.1µF
0.1µF
LTC1560-1
V
IN
–5V
1560_05.eps
+
15V
15V
0.1µF
0.1µF
1
2
3
4
8
7
6
5
5V
3
1k
2
4
VOUT
7
LT1360 8
0.1µF
(OR 5V)
0.01µF
0.01µF
8.1k
0.1µF
LTC1560-1
VIN
300pF
–5V
1560_06.eps
300pF
To achieve the full high frequency
performance from the filter, a small
resistor (about 200 ) should be added
at the output of the device to isolate
any capacitive load greater than 20pF.
Figure 5 shows a typical application
circuit to be used for any AC
performance measurements of the
LTC1560-1. Any high speed, high
slew-rate operational amplifier such
as the LT1360 can serve as the buffer.
To correctly evaluate the high fre-
quency distortion performance of the
LTC1560-1 requires a very low dis-
tortion input signal, either from a
very high quality signal generator or,
if such a source is not available, from
a source that has been filtered to
control its harmonic content.
Applications and
Experimental Results
The LTC1560-1 can be used not only
as a single device (as shown in Fig-
ures 2 and 5) but also as part of a
more complete frequency-shaping
system. Two representative examples
follow.
Highpass-Lowpass Filter
As a typical application in communi-
cation systems, where there is a need
to reject DC and some low frequency
signals, a 2nd order RC highpass
network can be inserted in front of the
LTC1560-1 to obtain a highpass-low-
pass response. Figures 6 and 7 depict
the network and its measured fre-
quency response, respectively. Notice
that the second resistor in the high-
pass filter is the input resistance of
the LTC1560-1, which is about 8.1k.
Delay-Equalized Elliptic Filter
Although elliptic filters offer high Q
and a sharp transition band, they
lack a constant group delay in the
passband, which implies more ring-
ing in the time-domain step response.
In order to minimize the delay ripple
in the passband of the LTC1560-1, an
allpass filter (delay equalizer) is cas-
caded with the LTC1560-1, as shown
in Figure 8. Figures 9 and 10 illus-
trate the eye diagrams before and
after the equalization, respectively.
An eye diagram is a qualitative
representation of the time-domain
response of a digital communication
system. It shows how susceptible the
system is to intersymbol interference
(ISI). Intersymbol interference is
caused by erroneous decisions in the
receiver due to pulse overlapping and
decaying oscillations of a previous
symbol. A pseudorandom 2-level se-
quence has been used as the input of
the LTC1560-1 to generate these eye
diagrams. The larger eye opening in
Figure 5. A typical circuit for evaluating the full performance of the LTC1560-1
Figure 6. A highpass-lowpass filter
Figure 7. Measured frequency response of
Figure 6’s circuit
Linear Technology Magazine • February 1997
13
DESIGN FEATURES
+
+
1
2
3
4
8
7
6
5
5V
2
6.49k
6.65k
49.9
22pF 0.1µF
0.1µF
22pF
9.75k
20k
40.2k
3V
OUT
1/2 LT1364 16
5
4
8
5V
1/2 LT1364
15V
7
0.1µF
(OR 5V)
0.01µF
0.01µF0.1µF
LTC1560-1
V
IN
–5V
1560_08.eps
Figure 10, an indication of the equal-
ization effect, leads to reduced ISI.
Note that in Figure 8, the equalizer
section has a gain of 2 for driving and
back-terminating 50 cable and load.
For a simple unterminated gain-of-1
equalizer, the 40.2k resistor changes
to 20k and the 49.9 r esistor is re-
Figure 8. Augmenting the LTC1560-1 for improved delay flatness
Figure 9. 2-level eye diagram of the LTC1560-1 before equalization Figure 10. 2-level eye diagram of the equalized filter
for
the latest information
on LTC products, 
visit
www.linear-tech.com
Mojitaba Atarodi contributed significant portions
of this article.
moved from the circuit. The 22pF
capacitors are 1% or 2% dipped silver
mica or COG ceramic.
Conclusions
The LTC1560-1 is a 5th order elliptic
lowpass filter that features a 10-bit
gain linearity at signal ranges up to
1MHz. Being small and user friendly,
the LTC1560-1 is suitable for any
compact design. It is a monolithic
replacement for larger, more expen-
sive and less accurate solutions in
communications, data acquisitions,
medical instrumentation and other
applications.
Linear Technology Magazine • February 1997
14
DESIGN FEATURES
The LTC1594 and LTC1598:
Micropower 4- and 8-Channel
12-Bit ADCs by Kevin R. Hoskins and Marco Pan
Introduction
Data acquisition applications that
require low power dissipation fall into
two general areas: products that re-
quire highly efficient power use, such
as battery-powered portable test
equipment and remotely located data
logging equipment, and products that
either operate in high temperature
environments or must not contribute
to increasing ambient temperature.
To help meet these requirements, Lin-
ear Technology has introduced the
LTC1594 and LTC1598.
Micropower ADCs
in Small Packages
The LTC1594 and LTC1598 are
micropower 12-bit ADCs that feature
a 4- and 8-channel multiplexer, re-
spectively. The LTC1594 is available
in a 16-pin SO package and the
LTC1598 is available in a 24-pin SSOP
package. Each ADC includes a simple,
efficient serial interface that reduces
interconnects and, thereby, possible
sources of corrupting digital noise.
Reduced interconnections also reduce
board size and allow the use of pro-
cessors having fewer I/O pins, both of
which help reduce system costs. Small
packages also shorten the distance
between the ADC and its supply and
voltage reference bypass components.
This reduces lead inductance and
allows bypass components to operate
as efficiently as possible.
Conserve Power with
Auto Shutdown Operation
The LTC1594 and LTC1598 include
an auto shutdown feature that re-
duces power dissipation when the
converter is inactive (whenever the
CS signal is a logic high). Nominal
power dissipation while either con-
verter is clocked at 320kHz is typically
1.6mW. The curve in Figure 1 indi-
cates the amount of current drawn by
this MUXed 12-bit ADC family for
sample rates up to 16.8ksps. As an
example, when converting at 4ksps,
the dissipation is just 450µW and
270µW for the 5V and 3V parts,
respectively.
Supply Flexibility:
2.7V or 5V
To increase applications flexibility,
the LTC1594 and LTC1598 are also
available as 3V parts (LTC1594L and
LTC1598L), which are tested for 2.7V
operation. The LTC1594L and
LTC1598L typically draw 160µA at
maximum conversion rate, one-half
of the supply current drawn by the 5V
parts. Nominal power dissipation
while either converter is clocked at
200kHz (10.5ksps) is typically 800µW.
SAMPLE FREQUENCY (kHz)
0.1
1
SUPPLY CURRENT (µA)
10
100
1000
1 10 100
1598_01.eps
T
A
= 25°C
V
CC
= 5V
V
REF
= 5V
f
CLK
= 320kHz
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
CH5
CH6
CH7
GND
CLK
CS MUX
D
IN
COM
GND
CS ADC
D
OUT
NC
CH4
CH3
CH2
CH1
CH0
V
CC
MUXOUT
ADCIN
V
REF
V
CC
CLK
NC
LTC1598
ANALOG INPUTS
0V TO 5V
RANGE
R4, 7.5k R2, 7.5k
1µF
5V
1598_02.eps
DATA IN
CHIP SELECT
CLOCK
DATA OUT
+
1µF
C6
0.015µF
C4
0.03µF
C2
0.1µF
R1, 7.5k R3, 7.5k
C3
0.03µF
C5
0.015µF
+
C1
0.1µF
5V
1/2
LT1368
1/2
LT1368
Figure 1. Supply current vs sample rate
Figure 2. A simple data acquisition system takes advantage of the LTC1598’s MUXOUT/ADCIN pins to filter analog signals prior to ADC conversion.
Linear Technology Magazine • February 1997
15
DESIGN FEATURES
Good DC Performance
The DC specs include excellent dif-
ferential nonlinearity (DNL) of
±3/4LSB, as required by pen-screen
and other monitoring applications.
No missing codes are guaranteed over
temperature.
Versatile, Flexible Serial I/O
The serial interface found on the
LTC1594 and LTC1598 is designed
for ease of use, flexibility, minimal
interconnections and I/O compatibil-
ity with QSPI, SPI, MICROWIRE™
and other serial interfaces. The MUX
and the ADC have separate chip se-
lect (CS) and serial clock inputs, which
adds versatility. The remaining serial
interface signals are data input (D
IN
)
and data output (D
OUT
). The maxi-
mum serial clock frequencies are
320kHz and 200kHz for the 5V and
3V parts, respectively.
Latch-up Proof MUX Inputs
The LTC1594’s and LTC1598’s input
MUXes are designed to handle input
voltages that exceed the nominal in-
put range, GND to the supply voltage,
without latch-up. Although an over-
driven, unselected channel may
corrupt a selected, correctly driven
channel, no latch-up occurs and cor-
rect conversion results resume when
the offending input voltage is removed.
The MUX inputs remain latch-up proof
for input currents up to ±200mA over
temperature.
Individual ADC
and MUX Chip Selects
Enhance Flexibility
The LTC1594 and LTC1598 feature
separate chip selects for ADC and
MUX. This allows the user to select a
particular channel once for multiple
conversions. This has the following
benefits: first, it eliminates the over-
head of sending D
IN
word for the same
channel each time for each conver-
sion; second, it avoids possible
glitches that may occur if a slow-
settling antialiasing filter is used; and
third, it sets the gain once for mul-
tiple conversions if the MUXOUT/
ADCIN loop is used to create a pro-
grammable gain amplifier (PGA).
MUXOUT/ADCIN
Loop Economizes
Signal Conditioning
The MUXOUT and ADCIN pins form a
very flexible external loop that allows
PGA and/or processing analog input
signals prior to conversion. This loop
is also a cost effective way to perform
the conditioning, because only one
circuit is needed instead of one for
each channel. Figure 2 shows the
loop being used to antialias filter sev-
eral analog inputs. The output signal
of the selected MUX channel, present
on the MUXOUT pin, is applied to R1
of the Sallen-Key filter. The filter band
limits the analog signal and its out-
put is applied to ADCIN. The LT1368
rail-to-rail op amps used in the filter
will, when lightly loaded as in this
application, swing to within 8mV of
the positive supply voltage. Since only
one circuit is used for all channels,
each channel sees the same filter
characteristics.
1598_03.eps
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
20
21
22
23
24
1
2
3
64R
32R
16R
8R
4R
2R
R
R
+
8 COM
18 MUXOUT
GND
4, 9
10
6
5, 14
11
7
CS ADC
CS MUX
CLK
DOUT
DIN
12
13
NC
NC
12-BIT
SAMPLING
ADC
8-CHANNEL
MUX
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
LTC1391
5V
1µF
ADCIN
17 16 15, 19 1µF
0.1µF
5V
1µF
5V
VREF VCC
+
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
V+
D
V
DOUT
DIN
CS
CLK
GND
1/2 LT1368
LTC1598
µP/µC
Figure 3. Using the MUXOUT/ADCIN loop of the LTC1598 to form a PGA with eight gains in a noninverting configuration
MICROWIRE is a trademark of National Semiconductor
Corp.
Linear Technology Magazine • February 1997
16
DESIGN FEATURES
Using MUXOUT/
ADCIN Loop as PGA
Figure 3 shows the LTC1598’s
MUXOUT/ADCIN loop and an LT1368
being used to create a single-channel
PGA with eight noninverting gains.
Combined with the LTC1391, as
shown in Figure 3, the system can
expand to eight channels and eight
gains for each channel. Using the
LTC1594, the PGA is reduced to four
gains. The output of the LT1368 drives
the ADCIN and the resistor ladder.
The resistors above the selected MUX
channel form the feedback for the
LT1368. The loop gain for this ampli-
fier is (R
S
1/R
S
2) + 1. R
S
1 is the
summation of the resistors above the
selected MUX channel and R
S
2 is the
summation of the resistors below the
selected MUX channel. If CH0 is se-
lected, the loop gain is 1 since R
S
1 is
0. Table 1 shows the gain for each
MUX channel. The LT1368 dual rail-
to-rail op amp is designed to operate
with 0.1µF load capacitors. These
capacitors provide frequency compen-
sation for the amplifiers, help reduce
the amplifiers’ output impedance and
improve supply rejection at high fre-
quencies. Because the LT1368’s I
B
is
low, the R
ON
of the selected channel
will not affect the loop gain given by
the formula above. In the case of the
inverting configuration of Figure 4,
the selected channel’s R
ON
will be
added to the resistor that sets the
loop gain.
8-Channel, Differential,
12-Bit A/D System Using
the LTC1391 and LTC1598
The LTC1598 can be combined with
the LTC1391 8-channel, serial-inter-
face analog multiplexer to create a
differential A/D system. Figure 5
shows the complete 8-channel, dif-
ferential A/D circuit. The system uses
the LTC1598’s MUX as the nonin-
verting input multiplexer and the
LTC1391 as inverting input multi-
plexer. The LTC1598’s MUXOUT
drives the ADCIN directly. The
inverting multiplexer’s output is ap-
plied to the LTC1598’s COM input.
The LTC1598 and LTC1391 share the
CS, D
IN
, and CLK control signals.
1598_04.eps
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
20
21
22
23
24
1
2
3
128R
64R
32R
16R
8R
4R
2R
R
+
8 COM GND
4, 9
10
6
5, 14
7
11
CS ADC
CS MUX
CLK
D
IN
D
OUT
12
13
NC
NC
12-BIT
SAMPLING
ADC
ADCINMUXOUT
18 17 16 15, 19 1µF
0.1µF
5V
5V
V
REF
V
CC
+
1/2 LT1368
128R
LTC1598
1598_05.eps
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
20
21
22
23
24
1
2
3
+
8-CHANNEL
MUX
8 COM GND
4, 9
10
6
5, 14
7
11
CS ADC
CS MUX
CLK
D
IN
D
OUT
12
13
NC
NC
12-BIT
SAMPLING
ADC
ADCINMUXOUT
18 17 16 15, 19 1µF
5V
V
REF
V
CC
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
LTC1391
CH0
CH7
D
IN
CLK
CS
D
OUT
5V
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
V+
D
V
D
OUT
D
IN
CS
CLK
GND
LTC1598
Figure 4. Using the MUXOUT/ADCIN loop of the LTC1598 to form a PGA with
eight inverting gains
Figure 5. Using the LTC1598 and LTC1391 as an 8-channel, differential 12-bit ADC system:
opening the indicated connection and shorting the dashed connection daisy-chains the
external and internal MUXes, increasing channel-selection flexibility.
continued on page 20
XUM
lennahC
gnitrevninoN
niaG
gnitrevnI
niaG
011
122
244
388
46161
52323
64646
7821821
Table 1. PGA gain for each MUX channel of
Figures 3 and 4
Linear Technology Magazine • February 1997
17
DESIGN FEATURES
+
+
+
+
+
VFB
5
RSENSE
(OPTIONAL)
LBI
1µA
RUN
LBOUT
LB
V
C
1SHOT
VFB
IN
1.23V
REFERENCE
ON
LBI
GND
READY
1.23V
1474_01.eps
5µS
ON
ON
100mV VIN
VIN
VOUT
SENSE
SW
1×25×
CONNECTION NOT PRESENT IN LTC1474
CONNECTION PRESENT IN LTC1474 ONLY
OUT
LTC1474 and LTC1475 High Efficiency
Switching Regulators Draw Only
10µA Supply Current by Greg Dittmer
Introduction
Maximizing battery life, one of the key
design requirements for all battery-
powered products, is now easier with
Linear Technology’s new family of
ultralow quiescent current, high
efficiency step-down regulator ICs,
the LTC1474 and LTC1475. The
LTC1474/LTC1475 are step-down
regulators with on-chip P-channel
MOSFET power switches. These regu-
lators draw only 10µA supply current
at no load while maintaining the out-
put voltage. With the on-chip switch
(1.3 at V
IN
= 10V), only four external
components are necessary to make a
complete, high efficiency (up to 92%)
step-down regulator. Low component
count and the LTC1474/LTC1475’s
tiny MSOP packages provide a mini-
mum-area solution to meet the limited
space requirements of portable appli-
cations. Wide supply voltage range
(3V–18V) and 100% duty cycle capa-
bility for low dropout allow maximum
energy to be extracted from the bat-
tery, making the LTC1474/LTC1475
ideal for moderate current (up to
300mA) battery-powered applications.
The peak inductor current is pro-
grammable via an optional current
sense resistor to allow the design to
be optimized for a particular applica-
tion and to provide short-circuit pro-
tection and excellent start-up
behavior. Other features include Burst
Mode™ operation to maintain high
efficiency over almost four decades of
load current, an on-chip low-battery
comparator and a shutdown mode to
further reduce supply current to 6µA.
The LTC1475 provides on/off control
with push-button switches for use in
handheld products.
The LTC1474/LTC1475 are avail-
able in adjustable output voltage
versions, in 8-pin MSOP and SO
packages.
Figure 1. LTC1474/LTC1475 functional block diagram
Linear Technology Magazine • February 1997
18
DESIGN FEATURES
1474_02.eps
MODE
RUN
ONE-SHOT
1.23V REFERENCE
INDUCTOR
CURRENT
CURRENT
COMPARATOR
VOLTAGE
COMPARATOR
LOW-BATTERY
COMPARATOR
SHUTDOWNSLEEP5µS OFF-TIMEON
OFFOFFOFFON (100µA)
OFFOFFON (10µA)ON (5µA)
OFFON (5µA)
ON (4µA)
ON (1µA)
High Performance on a
Microampere Budget
The functional block diagram, shown
in Figure 1, provides a study in power
management. LTC1474/LTC1475
control the output voltage by charg-
ing the output capacitor in short burst
cycles using Burst Mode operation.
The peak current in each burst cycle
(up to 400mA) is set by the external
sense resistor. As the load increases,
the frequency of the burst cycles in-
creases (up to a maximum of 170kHz)
to maintain the charge in the output
capacitor. The burst cycle begins when
the output voltage falls below the
lower threshold of the voltage com-
parator (V). The P-channel power
switch turns on to ramp the inductor
current up until either the current
comparator (C) trips on peak current
or the voltage comparator trips on the
upper voltage threshold. At this time
the one-shot is triggered and begins
the 5µs off-time period, during which
the switch is turned off and inductor
current ramps down. If, at the end of
the off-time, the output voltage is
below the upper voltage comparator
threshold, the switch is turned on
again to begin another cycle. If the
upper voltage threshold is exceeded,
however, the switch remains off and
the output capacitor supplies the load
current. The switch remains off until
the load current discharges the out-
put capacitor below the lower voltage
threshold.
The ultralow supply current and
very high efficiency at light loads are
achieved by powering only those func-
tions that are necessary at any given
time. Figure 2 is a summary of the
current used by each of the functions
in each of the different operating
modes. During sleep mode, when the
output capacitor is supplying the load,
only the 1.23V reference, the voltage
comparator and low-battery compara-
tor are on; together they draw only
10µA of supply current to perform
their functions. These three functions
are on at all times except during
shutdown. During shutdown, the volt-
age comparator is turned off to save
an additional 5µA. The current com-
parator, which, as a result of its speed
requirement necessarily draws more
current, is only turned on during the
switch on-time, when it is needed to
monitor the switch current. When the
current or voltage comparator trips,
the current comparator is turned off
and the one-shot timer is triggered,
drawing 10µA during its 5µs time-
out period. When the one-shot times
out it turns off, reducing the supply
current to the 10µA needed for the
voltage comparator, reference and low-
battery comparator, until a new burst
cycle begins.
In Control of
Inductor Current
Excessive peak inductor current can
be a liability. Lower peak current
offers the advantages of smaller volt-
age ripple (V = I
PEAK
× ESR), lower
noise and less stress on alkaline bat-
teries and other circuit components.
Also, lower peaks allow the use of
inductors with smaller physical size.
The LTC1474/LTC1475 provide flex-
ibility by allowing the peak switch/
inductor current to be programmed
with an optional sense resistor to
provide just enough to meet the load
requirement. Without a sense resis-
tor (that is, with pins 6 and 7 shorted)
the current limit defaults to its maxi-
mum of 400mA. Using the default
current limit eliminates the need for a
sense resistor and associated de-
coupling capacitor.
A sense MOSFET (a portion of the
main power MOSFET) is used to di-
vert a sample (about 5%) of the switch
current through the internal 5 sense
resistor. The internal current com-
parator monitors the voltage drop
across the series combination of the
internal and external sense resistors
and trips when this voltage drop ex-
ceeds 100mV. This results in a peak
current of I
PEAK
= 0.1/(0.25 + R
SENSE
)
+ 0.2 × (V
IN
– V
OUT
)/L. The second
term in the above equation is the
result of overshoot of the peak cur-
rent due to delays in the current
comparator and must be taken into
account at lower inductances and
higher supply voltages to guarantee
that maximum current ratings of the
inductor and switch are not exceeded.
Note that worst case will occur during
a short circuit, when V
OUT
= 0.
3.3V/200mA
Step-Down Regulator
A typical application circuit using the
LTC1474 is shown in Figure 3. This
circuit supplies a 200mA load at 3.3V
with an input supply range of 4V–18V
(3.3V at no load). The 0.1 sense
resistor reduces the peak current to
about 285mA, which is the minimum
level necessary to meet the 200mA
load current requirement with a
Figure 2. Supply current breakdown during each operational mode
Linear Technology Magazine • February 1997
19
DESIGN FEATURES
100µH inductor. The peak can be
reduced further if a higher value in-
ductor is used. Since the output
capacitor dominates the output volt-
age ripple, an AVX TPS series low ESR
(150m ) output capacitor is used to
provide a good compromise between
size and low ESR. With this capacitor
the output ripple is less than 50mV.
Efficiency Considerations
The efficiency curves for the 3.3V/
200mA regulator at various supply
voltages are shown in Figure 4. Note
the flatness of the curves over the
upper three decades of load current
and that the efficiency remains high
down to extremely light loads. Effi-
ciency at light loads depends on low
quiescent current. The curves are flat
because all significant sources of loss
except for the 10µA standby cur-
rent—I
2
R losses in the switch, catch
diode losses, gate charge losses to
turn on the switch and burst cycle DC
supply current losses—are identical
during each burst cycle. The only
variable is the rate at which the burst
cycles occur. Since burst frequency is
proportional to load, the loss as a
percentage of load remains relatively
constant. The efficiency drops off as
the load decreases below about 1mA
because the non-load-dependent
10µA standby current loss then con-
stitutes a more significant percentage
of the output power. This loss is pro-
portional to V
IN
and thus its effect is
more pronounced at higher V
IN
.
Care must be used in selecting the
catch diode to maximize both low and
high current efficiency. Low reverse
leakage current is critical for maxi-
mizing low current efficiency because
the leakage can potentially approach
the magnitude of the LTC1474/
LTC1475 supply current. Low for-
ward drop is critical for high current
efficiency because loss is proportional
to forward drop. These are conflicting
parameters, but the MBR0530 0.5A
Schottky diode used in the Figure 3 is
a good compromise. Lower induc-
tances also help by minimizing DCR
without increasing the inductor size.
However, lower inductances also re-
duce the maximum available output
power for a given I
PEAK
due to the fixed
s off-time and may also increase
the peak current overshoot due to
high di/dt (see formula for I
PEAK
).
LTC1475 Push-Button
On/Off Operation
The LTC1475 provides the option of
push-button control of run and shut-
down modes for handheld products.
In contrast to the LTC1474’s run/
shutdown mode, which is controlled
by a voltage level at the RUN pin
(ground = shutdown, open/high =
run), the LTC1475 run/shutdown
mode is controlled by an internal S/R
flip-flop that is set (run mode) by a
momentary ground at the RUN pin
and reset (shutdown mode) by a mo-
mentary ground at the LBI pin (see
Figure 5). This provides simple on/off
control with two push-button
switches. The simplest implementa-
tion of this function is shown in Figure
6, with normally open push-button
switches connected to the RUN and
LBI pins. Note that because the switch
on LBI is normally open, it doesn’t
+
10pF
1.69M
1M
100k
1474_03.eps
RUN
V
IN
SENSE
SW
V
FB
LBO
0.1µF10µF
25V
1000pF
LBI
LTC1474
LBI LBO
MBR0530
GND
8
7
6
5
1
V
OUT
3.3V/200mA V
IN
4V-18V
RUN
D:
L:
C
OUT
:
C
IN
:
MBR0530
SUMIDA CDRH74
TPSC107006R0150
THC50EIE106Z
2
3
4
0.1
100µF
6.3V
L
100µH
LOAD CURRENT (mA)
50
60
70
80
90
100
EFFICIENCY (%)
200
1474_04.eps
0.02 0.2 2 20
L= 100 µH
V
OUT
= 3.3V
R
SENSE
= 0.1
V
IN
= 5V V
IN
= 10V
V
IN
= 15V
LBI
RUN
MODE RUN SHUTDOWN RUN
MODE RUN SHUTDOWN RUN
RUN
LTC1474
LTC1475
RUN OVERRIDES SHUTDOWN 
WHILE RUN IS LOW
+
100k
10µF
1474_05.eps
V
IN
SENSE
SW
OFF
V
FB
V
FB
V
OUT
V
FB
LBO
LTC1475
V
BATT
V
BATT
GND
8
7
6
5
1
2
3
4
ON RUN
LBI/SD
1M 2.2M
100µH
100µF
Figure 3. LTC1474 3.3V/200mA step-down regulator Figure 4. Efficiency vs load for Figure 3’s
circuit
Figure 5. Comparison of RUN/SHUTDOWN
operation for the LTC1474 and LTC1475 Figure 6. LTC1475 step-down regulator with push-button on/off control
Linear Technology Magazine • February 1997
20
DESIGN FEATURES
affect the normal operation of this
input to the low-battery comparator.
With a resistor divider network con-
nected to the LBI to monitor the input
supply voltage level, the voltage at
this pin will normally be above the
low-battery trip threshold of 1.23V.
When this pin is pulled below 0.7V by
depressing the switch, the internal
flip-flop is reset to invoke shutdown.
Figure 7 shows an example of push-
button on/off control of a LTC1475
microcontroller application with a
single push button. The push button
is connected to the microcontroller as
a discrete input so that the
microcontroller can monitor the state
of the push button. The LTC1475 LBI
pin is connected to one of the
microcontroller’s open-drain discrete
outputs so that it can force the
LTC1475 off when it detects a
depressed push button. Because the
LTC1475 supplies power to
the microcontroller, once the micro-
controller is off, it can no longer turn
the LTC1475 back on. However, since
the push button is also connected
directly to the RUN pin, the LTC1475
can be turned back on directly from
the push button without the micro-
controller. The LTC1475 then powers
up the microcontroller. The discrete
inputs of most microcontrollers have
a reverse biased diode between the
input and supply; thus a blocking
diode with less than 1µA leakage is
necessary to prevent the powered
down microcontroller from pulling
down on the RUN pin.
Conclusion
The LTC1474 and LTC1475 ultralow
quiescent current step-down regula-
tor ICs provide a perfect solution for
low to moderate current (up to 300mA)
battery-powered applications where
high efficiency and maximizing bat-
tery life are critical. The 10µA no-load
supply current requirement ensures
that little battery energy is wasted on
the regulator. The internal P-channel
power switch, MSOP package and the
need for as few as four additional
components result in a very compact
solution, and the current program-
mability and wide supply-voltage
range provide the flexibility neces-
sary to optimize the design for a variety
of applications.