110 100 10k 100k
FREQUENCY (Hz)
1
10
100
1k
VOLTAGE NOISE (nV/
Hz)
V+ = 5.5V
V+ = 2.5V
CCM
IIN
RF
VOUT
+
-
+
-
VB
CF
CD
VOUT
IIN - RF
=
CIN = CD + CCM
LMV791, LMV792
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
LMV791/LMV792 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers with
Shutdown
Check for Samples: LMV791,LMV792
1FEATURES DESCRIPTION
The LMV791 (Single) and the LMV792 (Dual) low
2(Typical 5V Supply, Unless Otherwise Noted) noise, CMOS input operational amplifiers offer a low
• Input Referred Voltage Noise 5.8 nV/√Hz input voltage noise density of 5.8 nV/√Hz while
• Input Bias Current 100 fA consuming only 1.15 mA (LMV791) of quiescent
current. The LMV791 and LMV792 are unity gain
• Unity Gain Bandwidth 17 MHz stable op amps and have gain bandwidth of 17 MHz.
• Supply Current per Channel Enable Mode The LMV791/ LMV792 have a supply voltage range
– LMV791 1.15 mA of 1.8V to 5.5V and can operate from a single supply.
The LMV791/LMV792 each feature a rail-to-rail
– LMV792 1.30 mA output stage capable of driving a 600Ωload and
• Supply Current per Channel in Shutdown sourcing as much as 60 mA of current.
Mode 0.02 µA The LMV791 family provides optimal performance in
• Rail-to-Rail Output Swing low voltage and low noise systems. A CMOS input
– @ 10 kΩLoad 25 mV from Rail stage, with typical input bias currents in the range of
– @ 2 kΩLoad 45 mV from Rail a few femtoAmperes, and an input common mode
voltage range which includes ground make the
• Ensured 2.5V and 5.0V Performance LMV791 and the LMV792 ideal for low power sensor
• Total Harmonic Distortion 0.01% @1 kHz, 600Ωapplications. The LMV791 family has a built-in enable
• Temperature Range −40°C to 125°C feature which can be used to optimize power
dissipation in low power applications.
APPLICATIONS The LMV791/LMV792 are manufactured using TI’s
• Photodiode Amplifiers advanced VIP50 process and are offered in a 6-pin
SOT and a 10-pin VSSOP package respectively.
• Active Filters and Buffers
• Low Noise Signal Processing
• Medical Instrumentation
• Sensor Interface Applications
Typical Application
Figure 1. Photodiode Transimpedance Amplifier Figure 2. Input Referred Voltage Noise vs.
Frequency
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2005–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
ESD Tolerance(3) Human Body Model 2000V
Machine Model 200V
Charge-Device Model 1000V
VIN Differential ±0.3V
Supply Voltage (V+– V−) 6.0V
Input/Output Pin Voltage V++0.3V, V−−0.3V
Storage Temperature Range −65°C to 150°C
Junction Temperature(4) +150°C
Soldering Information Infrared or Convection (20 sec) 235°C
Wave Soldering Lead Temp (10 sec) 260°C
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see the Electrical Characteristics Tables.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Human Body Model is 1.5 kΩin series with 100 pF. Machine Model is 0Ωin series with 200 pF
(4) The maximum power dissipation is a function of TJ(MAX),θJA. The maximum allowable power dissipation at any ambient temperature is
PD= (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.
Operating Ratings(1)
Temperature Range(2) −40°C to 125°C
Supply Voltage (V+– V−)
−40°C ≤TA≤125°C 2.0V to 5.5V
0°C ≤TA≤125°C 1.8V to 5.5V
Package Thermal Resistance (θJA)(2) 6-Pin SOT 170°C/W
10-Pin VSSOP 236°C/W
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see the Electrical Characteristics Tables.
(2) The maximum power dissipation is a function of TJ(MAX),θJA. The maximum allowable power dissipation at any ambient temperature is
PD= (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.
2.5V Electrical Characteristics
Unless otherwise specified, all limits are ensured for TA= 25°C, V+= 2.5V, V−= 0V, VCM = V+/2 = VO, VEN = V+.Boldface
limits apply at the temperature extremes.
Symbol Parameter Conditions Min Typ Max Units
(1) (2) (1)
VOS Input Offset Voltage 0.1 ±1.35 mV
±1.65
TC VOS Input Offset Voltage Temperature Drift LMV791(3) −1.0 μV/°C
LMV792(3) −1.8
IBInput Bias Current VCM = 1.0V(4) (5) −40°C ≤TA≤85 °C 0.05 1
25 pA
−40°C ≤TA≤85 °C 0.05 1
100
IOS Input Offset Current VCM = 1.0V(5) 10 fA
(1) Limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlations using the
statistical quality control (SQC) method.
(2) Typical values represent the parametric norm at the time of characterization.
(3) Offset voltage average drift is determined by dividing the change in VOS by temperature change.
(4) Positive current corresponds to current flowing into the device.
(5) This parameter is specified by design and/or characterization and is not tested in production.
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
2.5V Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured for TA= 25°C, V+= 2.5V, V−= 0V, VCM = V+/2 = VO, VEN = V+.Boldface
limits apply at the temperature extremes.
Symbol Parameter Conditions Min Typ Max Units
(1) (2) (1)
CMRR Common Mode Rejection Ratio 0V ≤VCM ≤1.4V 80 94 dB
75
PSRR Power Supply Rejection Ratio 2.0V ≤V+≤5.5V, VCM = 0V 80 100
75 dB
1.8V ≤V+≤5.5V, VCM = 0V 80 98
CMVR Common Mode Voltage Range CMRR ≥60 dB −0.3 1.5 V
CMRR ≥55 dB −0.3 1.5
AVOL Open Loop Voltage Gain VOUT = 0.15V to 2.2V, LMV791 85 98
RLOAD = 2 kΩto V+/2 80
LMV792 82 92 dB
78
VOUT = 0.15V to 2.2V, 88 110
RLOAD = 10 kΩto V+/2 84
VOUT Output Voltage Swing High RLOAD = 2 kΩto V+/2 25 75
82
RLOAD = 10 kΩto V+/2 20 65
71 mV from
either rail
Output Voltage Swing Low RLOAD = 2 kΩto V+/2 30 75
78
RLOAD = 10 kΩto V+/2 15 65
67
IOUT Output Current Sourcing to V−35 47
VIN = 200 mV(6) 28 mA
Sinking to V+7 15
VIN = –200 mV(6) 5
ISSupply Current per Amplifier Enable Mode LMV791 0.95 1.30
VEN ≥2.1V 1.65 mA
LMV792 1.1 1.50
per channel 1.85
Shutdown Mode, VEN < 0.4 0.02 1 μA
per channel 5
SR Slew Rate AV= +1, Rising (10% to 90%) 8.5 V/μs
AV= +1, Falling (90% to 10%) 10.5
GBW Gain Bandwidth 14 MHz
enInput Referred Voltage Noise Density f = 1 kHz 6.2 nV/√Hz
inInput Referred Current Noise Density f = 1 kHz 0.01 pA/√Hz
ton Turn-on Time 140 ns
toff Turn-off Time 1000 ns
VEN Enable Pin Voltage Range Enable Mode 2.1 2 to 2.5 V
Shutdown Mode 0 to 0.5 0.4
IEN Enable Pin Input Current Enable Mode VEN = 2.5V (7) 1.5 3 μA
Shutdown Mode VEN = 0V (7) 0.003 0.1
THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV= 1, RLOAD = 600Ω0.01 %
(6) The short circuit test is a momentary test, the short circuit duration is 1.5 ms.
(7) Positive current corresponds to current flowing into the device.
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5V Electrical Characteristics
Unless otherwise specified, all limits are ensured for TA= 25°C, V+= 5V, V−= 0V, VCM = V+/2 = VO, VEN = V+.Boldface limits
apply at the temperature extremes.
Symbol Parameter Conditions Min Typ Max Units
(1) (2) (1)
VOS Input Offset Voltage 0.1 ±1.35 mV
±1.65
TC VOS Input Offset Voltage Temperature Drift LMV791(3) −1.0 μV/°C
LMV792(3) −1.8
IBInput Bias Current VCM = 2.0V(4) (5) −40°C ≤TA≤85°C 0.1 1
25 pA
−40°C ≤TA≤125°C 0.1 1
100
IOS Input Offset Current VCM = 2.0V(5) 10 fA
CMRR Common Mode Rejection Ratio 0V ≤VCM ≤3.7V 80 100 dB
75
PSRR Power Supply Rejection Ratio 2.0V ≤V+≤5.5V, VCM = 0V 80 100
75 dB
1.8V ≤V+≤5.5V, VCM = 0V 80 98
CMVR Common Mode Voltage Range CMRR ≥60 dB −0.3 4 V
CMRR ≥55 dB −0.3 4
AVOL Open Loop Voltage Gain VOUT = 0.3V to 4.7V, LMV791 85 97
RLOAD = 2 kΩto V+/2 80
LMV792 82 89 dB
78
VOUT = 0.3V to 4.7V, 88 110
RLOAD = 10 kΩto V+/2 84
VOUT Output Voltage Swing High RLOAD = 2 kΩto V+/2 35 75
82
RLOAD = 10 kΩto V+/2 25 65
71
Output Voltage Swing Low RLOAD = 2 kΩto V+/2 LMV791 42 75 mV from
78 either rail
LMV792 45 80
83
RLOAD = 10 kΩto V+/2 20 65
67
IOUT Output Current Sourcing to V−45 60
VIN = 200 mV(6) 37 mA
Sinking to V+10 21
VIN = –200 mV(6) 6
ISSupply Current per Amplifier Enable Mode LMV791 1.15 1.40
VEN ≥4.6V 1.75 mA
LMV792 1.30 1.70
per channel 2.05
Shutdown Mode (VEN ≤0.4V) 0.14 1 μA
5
SR Slew Rate AV= +1, Rising (10% to 90%) 6.0 9.5 V/μs
AV= +1, Falling (90% to 10%) 7.5 11.5
GBW Gain Bandwidth 17 MHz
(1) Limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlations using the
statistical quality control (SQC) method.
(2) Typical values represent the parametric norm at the time of characterization.
(3) Offset voltage average drift is determined by dividing the change in VOS by temperature change.
(4) Positive current corresponds to current flowing into the device.
(5) This parameter is specified by design and/or characterization and is not tested in production.
(6) The short circuit test is a momentary test, the short circuit duration is 1.5 ms.
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Product Folder Links: LMV791 LMV792
OUTPUT
V-
+IN
V+
-IN
+-
1
2
3
5
4
6
EN
OUT A
IN A-
IN A+
V-
EN A
1
2
3
4
5EN B
IN B+
IN B-
OUT B
V+
6
7
8
9
10
+
-
+
-
LMV791, LMV792
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
5V Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured for TA= 25°C, V+= 5V, V−= 0V, VCM = V+/2 = VO, VEN = V+.Boldface limits
apply at the temperature extremes.
enInput Referred Voltage Noise Density f = 1 kHz 5.8 nV/√Hz
inInput Referred Current Noise Density f = 1 kHz 0.01 pA/√Hz
ton Turn-on Time 110 ns
toff Turn-off Time 800 ns
VEN Enable Pin Voltage Range Enable Mode 4.6 4.5 to 5 V
Shutdown Mode 0 to 0.5 0.4
IEN Enable Pin Input Current Enable Mode VEN = 5.0V(7) 5.6 10 μA
Shutdown Mode VEN = 0V(7) 0.005 0.2
THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV= 1, RLOAD = 600Ω0.01 %
(7) Positive current corresponds to current flowing into the device.
Connection Diagram
Figure 3. 6-Pin SOT Figure 4. 10-Pin VSSOP
Top View Top View
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Product Folder Links: LMV791 LMV792
-40°C
-0.3 0.4 1.1 1.8
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
0.55
0.6
VOS (mV)
VCM (V)
V+ = 2.5V
125°C
25°C
-0.3 0.6 1.5 2.4 3.3 4.2
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
VOS (mV)
VCM (V)
-40°C
25°C
125°C
V+ = 5V
1.5 2.5 3.5 4.5 5.5
0
1
1.2
1.4
1.6
1.8
SUPPLY CURRENT (PA)
V+ (V)
0.8
0.6
0.4
0.2
125°C
25°C
-40°C
6.0
-0.3 0 0.3 0.6 0.9 1.2
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
VOS (mV)
VCM (V)
125°C
25°C
-40°C
V+ = 1.8V
1.5 2.5 3.5 4.5 5.5 6
0
0.4
0.8
1.2
1.6
2
SUPPLY CURRENT (mA)
V+ (V)
25°C
-40°C
125°C
1.5 2.5 3.5 4.5 5.5 6.0
SUPPLY CURRENT (mA)
V+ (V)
0
0.4
0.8
1.2
1.6
2
25°C
-40°C
125°C
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
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Typical Performance Characteristics
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Supply Current vs. Supply Voltage (LMV791) Supply Current vs. Supply Voltage (LMV792)
Figure 5. Figure 6.
Supply Current vs.\ Supply Voltage in Shutdown Mode VOS vs. VCM
Figure 7. Figure 8.
VOS vs. VCM VOS vs. VCM
Figure 9. Figure 10.
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Product Folder Links: LMV791 LMV792
-40°C
25°C
125°C
0 0.5 1 1.5 2 2.5
-0.1
0.1
0.3
0.5
0.7
0.9
1.5
SUPPLY CURRENT (mA)
ENABLE PIN VOLTAGE (V)
1.1
1.3 V+ = 2.5V
0 1 2 3 4 5
-0.1
0.4
0.9
1.4
1.9
2.4
SUPPLY CURRENT (mA)
ENABLE PIN VOLTAGE (V)
125°C
25°C
-40°C
-40°C
125°C
V+ = 5V
1.5 2.5 3.5 4.5 5.5 6.0
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
VOS (mV)
V+ (V)
-40°C
125°C
25°C
13
SLEW RATE (V/Ps)
1.8 2.3 2.8 3.3 3.8 4.3 4.8 5.3
6
7
8
9
10
11
V+ (V)
12
RISING
FALLING
5.5
LMV791, LMV792
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
VOS vs. Supply Voltage Slew Rate vs. Supply Voltage
Figure 11. Figure 12.
Supply Current vs. Enable Pin Voltage (LMV791) Supply Current vs. Enable Pin Voltage(LMV791)
Figure 13. Figure 14.
Supply Current vs. Enable Pin Voltage (LMV792) Supply Current vs. Enable Pin Voltage (LMV792)
Figure 15. Figure 16.
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0 1 2 3 4 5
0
5
10
15
20
25
30
ISINK (mA)
VOUT (V)
125°C
25°C
-40°C
-40°C
0 1 2 3 4 5
0
10
20
30
40
50
60
70
ISOURCE (mA)
VOUT (V)
125°C
25°C
1 2 3 4 5 6
0
5
10
15
20
25
30
35
ISINK (mA)
V+ (V)
25°C
125°C
-40°C
0 1 2 3 4
-3
-2.5
-2
-1.5
-1
-0.5
0
0.5
1
1.5
IBIAS (pA)
VCM (V)
-40°C
25°C
V+ = 5V
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Input Bias Current vs. VCM Input Bias Current vs. VCM
Figure 17. Figure 18.
Sourcing Current vs. Supply Voltage Sinking Current vs. Supply Voltage
Figure 19. Figure 20.
Sourcing Current vs. Output Voltage Sinking Current vs. Output Voltage
Figure 21. Figure 22.
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1.8 2.5 3.2 3.9 4.6 5.3 6
V+ (V)
0
20
40
60
80
100
120
VOUT FROM RAIL (mV)
RLOAD = 600:
-40°C
25°C 125°C
1.8 2.5 3.2 3.9 4.6 5.3 6
0
10
20
30
40
50
60
70
80
90
100
VOUT FROM RAIL (mV)
V+ (V)
125°C
25°C
-40°C
RLOAD = 600:
1.8 2.5 3.2 3.9 4.6 5.3 6
0
5
10
15
20
25
30
35
40
45
50
VOUT FROM RAIL (mV)
V+ (V)
125°C
25°C
-40°C
RLOAD = 2 k:
1.8 2.5 3.2 3.9 4.6 5.3 6
V+ (V)
0
5
10
15
20
25
VOUT FROM RAIL (mV)
-40°C
25°C
125°C
RLOAD = 10 k:
1.8 2.5 3.2 3.9 4.6 5.3 6
V+ (V)
0
5
10
15
20
25
30
35
40
VOUT FROM RAIL (mV)
125°C
25°C
-40°C
RLOAD = 10 k:
LMV791, LMV792
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage
Figure 23. Figure 24.
Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage
Figure 25. Figure 26.
Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage
Figure 27. Figure 28.
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0.02 0.2 2
OUTPUT AMPLITUDE (V)
-140
-120
-100
-80
-60
-40
-20
0
THD+N (dB)
V+ = 2.75V
V- = -2.75V
AV = +2
RLOAD = 100 k:
RLOAD = 600:
4
10 100 1k 10k 100k
FREQUENCY (Hz)
0
0.001
0.002
0.003
0.004
0.005
0.006
THD+N (%)
RL = 600:
RL = 100 k:
V+ = 1.2V
V- = 0.6V
VO = 0.9 VPP
AV = +2
OUTPUT AMPLITUDE (V)
0.02 0.2 2
-120
-100
-80
-60
-40
-20
0
THD+N (dB)
RLOAD = 600:
RLOAD = 100 k:
V+ = 1.2V
V- = -0.6V
AV = +2
110 100 10k 100k
FREQUENCY (Hz)
1
10
100
1k
VOLTAGE NOISE (nV/
Hz)
V+ = 5.5V
V+ = 2.5V
400 nV/DIV
1S/DIV
VS = ±2.5V
VCM = 0.0V
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Input Referred Voltage Noise vs. Frequency Time Domain Voltage Noise
Figure 29. Figure 30.
Overshoot and Undershoot vs. CLOAD THD+N vs. Peak-to-Peak Output Voltage (VOUT)
Figure 31. Figure 32.
THD+N vs. Peak-to-Peak Output Voltage (VOUT) THD+N vs. Frequency
Figure 33. Figure 34.
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160
1k 100k 100M
FREQUENCY (Hz)
0
60
CROSSTALK REJECTION RATION (dB)
10M
1M
10k
120
100
40
20
80
140
10 mV/DIV
200 ns/DIV
INPUT = 20 mVPP
f = 1 MHz
V+ = 2.5V
10 1k 10k 100k 1M 10M 100M
FREQUENCY (Hz)
0.01
0.1
1
10
100
OUTPUT IMPEDANCE (:)
100
-40
-20
0
20
60
80
100
120
GAIN (dB)
40
-60
PHASE (°)
-40
-20
0
20
60
80
100
120
40
-60
FREQUENCY (Hz)
10k 100k 1M 10M 100M
PHASE
GAIN
RLOAD = 600:10k: 10 M:
1k 10k 100k 1M 10M 100M
-40
-20
0
20
60
80
100
120
GAIN (dB)
FREQUENCY (Hz)
40
-60
PHASE (°)
-40
-20
0
20
60
80
100
120
40
-60
CL = 20 pF
CL = 50 pF
CL = 100 pF
GAIN
CL = 100 pF
CL = 50 pF
CL = 20 pF
PHASE
10 100 1k 10k 100k
FREQUENCY (Hz)
0
0.001
0.002
0.003
0.004
0.005
0.006
THD+N (%)
RL = 600:
RL = 100 k:
V+ = 2.5V
V- = 2.5V
VO = 4 VPP
AV = +2
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SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
THD+N vs. Frequency Open Loop Gain and Phase with Capacitive Load
Figure 35. Figure 36.
Open Loop Gain and Phase with Resistive Load Closed Loop Output Impedance vs. Frequency
Figure 37. Figure 38.
Crosstalk Rejection Small Signal Transient Response, AV= +1
Figure 39. Figure 40.
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10 1k 100k 10M
FREQUENCY (Hz)
-100
-60
-40
0
POSITIVE PSRR (dB)
1M10k
100
-20
-80 1.8V
5.5V
-10
0
10
20
30
40
50
PHASE MARGIN (°)
10 100 1000
CLOAD (pF)
RLOAD = 10 M:
V+ = 5V
RLOAD = 10 k:
RLOAD = 600:
-10
0
10
20
30
40
50
PHASE MARGIN (°)
10 100 1000
CLOAD (pF)
RLOAD = 10 M:
V+ = 2.5V
RLOAD = 10 k:
RLOAD = 600:
200 mV/DIV
800 ns/DIV
INPUT = 1 VPP
f = 200 kHz
V+ = 5V
800 ns/DIV
200 mV/DIV
INPUT = 1 VPP
f = 200 kHz
V+ = 2.5V
10 mV/DIV
200 ns/DIV
INPUT = 20 mVPP
f = 1 MHz
V+ = 5V
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Large Signal Transient Response, AV= +1 Small Signal Transient Response, AV= +1
Figure 41. Figure 42.
Large Signal Transient Response, AV= +1 Phase Margin vs. Capacitive Load (Stability)
Figure 43. Figure 44.
Phase Margin vs. Capacitive Load (Stability) Positive PSRR vs. Frequency
Figure 45. Figure 46.
12 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMV791 LMV792
0 1 2 3 4
0
5
10
15
20
25
CCM (pF)
VCM (V)
V+ = 5V
-120
-100
-80
-60
-40
-20
NEGATIVE PSRR (dB)
10 100 1k 10k 100k 1M 10M
FREQUENCY (Hz)
V+ = 1.8V
V+ = 5.5V
10 1k 1M
FREQUENCY (Hz)
0
40
120
CMRR (dB)
100k
10k
100
100
60
20
80
V+ = 2.5V
V+ = 5V
LMV791, LMV792
www.ti.com
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Typical Performance Characteristics (continued)
Unless otherwise specified, TA= 25°C, V–= 0, V+= Supply Voltage = 5V, VCM = V+/2, VEN = V+.
Negative PSRR vs. Frequency CMRR vs. Frequency
Figure 47. Figure 48.
Input Common Mode Capacitance vs. VCM
Figure 49.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LMV791 LMV792
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
APPLICATION INFORMATION
ADVANTAGES OF THE LMV791/LMV792
Wide Bandwidth at Low Supply Current
The LMV791 and LMV792 are high performance op amps that provide a unity gain bandwidth of 17 MHz while
drawing a low supply current of 1.15 mA. This makes them ideal for providing wideband amplification in portable
applications. The enable and shutdown feature can also be used to design more power efficient systems that
offer wide bandwidth and high performance while consuming less average power.
Low Input Referred Noise and Low Input Bias Current
The LMV791/LMV792 have a very low input referred voltage noise density (5.8 nV/√Hz at 1 kHz). A CMOS input
stage ensures a small input bias current (100 fA) and low input referred current noise (0.01 pA/√Hz). This is very
helpful in maintaining signal fidelity, and makes the LMV791 and LMV792 ideal for audio and sensor based
applications.
Low Supply Voltage
The LMV791 and the LMV792 have performance ensured at 2.5V and 5V supply. The LMV791 family is ensured
to be operational at all supply voltages between 2.0V and 5.5V, for ambient temperatures ranging from −40°C to
125°C, thus utilizing the entire battery lifetime. The LMV791 and LMV792 are also ensured to be operational at
1.8V supply voltage, for temperatures between 0°C and 125°C. This makes the LMV791 family ideal for usage in
low-voltage commercial applications.
RRO and Ground Sensing
Rail-to-rail output swing provides maximum possible dynamic range at the output. This is particularly important
when operating at low supply voltages. An innovative positive feedback scheme is used to boost the current drive
capability of the output stage. This allows the LMV791 and the LMV792 to source more than 40 mA of current at
1.8V supply. This also limits the performance of the LMV791 family as comparators, and hence the usage of the
LMV791 and the LMV792 in an open-loop configuration is not recommended. The input common-mode range
includes the negative supply rail which allows direct sensing at ground in single supply operation.
Enable and Shutdown Features
The LMV791 family is ideal for battery powered systems. With a low supply current of 1.15 mA and a shutdown
current of 140 nA typically, the LMV791 and LMV792 allow the designer to maximize battery life. The enable pin
of the LMV791 and the LMV792 allows the op amp to be turned off and reduce its supply current to less than 1
μA. To power on the op amp the enable pin should be higher than V+- 0.5V, where V+is the positive supply. To
disable the op amp, the enable pin voltage should be less than V−+ 0.5V, where V−is the negative supply.
Small Size
The small footprint of the LMV791 and the LMV792 package saves space on printed circuit boards, and enables
the design of smaller electronic products, such as cellular phones, pagers, or other portable systems. Long
traces between the signal source and the opamp make the signal path susceptible to noise. By using a physically
smaller LMV791 and LMV792 package, the opamp can be placed closer to the signal source, reducing noise
pickup and increasing signal integrity.
CAPACITIVE LOAD TOLERANCE
The LMV791 and LMV792 can directly drive 120 pF in unity-gain without oscillation. The unity-gain follower is the
most sensitive configuration to capacitive loading. Direct capacitive loading reduces the phase margin of
amplifiers. The combination of the amplifier’s output impedance and the capacitive load induces phase lag. This
results in either an underdamped pulse response or oscillation. To drive a heavier capacitive load, the circuit in
Figure 50 can be used.
In Figure 50, the isolation resistor RISO and the load capacitor CLform a pole to increase stability by adding more
phase margin to the overall system. The desired performance depends on the value of RISO. The bigger the RISO
resistor value, the more stable VOUT will be. Increased RISO would, however, result in a reduced output swing and
short circuit current.
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Product Folder Links: LMV791 LMV792
+¨
©
§
¨
©
§
-1
2CIN
P1,2 = 1
R1
1
R2r1
R1
1
R2
+
2
-4 A0CIN
R2
-R2/R1
1 + s
¨
©
§
¨
©
§
+s2
A0
CIN R2
¨
©
§
¨
©
§
VOUT
VIN (s) =
A0 R1
R1 + R2
CIN
R1
R2
VOUT
+
-
+
-
VIN
+
-
VOUT
VIN
R2
R1
AV = - = -
CF
LMV791, LMV792
www.ti.com
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Figure 50. Isolation of CLto Improve Stability
INPUT CAPACITANCE AND FEEDBACK CIRCUIT ELEMENTS
The LMV791 family has a very low input bias current (100 fA) and a low 1/f noise corner frequency (400 Hz),
which makes it ideal for sensor applications. However, to obtain this performance a large CMOS input stage is
used, which adds to the input capacitance of the op-amp, CIN. Though this does not affect the DC and low
frequency performance, at higher frequencies the input capacitance interacts with the input and the feedback
impedances to create a pole, which results in lower phase margin and gain peaking. This can be controlled by
being selective in the use of feedback resistors, as well as by using a feedback capacitance, CF. For example, in
the inverting amplifier shown in Figure 51, if CIN and CFare ignored and the open loop gain of the op amp is
considered infinite then the gain of the circuit is −R2/R1. An op amp, however, usually has a dominant pole, which
causes its gain to drop with frequency. Hence, this gain is only valid for DC and low frequency. To understand
the effect of the input capacitance coupled with the non-ideal gain of the op amp, the circuit needs to be
analyzed in the frequency domain using a Laplace transform.
Figure 51. Inverting Amplifier
For simplicity, the op amp is modelled as an ideal integrator with a unity gain frequency of A0. Hence, its transfer
function (or gain) in the frequency domain is A0/s. Solving the circuit equations in the frequency domain, ignoring
CFfor the moment, results in an expression for the gain shown in Equation 1.
(1)
It can be inferred from the denominator of the transfer function that it has two poles, whose expressions can be
obtained by solving for the roots of the denominator and are shown in Equation 2.
(2)
Equation 2 shows that as the values of R1and R2are increased, the magnitude of the poles, and hence the
bandwidth of the amplifier, is reduced. This theory is verified by using different values of R1and R2in the circuit
shown in Figure 50 and by comparing their frequency responses. In Figure 52 the frequency responses for three
different values of R1and R2are shown. When both R1and R2are 1 kΩ, the response is flattest and widest;
whereas, it narrows and peaks significantly when both their values are changed to 10 kΩor 30 kΩ. So it is
advisable to use lower values of R1and R2to obtain a wider and flatter response. Lower resistances also help in
high sensitivity circuits since they add less noise.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LMV791 LMV792
10k 100k 1M 10M
FREQUENCY (Hz)
-40
-30
-20
-10
0
10
20
GAIN (dB)
CF = 0 pF
CF = 5 pF
CF = 2 pF
R1, R2 = 30 k:
AV = -1
10k 100k 1M 10M 100M
FREQUENCY (Hz)
-25
-20
-15
-10
-5
0
5
10
15
GAIN (dB)
R1, R2 = 30 k:
AV = -1
R1, R2 = 10 k:
R1, R2 = 1 k:
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
Figure 52. Gain Peaking Caused by Large R1, R2
A way of reducing the gain peaking is by adding a feedback capacitance CFin parallel with R2. This introduces
another pole in the system and prevents the formation of pairs of complex conjugate poles which cause the gain
to peak. Figure 53 shows the effect of CFon the frequency response of the circuit. Adding a capacitance of 2 pF
removes the peak, while a capacitance of 5 pF creates a much lower pole and reduces the bandwidth
excessively.
Figure 53. Gain Peaking Eliminated by CF
AUDIO PREAMPLIFIER WITH BANDPASS FILTERING
With low input referred voltage noise, low supply voltage and low supply current, and a low harmonic distortion,
the LMV791 family is ideal for audio applications. Its wide unity gain bandwidth allows it to provide large gain for
a wide range of frequencies and it can be used to design a preamplifier to drive a load of as low as 600Ωwith
less than 0.01% distortion. Two amplifier circuits are shown in Figure 54 and Figure 55.Figure 54 is an inverting
amplifier, with a 10 kΩfeedback resistor, R2, and a 1kΩinput resistor, R1, and hence provides a gain of −10.
Figure 55 is a non-inverting amplifier, using the same values of R1and R2, and provides a gain of 11. In either of
these circuits, the coupling capacitor CC1 decides the lower frequency at which the circuit starts providing gain,
while the feedback capacitor CFdecides the frequency at which the gain starts dropping off. Figure 56 shows the
frequency response of the inverting amplifier with different values of CF.
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Product Folder Links: LMV791 LMV792
25
1100 10k 1M
FREQUENCY (Hz)
-20
-5
10
GAIN (dB)
100k1k
10
20
15
5
0
-10
-15
CF = 10 pF
CF = 100 pF
CF = 1 nF
CC2 RB1
RB2
CF
CC1
V+
AV = 1 + R2
R1
+
-
VIN
+
-
R1
1 k:
R2
10 k:
VOUT
+
-
= 11
CC1
+VOUT
+
-
-
CF
VIN
+
-
RB1
V+
RB2
CC2
R2
R1
AV = - = -10
R2
10 k:
R1
1 k:
LMV791, LMV792
www.ti.com
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Figure 54. Inverting Audio Preamplifier
Figure 55. Non-inverting Audio Preamplifier
Figure 56. Frequency Response of the Inverting Audio Preamplifier
TRANSIMPEDANCE AMPLIFIER
CMOS input op amps are often used in transimpedance applications as they have an extremely high input
impedance. A transimpedance amplifier converts a small input current into a voltage. This current is usually
generated by a photodiode. The transimpedance gain, measured as the ratio of the output voltage to the input
current, is expected to be large and wide-band. Since the circuit deals with currents in the range of a few nA, low
noise performance is essential. The LMV791/LMV792 are CMOS input op amps providing wide bandwidth and
low noise performance, and are hence ideal for transimpedance applications.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LMV791 LMV792
RFCINA0
CF = 2SRFA0
1 + 1 + 4S
CCM
IIN
RF
VOUT
+
-
+
-
VB
CF
CD
VOUT
IIN - RF
=
CIN = CD + CCM
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
Usually, a transimpedance amplifier is designed on the basis of the current source driving the input. A
photodiode is a very common capacitive current source, which requires transimpedance gain for transforming its
miniscule current into easily detectable voltages. The photodiode and amplifier’s gain are selected with respect to
the speed and accuracy required of the circuit. A faster circuit would require a photodiode with lesser
capacitance and a faster amplifier. A more sensitive circuit would require a sensitive photodiode and a high gain.
A typical transimpedance amplifier is shown in Figure 57. The output voltage of the amplifier is given by the
equation VOUT =−IINRF. Since the output swing of the amplifier is limited, RFshould be selected such that all
possible values of IIN can be detected.
The LMV791/LMV792 have a large gain-bandwidth product (17 MHz), which enables high gains at wide
bandwidths. A rail-to-rail output swing at 5.5V supply allows detection and amplification of a wide range of input
currents. A CMOS input stage with negligible input current noise and low input voltage noise allows the
LMV791/LMV792 to provide high fidelity amplification for wide bandwidths. These properties make the
LMV791/LMV792 ideal for systems requiring wide-band transimpedance amplification.
Figure 57. Photodiode Transimpedance Amplifier
As mentioned earlier, the following parameters are used to design a transimpedance amplifier: the amplifier gain-
bandwidth product, A0; the amplifier input capacitance, CCM; the photodiode capacitance, CD; the
transimpedance gain required, RF; and the amplifier output swing. Once a feasible RFis selected using the
amplifier output swing, these numbers can be used to design an amplifier with the desired transimpedance gain
and a maximally flat frequency response.
An essential component for obtaining a maximally flat response is the feedback capacitor, CF. The capacitance
seen at the input of the amplifier, CIN, combined with the feedback capacitor, RF, generate a phase lag which
causes gain-peaking and can destabilize the circuit. CIN is usually just the sum of CDand CCM. The feedback
capacitor CFcreates a pole, fPin the noise gain of the circuit, which neutralizes the zero in the noise gain, fZ,
created by the combination of RFand CIN. If properly positioned, the noise gain pole created by CFcan ensure
that the slope of the gain remains at 20 dB/decade till the unity gain frequency of the amplifier is reached, thus
ensuring stability. As shown in Figure 58, fPis positioned such that it coincides with the point where the noise
gain intersects the op amp’s open loop gain. In this case, fPis also the overall 3 dB frequency of the
transimpedance amplifier. The value of CFneeded to make it so is given by Equation 3. A larger value of CF
causes excessive reduction of bandwidth, while a smaller value fails to prevent gain peaking and instability.
(3)
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Product Folder Links: LMV791 LMV792
+
RF
-
CF
IF RA < < RF
CFc = ¨
©
§1 + RB
RA
¨
©
§
RA
CFc
RB
OP AMP
OPEN LOOP
GAIN
NOISE GAIN WITH NO CF
NOISE GAIN WITH CF
fZfPA0
fZ = 1
2SRFCIN
fP = A0
2SRF(CIN+CF)
GAIN
FREQUENCY
LMV791, LMV792
www.ti.com
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Figure 58. CFSelection for Stability
Calculating CFfrom Equation 3 can sometimes return unreasonably small values (<1 pF), especially for high
speed applications. In these cases, its often more practical to use the circuit shown in Figure 59 in order to allow
more reasonable values. In this circuit, the capacitance CF′is (1+ RB/RA) time the effective feedback capacitance,
CF. A larger capacitor can now be used in this circuit to obtain a smaller effective capacitance.
For example, if a CFof 0.5 pF is needed, while only a 5 pF capacitor is available, RBand RAcan be selected
such that RB/RA= 9. This would convert a CF′of 5 pF into a CFof 0.5 pF. This relationship holds as long as RA
<< RF.
Figure 59. Obtaining Small CFfrom large CF′
LMV791 AS A TRANSIMPEDANCE AMPLIFIER
The LMV791 was used to design a number of amplifiers with varying transimpedance gains and source
capacitances. The gains, bandwidths and feedback capacitances of the circuits created are summarized in
Table 1. The frequency responses are presented in Figure 60 and Figure 61. The feedback capacitances are
slightly different from the formula in Equation 3, since the parasitic capacitance of the board and the feedback
resistor RFhad to be accounted for.
Table 1.
Transimpedance, ATI CIN CF3 dB Frequency
470000 50 pF 1.5 pF 350 kHz
470000 100 pF 2.0 pF 250 kHz
470000 200 pF 3.0 pF 150 kHz
47000 50 pF 4.5 pF 1.5 MHz
47000 100 pF 6.0 pF 1 MHz
47000 200 pF 9.0 pF 700 kHz
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LMV791 LMV792
10k 100k 1M 10M
FREQUENCY (Hz)
50
55
60
65
70
75
80
85
90
95
100
GAIN (dB)
CIN = 50 pF, CF = 4.5 pF
CIN = 100 pF, CF = 6 pF
CIN = 200 pF, CF = 9 pF
10k 100k 1M 10M
FREQUENCY (Hz)
50
60
70
80
90
100
110
120
130
GAIN (dB)
CIN = 50 pF, CF = 1.5 pF
CIN = 100 pF, CF = 2 pF
CIN = 200 pF, CF = 3 pF
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
Figure 60. Frequency Response for ATI = 470000
Figure 61. Frequency Response for ATI = 47000
HIGH GAIN WIDEBAND TRANSIMPEDANCE AMPLIFIER USING THE LMV792
The LMV792, dual, low noise, wide bandwidth, CMOS input op amp IC can be used for compact, robust and
integrated solutions for sensing and amplifying wide-band signals obtained from sensitive photodiodes. One of
the two op amps available can be used to obtain transimpedance gain while the other can be used for amplifying
the output voltage to further enhance the transimpedance gain. The wide bandwidth of the op amps (17 MHz)
ensures that they are capable of providing high gain for a wide range of frequencies. The low input referred noise
(5.8 nV/√Hz) allows the amplifier to deliver an output with a high SNR (signal to noise ratio). The small VSSOP-
10 footprint saves space on printed circuit boards and allows ease of design in portable products.
The circuit shown in Figure 62, has the first op amp acting as a transimpedance amplifier with a gain of 47000,
while the second stage provides a voltage gain of 10. This provides a total transimpedance gain of 470000 with a
−3 dB bandwidth of about 1.5 MHz, for a total input capacitance of 50 pF. The frequency response for the circuit
is shown in Figure 63
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Product Folder Links: LMV791 LMV792
+
VOUT
+
-
-
RACF
VIN = KI
+
-
IR RADIATION
INTENSITY, I
VOUT RA
K(RA + RB)
I =
IR SENSOR
RB
10k 100k 1M 10M
FREQUENCY (Hz)
60
70
80
90
100
110
120
GAIN (dB)
CIN = 50 pF
CF = 4.5 pF
+
-
+
-
792B
792A
CIN = 50 pF
47 k:
4.5 pF
1 k:
0.1 PF
10 k:
+
-
VOUT
IIN
VOUT
IIN = 470,000
ATI =
LMV791, LMV792
www.ti.com
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
Figure 62. 1.5 MHz Transimpedance Amplifier, with ATI = 470000
Figure 63. 1.5 MHz Transimpedance Amplifier Frequency Response
SENSOR INTERFACES
The low input bias current and low input referred noise of the LMV791 and LMV792 make them ideal for sensor
interfaces. These circuits are required to sense voltages of the order of a few μV, and currents amounting to less
than a nA, and hence the op amp needs to have low voltage noise and low input bias current. Typical
applications include infra-red (IR) thermometry, thermocouple amplifiers and pH electrode buffers. Figure 64 is
an example of a typical circuit used for measuring IR radiation intensity, often used for estimating the
temperature of an object from a distance. The IR sensor generates a voltage proportional to I, which is the
intensity of the IR radiation falling on it. As shown in Figure 64, K is the constant of proportionality relating the
voltage across the IR sensor (VIN) to the radiation intensity, I. The resistances RAand RBare selected to provide
a high gain to amplify this voltage, while CFis added to filter out the high frequency noise.
Figure 64. IR Radiation Sensor
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LMV791 LMV792
LMV791, LMV792
SNOSAG6F –SEPTEMBER 2005–REVISED MARCH 2013
www.ti.com
REVISION HISTORY
Changes from Revision E (March 2013) to Revision F Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 21
22 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMV791 LMV792
PACKAGE OPTION ADDENDUM
www.ti.com 7-Oct-2013
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LMV791MK/NOPB ACTIVE SOT DDC 6 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AS1A
LMV791MKX/NOPB ACTIVE SOT DDC 6 3000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AS1A
LMV792MM/NOPB ACTIVE VSSOP DGS 10 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AX2A
LMV792MMX/NOPB ACTIVE VSSOP DGS 10 3500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM AX2A
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
PACKAGE OPTION ADDENDUM
www.ti.com 7-Oct-2013
Addendum-Page 2
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LMV791MK/NOPB SOT DDC 6 1000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMV791MKX/NOPB SOT DDC 6 3000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMV792MM/NOPB VSSOP DGS 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LMV792MMX/NOPB VSSOP DGS 10 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LMV791MK/NOPB SOT DDC 6 1000 210.0 185.0 35.0
LMV791MKX/NOPB SOT DDC 6 3000 210.0 185.0 35.0
LMV792MM/NOPB VSSOP DGS 10 1000 210.0 185.0 35.0
LMV792MMX/NOPB VSSOP DGS 10 3500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 2
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