a Low Cost, Low Power Instrumentation Amplifier AD620 CONNECTION DIAGRAM FEATURES EASY TO USE Gain Set with One External Resistor (Gain Range 1 to 1000} Wide Power Supply Range (62.3 V to 618 V) Higher Performance than Three Op Amp IA Designs Available in 8-Pin DIP and SOIC Packaging Low Power, 1.3 mA max Supply Current 8-Pin Plastic Mini-DIP (N), Cerdip (Q) and SOIC (R) Packages EXCELLENT DC PERFORMANCE ("A GRADE") 125 mV max, Input Offset Voltage (50 mV max "B" Grade) 1 mV/8C max, Input Offset Drift 2.0 nA max, Input Bias Current 93 dB min Common-Mode Rejection Ratio (G = 10) LOW NOISE 9 nV/Hz, @ 1 kHz, Input Voltage Noise 0.28 mV p-p Noise (0.1 Hz to 10 Hz) 7 +VS +IN 3 6 OUTPUT G -VS 4 AD620 5 REF TOP VIEW The AD620, with its high accuracy of 40 ppm maximum nonlinearity, low offset voltage of 50 V max and offset drift of 0.6 V/C max, is ideal for use in precision data acquisition systems, such as weigh scales and transducer interfaces. Furthermore, the low noise, low input bias current, and low power of the AD620 make it well suited for medical applications such as ECG and noninvasive blood pressure monitors. APPLICATIONS Weigh Scales ECG and Medical Instrumentation Transducer Interface Data Acquisition Systems Industrial Process Controls Battery Powered and Portable Equipment PRODUCT DESCRIPTION The AD620 is a low cost, high accuracy instrumentation amplifier which requires only one external resistor to set gains of 1 to 1000. Furthermore, the AD620 features 8-pin SOIC and DIP packaging that is smaller than discrete designs, and offers lower 30,000 The low input bias current of 1.0 nA max is made possible with the use of Supereta processing in the input stage. The AD620 works well as a preamplifier due to its low input voltage noise of 9 nV/Hz at 1 kHz, 0.28 V p-p in the 0.1 Hz to 10 Hz band, 0.1 pA/Hz input current noise. Also, the AD620 is well suited for multiplexed applications with its settling time of 15 s to 0.01% and its cost is low enough to enable designs with one in amp per channel. 10,000 25,000 3 OP-AMP IN-AMP (3 OP-07s) 20,000 1,000 RTI VOLTAGE NOISE (0.1 -10Hz) - V p-p TOTAL ERROR, PPM OF FULL SCALE 8 R power (only 1.3 mA max supply current), making it a good fit for battery powered, portable (or remote) applications. EXCELLENT AC SPECIFICATIONS 120 kHz Bandwidth (G = 100) 15 ms Settling Time to 0.01% 15,000 AD620A 10,000 RG TYPICAL STANDARD BIPOLAR INPUT IN-AMP 100 10 G = 100 AD620 SUPERETA BIPOLAR INPUT IN-AMP 1 5,000 0 RG 1 -IN 2 0 5 10 15 20 SUPPLY CURRENT - mA 0.1 1k 10k 100k 1M 10M 100M SOURCE RESISTANCE - Three Op Amp IA Designs vs. AD620 Total Voltage Noise vs. Source Resistance Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 REV. D AD620-SPECIFICATIONS (Typical @ +258C, VS = 615 V, and RL = 2 kV, unless otherwise noted) Model Conditions GAIN Gain Range Gain Error2 G=1 G = 10 G = 100 G = 1000 Nonlinearity, G = 1-1000 G = 1-100 Gain vs. Temperature G = 1 + (49.4 k/R G) VOLTAGE OFFSET Input Offset, VOSI Over Temperature Average TC Output Offset, V OSO (Total RTI Error = V OSI + VOSO/G) VS = 5 V to 15 V VS = 5 V to 15 V VS = 5 V to 15 V VS = 15 V VS = 5 V VS = 5 V to 15 V VS = 5 V to 15 V Over Temperature Average TC Offset Referred to the Input vs. Supply (PSR) G=1 G = 10 G = 100 G = 1000 VOUT = 10 V Min AD620A Typ Max 1 VOUT = -10 V to +10 V, RL = 10 k RL = 2 k Gain <10002 Over Temperature Common-Mode Rejection Ratio DC to 60 Hz with I k Source Imbalance G= 1 G = 10 G= 100 G= 1000 OUTPUT Output Swing Over Temperature Over Temperature Short Current Circuit AD620B Typ Max 1 10,000 Min AD620S1 Typ Max 1 Units 10,000 0.03 0.15 0.15 0.40 0.10 0.30 0.30 0.70 0.01 0.10 0.10 0.35 0.02 0.15 0.15 0.50 0.03 0.15 0.15 0.40 0.10 0.30 0.30 0.70 % % % % 10 10 40 95 -50 10 10 40 95 -50 10 10 40 95 -50 ppm ppm ppm/C 30 125 185 1.0 1000 1500 2000 15 15 50 85 0.6 500 750 1000 7.0 30 125 225 1.0 1000 1500 2000 15 V V V/C V V V V/C 0.3 400 5.0 0.1 200 2.5 0.3 400 5.0 VS = 2.3 V to 18 V 80 95 110 110 INPUT CURRENT Input Bias Current Over Temperature Average TC Input Offset Current Over Temperature Average TC INPUT Input Impedance Differential Common-Mode Input Voltage Range 3 Over Temperature 10,000 Min 100 120 140 140 0.5 3.0 0.3 VS = 2.3 V to 5 V VS = 5 V to 18 V 80 100 120 120 2.0 2.5 100 120 140 140 0.5 3.0 0.3 1.0 1.5 80 95 110 110 1.0 1.5 100 120 140 140 0.5 8.0 0.3 0.5 0.75 1.5 1.5 8.0 10i2 10i2 10i2 10i2 10i2 10i2 -VS + 1.9 -VS + 2.1 -VS + 1.9 -VS + 2.1 +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.4 -VS + 1.9 -VS + 2.1 -VS + 1.9 -VS + 2.1 +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.4 -VS + 1.9 -VS + 2.1 -VS + 1.9 -VS + 2.3 dB dB dB dB 2 4 1.0 2.0 +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.4 nA nA pA/C nA nA pA/C GipF GipF V V V V VCM = 0 V to 10 V 73 93 110 110 RL = 10 k, VS = 2.3 V to 5 V VS = 5 V to +18 V -VS + 1.1 -VS + 1.4 -VS + 1.2 -VS + 1.6 90 110 130 130 18 80 100 120 120 +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.5 -2- -VS + 1.1 -VS + 1.4 -VS + 1.2 -VS + 1.6 90 110 130 130 18 73 93 110 110 +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.5 -VS + 1.1 -VS + 1.6 -VS + 1.2 -VS + 2.3 90 110 130 130 18 dB dB dB dB +VS - 1.2 +VS - 1.3 +VS - 1.4 +VS - 1.5 V V V V mA REV. D AD620 Model Conditions DYNAMIC RESPONSE Small Signal -3 dB Bandwidth G=1 G = 10 G = 100 G = 1000 Slew Rate Settling Time to 0.01% 10 V Step G = 1-100 G = 1000 AD620A Typ Max Min 1000 800 120 12 1.2 0.75 AD620B Typ Max Min 1000 800 120 12 1.2 0.75 15 150 Min 0.75 15 150 AD620S1 Typ Max Units 1000 800 120 12 1.2 kHz kHz kHz kHz V/s 15 150 s s NOISE Voltage Noise, 1 kHz Input, Voltage Noise, e ni Output, Voltage Noise, e no RTI, 0.1 Hz to 10 Hz G=1 G = 10 G = 100-1000 Current Noise 0.1 Hz to 10 Hz Total RTI Noise = (e 2 ni ) + (eno / G )2 9 72 f = 1 kHz REFERENCE INPUT RIN IIN Voltage Range Gain to Output VIN+ , VREF = 0 POWER SUPPLY Operating Range 4 Quiescent Current Over Temperature VS = 2.3 V to 18 V 13 100 13 100 9 72 13 100 nV/Hz nV/Hz 3.0 0.55 0.28 100 10 3.0 6.0 0.55 0.8 0.28 0.4 100 10 3.0 6.0 0.55 0.8 0.28 0.4 100 10 V p-p V p-p V p-p fA/Hz pA p-p 20 +50 20 +50 20 +50 k A V +60 -VS + 1.6 +VS -1.6 1 0.0001 VS + 1.6 2.3 2.3 TEMPERATURE RANGE For Specified Performance 18 1.3 1.6 0.9 1.1 -40 to +85 NOTES 1 Does not include effects of external resistor R G. 2 One input grounded. G = 1. 3 This is defined as the same supply range which is used to specify PSR. 4 See Analog Devices military data sheet for 883B tested specifications. Specifications subject to change without notice. REV. D 9 72 -3- +60 +VS - 1.6 1 0.0001 0.9 1.1 -40 to +85 18 1.3 1.6 +60 -VS + 1.6 +VS - 1.6 1 0.0001 2.3 0.9 1.1 18 1.3 1.6 -55 to +125 V mA mA C AD620 ABSOLUTE MAXIMUM RATINGS 1 ORDERING GUIDE Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . . 650 mW Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . VS Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . 25 V Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite Storage Temperature Range (Q) . . . . . . . . . . -65C to +150C Storage Temperature Range (N, R) . . . . . . . . -65C to +125C Operating Temperature Range AD620 (A, B) . . . . . . . . . . . . . . . . . . . . . . -40C to +85C AD620 (S) . . . . . . . . . . . . . . . . . . . . . . . . -55C to +125C Lead Temperature Range (Soldering 10 seconds) . . . . . . . . . . . . . . . . . . . . . . . +300C Model Temperature Range Package Option* AD620AN AD620BN AD620AR AD620BR AD620A Chips AD620SQ/883B -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -55C to +125C N-8 N-8 R-8 R-8 Die Form Q-8 *N = Plastic DIP; Q = Cerdip; R = SOIC. ESD SUSCEPTIBILITY ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 volts, which readily accumulate on the human body and on test equipment, can discharge without detection. Although the AD620 features proprietary ESD protection circuitry, permanent damage may still occur on these devices if they are subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid any performance degradation or loss of functionality. NOTES 1 Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air: 8-Pin Plastic Package: JA = 95C/Watt 8-Pin Cerdip Package: JA = 110C/Watt 8-Pin SOIC Package: JA = 155C/Watt METALIZATION PHOTOGRAPH Dimensions shown in inches and (mm). Contact factory for latest dimensions. -4- REV. D AD620 Typical Characteristics (@ +258C, V = 615 V, R = 2 kV, unless otherwise noted) S L 50 2.0 SAMPLE SIZE = 360 +I B 1.0 INPUT CURRENT - nA PERCENTAGE OF UNITS 1.5 40 30 20 0.5 -I B 0 -0.5 -1.0 10 -1.5 0 -2.0 -80 -40 0 +40 +80 -75 -25 25 75 TEMPERATURE - C INPUT OFFSET VOLTAGE - V Figure 1. Typical Distribution of Input Offset Voltage 125 175 Figure 4. Input Bias Current vs. Temperature 50 2 CHANGE IN OFFSET VOLTAGE - V SAMPLE SIZE = 850 PERCENTAGE OF UNITS 40 30 20 10 0 -1200 -600 0 +600 1.5 1 0.5 0 +1200 0 1 INPUT BIAS CURRENT - pA Figure 2. Typical Distribution of Input Bias Current 2 3 WARM-UP TIME - Minutes 4 5 Figure 5. Change in Input Offset Voltage vs. Warm-Up Time 50 1000 SAMPLE SIZE = 850 Hz GAIN = 1 VOLTAGE NOISE - nV/ PERCENTAGE OF UNITS 40 30 20 10 100 GAIN = 10 10 GAIN = 100, 1,000 GAIN = 1000 BW LIMIT 0 -400 -200 0 +200 1 +400 1 INPUT OFFSET CURRENT - pA Figure 3. Typical Distribution of Input Offset Current REV. D 10 100 1k FREQUENCY - Hz 10k 100k Figure 6. Voltage Noise Spectral Density vs. Frequency, (G = 1-1000) -5- AD620-Typical Characteristics 1000 1s 100mV CURRENT NOISE - fA/ Hz 100 90 100 10 0% 10 1 10 100 FREQUENCY - Hz 1000 Figure 7. Current Noise Spectral Density vs. Frequency Figure 9. 0.1 Hz to 10 Hz Current Noise, 5 pA/Div RTI NOISE - 2.0 V/div TOTAL DRIFT FROM 25C TO 85C, RTI - V 100,000 10,000 FET INPUT IN-AMP 1000 100 AD620A 10 TIME - 1 sec/div 1k 10k 100k 1M SOURCE RESISTANCE - 10M Figure 10. Total Drift vs. Source Resistance Figure 8a. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1) +160 G = 1000 +140 G = 100 RTI NOISE - 0.1V/div +120 CMR - dB G = 10 +100 G=1 +80 +60 +40 +20 0 0.1 TIME - 1 sec/div 1 10 100 1k 10k 100k 1M FREQUENCY - Hz Figure 11. CMR vs. Frequency, RTI, Zero to 1 k Source Imbalance Figure 8b. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1000) -6- REV. D AD620 35 180 G = 10, 100, 1000 30 140 G = 1000 PSR - dB 120 100 G = 100 80 G = 10 60 G=1 20 15 10 5 G=1 40 25 BW LIMIT OUTPUT VOLTAGE - Volts p-p 160 G = 1000 G = 100 20 0.1 0 1 10 100 1k FREQUENCY - Hz 10k 100k 1k 1M 100k 1M FREQUENCY - Hz Figure 15. Large Signal Frequency Response Figure 12. Positive PSR vs. Frequency, RTI (G = 1-1000) +Vs -0.0 160 -0.5 INPUT VOLTAGE LIMIT - Volts (REFERRED TO SUPPLY VOLTAGES) 180 140 120 PSR - dB 10k 100 G = 1000 80 G = 100 60 G = 10 40 -1.0 -1.5 +1.5 +1.0 +0.5 G=1 20 0.1 1 10 100 1k 10k 100k -Vs +0.0 1M 0 5 FREQUENCY - Hz Figure 13. Negative PSR vs. Frequency, RTI (G = 1-1000) +Vs -0.0 OUTPUT VOLTAGE SWING - Volts (REFERRED TO SUPPLY VOLTAGES) GAIN - V/V 100 10 1 -0.5 R L= 10k -1.0 -1.5 R L= 2k +1.5 R L= 2k +1.0 +0.5 R L= 10k -Vs +0.0 1k 10k 100k 1M 0 10M FREQUENCY - Hz 5 10 15 SUPPLY VOLTAGE Volts 20 Figure 17. Output Voltage Swing vs. Supply Voltage, G = 10 Figure 14. Gain vs. Frequency REV. D 20 Figure 16. Input Voltage Range vs. Supply Voltage, G = 1 1000 0.1 100 10 15 SUPPLY VOLTAGE Volts -7- AD620 OUTPUT VOLTAGE SWING - Volts p-p 30 5V VS = 15V G = 10 1mV 100 90 20 10 10 0% 10s 0 0 100 1k LOAD RESISTANCE - 10k Figure 18. Output Voltage Swing vs. Load Resistance 5V Figure 21. Large Signal Response and Settling Time, G = 10 (0.5 mV = 001%) 1mV 20mV 100 100 90 90 10 10 0% 0% 10s 10s Figure 22. Small Signal Response, G = 10, RL = 2 k, CL = 100 pF Figure 19. Large Signal Pulse Response and Settling Time G = 1 (0.5 mV = 0.01%) 5V 20mV 100 100 90 90 10 10 0% 0% 10s 1mV 10s Figure 20. Small Signal Response, G = 1, RL = 2 k, CL = 100 pF Figure 23. Large Signal Response and Settling Time, G = 100 (0.5 mV = 0.01%) -8- REV. D AD620 20 20mV TO 0.01% 100 15 SETTLING TIME - s 90 10 TO 0.1% 10 5 0% 10s 0 0 5 10 15 OUTPUT STEP SIZE - Volts 20 Figure 27. Settling Time vs. Step Size (G = 1) Figure 24. Small Signal Pulse Response, G = 100, RL = 2 k, CL = 100 pF 1000 5V 5mV 100 SETTLING TIME - s 90 100 10 10 0% 100 s 1 1 10 100 1000 GAIN Figure 25. Large Signal Response and Settling Time, G = 1000 (0.5 mV = 0.01%) Figure 28. Settling Time to 0.01% vs. Gain, for a 10 V Step 20mV 10V 100 100 90 90 10 10 0% 2V 0% 50s Figure 29a. Gain Nonlinearity, G = 1, RL = 10 k (10 V = 1 ppm) Figure 26. Small Signal Pulse Response, G = 1000, RL = 2 k, CL = 100 pF REV. D -9- AD620 100V I1 2V 20A VB 20A I2 100 A1 90 A2 C1 10k C2 10k OUTPUT A3 R3 400 - IN R1 10k R2 Q1 R4 400 RG 0% GAIN SENSE REF +IN Q2 10 10k GAIN SENSE -VS Figure 31. Simplified Schematic of AD620 Figure 29b. Gain Nonlinearity, G = 100, RL = 10 k (100 V = 10 ppm) 1mV THEORY OF OPERATION The AD620 is a monolithic instrumentation amplifier based on a modification of the classic three op amp approach. Absolute value trimming allows the user to program gain accurately (to 0.15% at G = 100) with only one resistor. Monolithic construction and laser wafer trimming allow the tight matching and tracking of circuit components, thus insuring the high level of performance inherent in this circuit. 2V 100 90 10 0% Figure 29c. Gain Nonlinearity, G = 1000, RL = 10 k (1 mV = 100 ppm) 10k * INPUT 10V p-p 1k 10T 10k 100k VOUT +VS 11k 1k 2 100 7 1 G=1000 G=1 AD620 G=100 G=10 49.9 499 5.49k 6 The input transistors Q1 and Q2 provide a single differentialpair bipolar input for high precision (Figure 31), yet offer 10x lower Input Bias Current thanks to Supereta processing. Feedback through the Q1-A1-R1 loop and the Q2-A2-R2 loop maintains constant collector current of the input devices Q1, Q2 thereby impressing the input voltage across the external gain setting resistor RG. This creates a differential gain from the inputs to the A1/A2 outputs given by G = (R1 + R2)/RG + 1. The unity-gain subtracter A3 removes any common-mode signal, yielding a single-ended output referred to the REF pin potential. The value of RG also determines the transconductance of the preamp stage. As RG is reduced for larger gains, the transconductance increases asymptotically to that of the input transistors. This has three important advantages: (a) Open-loop gain is boosted for increasing programmed gain, thus reducing gainrelated errors. (b) The gain-bandwidth product (determined by C1, C2 and the preamp transconductance) increases with programmed gain, thus optimizing frequency response. (c) The input voltage noise is reduced to a value of 9 nV/Hz, determined mainly by the collector current and base resistance of the input devices. The internal gain resistors, R1 and R2, are trimmed to an absolute value of 24.7 k, allowing the gain to be programmed accurately with a single external resistor. 5 The gain equation is then 8 4 3 * ALL RESISTORS 1% TOLERANCE G= -VS 49.4 k +1 RG so that Figure 30. Settling Time Test Circuit RG = -10- 49.4 k G -1 REV. D AD620 Make vs. Buy: A Typical Bridge Application Error Budget The AD620 offers improved performance over "homebrew" three op amp IA designs, along with smaller size, less components and 10x lower supply current. In the typical application, shown in Figure 32, a gain of 100 is required to amplify a bridge output of 20 mV full scale over the industrial temperature range of -40C to +85C. The error budget table below shows how to calculate the effect various error sources have on circuit accuracy. Regardless of the system it is being used in, the AD620 provides greater accuracy, and at low power and price. In simple systems, absolute accuracy and drift errors are by far the most significant contributors to error. In more complex systems with an intelligent processor, an auto-gain/auto-zero cycle will remove all absolute accuracy and drift errors leaving only the resolution errors of gain nonlinearity and noise, thus allowing full 14-bit accuracy. Note that for the homebrew circuit, the OP07 specifications for input voltage offset and noise have been multiplied by 2. This is because a three op amp type in amp has two op amps at its inputs, both contributing to the overall input error. +10V 10k* 10k* OP-07D R = 350 R = 350 10k** RG 499 R = 350 AD620A 100** OP-07D 10k** R = 350 REFERENCE OP-07D PRECISION BRIDGE TRANSDUCER AD620A MONOLITHIC INSTRUMENTATION AMPLIFIER, G=100 10k* 10k* "HOMEBREW" IN-AMP, G=100 *0.02% RESISTOR MATCH, 3PPM/C TRACKING **DISCRETE 1% RESISTOR, 100PPM/C TRACKING SUPPLY CURRENT = 15mA MAX SUPPLY CURRENT = 1.3mA MAX Figure 32. Make vs. Buy Table I. Make vs. Buy Error Budget Error Source AD620 Circuit Calculation "Homebrew" Circuit Calculation Error, ppm of Full Scale AD620 Homebrew ABSOLUTE ACCURACY at TA = +25C Input Offset Voltage, V Output Offset Voltage, V Input Offset Current, nA CMR, dB 125 V/20 mV 1000 V/100/20 mV 2 nA x 350 /20 mV 110 dB3.16 ppm, x 5 V/20 mV (150 V x 2)/20 mV ((150 V x 2)/100)/20 mV (6 nA x 350 )/20 mV (0.02% Match x 5 V)/20 mV/100 16,250 14,500 14,118 14,791 10,607 10,150 14,153 10,500 Total Absolute Error 17,558 11,310 100 ppm/C Track x 60C (2.5 V/C x 2 x 60C)/20 mV (2.5 V/C x 2 x 60C)/100/20 mV 13,600 13,000 14,450 16,000 10,607 10,150 Total Drift Error 17,050 16,757 40 ppm (0.38 V p-p x 2)/20 mV 14,140 141,14 10,140 13,127 Total Resolution Error 14,154 101,67 Grand Total Error 14,662 28,134 DRIFT TO +85C Gain Drift, ppm/C Input Offset Voltage Drift, V/C Output Offset Voltage Drift, V/C (50 ppm + 10 ppm) x 60C 1 V/C x 60C/20 mV 15 V/C x 60C/100/20 mV RESOLUTION Gain Nonlinearity, ppm of Full Scale 40 ppm Typ 0.1 Hz-10 Hz Voltage Noise, V p-p 0.28 V p-p/20 mV G = 100, VS = 15 V. (All errors are min/max and referred to input.) REV. D -11- AD620 +5V 3k 20k 7 3 3k REF 8 3k 3k AD620B G=100 499 6 IN 5 1 2 10k ADC 4 AD705 DIGITAL DATA OUTPUT AGND 20k 1.3mA MAX 1.7mA 0.6mA MAX 0.10mA Figure 33. A Pressure Monitor Circuit which Operates on a +5 V Single Supply Pressure Measurement Medical ECG Although useful in many bridge applications such as weigh scales, the AD620 is especially suited for higher resistance pressure sensors powered at lower voltages where small size and low power become more significant. The low current noise of the AD620 allows its use in ECG monitors (Figure 34) where high source resistances of 1 M or higher are not uncommon. The AD620's low power, low supply voltage requirements, and space-saving 8-pin mini-DIP and SOIC package offerings make it an excellent choice for battery powered data recorders. Figure 33 shows a 3 k pressure transducer bridge powered from +5 V. In such a circuit, the bridge consumes only 1.7 mA. Adding the AD620 and a buffered voltage divider allows the signal to be conditioned for only 3.8 mA of total supply current. Small size and low cost make the AD620 especially attractive for voltage output pressure transducers. Since it delivers low noise and drift, it will also serve applications such as diagnostic noninvasive blood pressure measurement. Furthermore, the low bias currents and low current noise coupled with the low voltage noise of the AD620 improve the dynamic range for better performance. The value of capacitor C1 is chosen to maintain stability of the right leg drive loop. Proper safeguards, such as isolation, must be added to this circuit to protect the patient from possible harm. +3V PATIENT/CIRCUIT PROTECTION/ISOLATION 7 3 C1 R1 10k R4 1M R3 24.9k R2 24.9k 8 RG 8.25k AD620A 6 G=7 1 2 5 4 0.03Hz HIGH PASS FILTER G = 143 OUTPUT 1V/mV OUTPUT AMPLIFIER AD705J -3V Figure 34. A Medical ECG Monitor Circuit -12- REV. D AD620 Precision V-I Converter INPUT AND OUTPUT OFFSET VOLTAGE The AD620 along with another op amp and two resistors make a precision current source (Figure 35). The op amp buffers the reference terminal to maintain good CMR. The output voltage VX of the AD620 appears across R1 which converts it to a current. This current less only the input bias current of the op amp then flows out to the load. The low errors of the AD620 are attributed to two sources, input and output errors. The output error is divided by G when referred to the input. In practice, the input errors dominate at high gains and the output errors dominate at low gains. The total VOS for a given gain is calculated as: Total Error RTI = input error + (output error/G) Total Error RTO = (input error x G) + output error +Vs VIN+ REFERENCE TERMINAL 7 3 8 AD620 RG VIN- R1 5 1 2 The reference terminal potential defines the zero output voltage, and is especially useful when the load does not share a precise ground with the rest of the system. It provides a direct means of injecting a precise offset to the output, with an allowable range of 2 V within the supply voltages. Parasitic resistance should be kept to a minimum for optimum CMR. + Vx 6 4 I L -Vs AD705 I L= Vx R1 = INPUT PROTECTION [(VIN+ ) - (VIN- )] G R1 LOAD Figure 35. Precision Voltage-to-Current Converter (Operates on 1.8 mA, 3 V) GAIN SELECTION The AD620's gain is resistor programmed by RG: or more precisely, by whatever impedance appears between Pins 1 and 8. The AD620 is designed to offer accurate gains using 0.1%-1% resistors. Table II shows required values of RG for various gains. Note that for G = 1, the RG pins are unconnected (RG = ). For any arbitrary gain RG can be calculated by using the formula: RG = 49.4 k G -1 To minimize gain error avoid high parasitic resistance in series with RG, and to minimize gain drift RG should have a low TC-- less than 10 ppm/C for the best performance. The AD620 features 400 of series thin film resistance at its inputs, and will safely withstand input overloads of up to 15 V or 60 mA for several hours. This is true for all gains, and power on and off, which is particularly important since the signal source and amplifier may be powered separately. For longer time periods, the current should not exceed 6 mA (IIN VIN/400 ). For input overloads beyond the supplies, clamping the inputs to the supplies (using a low leakage diode such as an FD333) will reduce the required resistance, yielding lower noise. RF INTERFERENCE All instrumentation amplifiers can rectify out of band signals, and when amplifying small signals, these rectified voltages act as small dc offset errors. The AD620 allows direct access to the input transistor bases and emitters enabling the user to apply some first order filtering to unwanted RF signals (Figure 36), where RC < 1/(2 f) and where f the bandwidth of the AD620; C 150 pF. Matching the extraneous capacitance at Pins 1 and 8, and Pins 2 and 3 helps to maintain high CMR. RG Table II. Required Values of Gain Resistors 1% Std Table Value of RG, Calculated Gain 0.1% Std Table Value of RG, Calculated Gain 49.9 k 12.4 k 5.49 k 1.990 4.984 9.998 49.3 k 12.4 k 5.49 k 2.002 4.984 9.998 2.61 k 1.00 k 499 19.93 50.40 100.0 2.61 k 1.01 k 499 19.93 49.91 100.0 249 100 49.9 199.4 495.0 991.0 249 98.8 49.3 199.4 501.0 1,003 REV. D 1 8 2 7 3 6 4 5 C R -IN R +IN C Figure 36. Circuit to Attenuate RF Interference -13- AD620 problems by simply tying the REF pin to the appropriate "local ground." COMMON-MODE REJECTION Instrumentation amplifiers like the AD620 offer high CMR which is a measure of the change in output voltage when both inputs are changed by equal amounts. These specifications are usually given for a full-range input voltage change and a specified source imbalance. For optimal CMR the reference terminal should be tied to a low impedance point, and differences in capacitance and resistance should be kept to a minimum between the two inputs. In many applications shielded cables are used to minimize noise, and for best CMR over frequency the shield should be properly driven. Figures 37 and 38 show active data guards which are configured to improve ac common-mode rejections by "bootstrapping" the capacitances of input cable shields, thus minimizing the capacitance mismatch between the inputs. In order to isolate low level analog signals from a noisy digital environment, many data-acquisition components have separate analog and digital ground pins (Figure 39). It would be convenient to use a single ground line, however, current through ground wires and PC runs of the circuit card can cause hundreds of millivolts of error. Therefore, separate ground returns should be provided to minimize the current flow from the sensitive points to the system ground. These ground returns must be tied together at some point, usually best at the ADC package as shown. +VS - INPUT 0.1F 0.1F 1F 1F 7 2 6 -VS 3 5 6 6 + 4 11 4 AD620 VOUT AD620 RG 1F 7 1 100 AD585 S/H 7 9 11 15 1 AD574A ADC DIGITAL DATA OUTPUT 5 8 Figure 39. Basic Grounding Practice REFERENCE 4 + INPUT C +5V 2 AD648 100 DIGITAL P.S. ANALOG P.S. +15V C -15V 3 GROUND RETURNS FOR INPUT BIAS CURRENTS -V S Input bias currents are those currents necessary to bias the input transistors of an amplifier. There must be a direct return path Figure 37. Differential Shield Driver +V S - INPUT 2 7 100 AD548 + INPUT RG 2 RG 2 1 VOUT AD620 5 8 3 6 4 REFERENCE -V S Figure 38. Common-Mode Shield Driver GROUNDING Since the AD620 output voltage is developed with respect to the potential on the reference terminal, it can solve many grounding -14- REV. D AD620 for these currents; therefore when amplifying "floating" input +VS - INPUT 2 1 7 +VS VOUT AD620 RG sources such as transformers, or ac-coupled sources, there must be a dc path from each input to ground as shown in Figure 40. Refer to the Instrumentation Amplifier Application Guide (free from Analog Devices) for more information regarding in amp applications. 6 - INPUT 2 5 8 + INPUT 3 1 LOAD 4 VOUT AD620 RG REFERENCE 7 -VS 6 5 8 TO POWER SUPPLY GROUND 3 + INPUT LOAD 4 REFERENCE -VS Figure 40a. Ground Returns for Bias Currents with Transformer Coupled Inputs TO POWER SUPPLY GROUND Figure 40b. Ground Returns for Bias Currents with Thermocouple Inputs +VS - INPUT 2 1 7 VOUT AD620 RG 6 5 8 + INPUT 100k 100k LOAD 4 3 REFERENCE -VS TO POWER SUPPLY GROUND Figure 40c. Ground Returns for Bias Currents with AC Coupled Inputs REV. D -15- AD620 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). Plastic DIP (N-8) Package 5 0.25 (6.35) PIN 1 1 C15499b-12-4/93 8 0.31 (7.87) 4 0.30 (7.62) REF 0.39 (9.91) MAX 0.0350.01 (0.890.25) 0.1650.01 (4.190.25) 0.0110.003 (0.280.08) 0.180.03 (4.570.76) 0.125 (3.18) MIN 0.0180.003 (0.460.08) 0.10 (2.54) BSC 0.033 (0.84) NOM 15 0 SEATING PLANE Cerdip (Q-8) Package 0.005 (0.13) MIN 0.055 (1.40) MAX 8 5 0.310 (7.87) 0.220 (5.59) PIN 1 1 4 0.320 (8.13) 0.290 (7.37) 0.405 (10.29) MAX 0.060 (1.52) 0.015 (0.38) 0.200 (5.08) MAX 0.150 (3.81) MIN 0.200 (5.08) 0.125 (3.18) 0.015 (0.38) 0.008 (0.20) 15 0.023 (0.58) 0.014 (0.36) 0.100 0.070 (1.78) (2.54) 0.030 (0.76) BSC 0 SEATING PLANE SOIC (R-8) Package 0.150 (3.81) 8 5 0.244 (6.20) 0.228 (5.79) 1 4 0.197 (5.01) 0.189 (4.80) 0.102 (2.59) 0.094 (2.39) 0.010 (0.25) 0.004 (0.10) 0.050 (1.27) BSC 0.019 (0.48) 0.014 (0.36) 0.020 (0.051) x 45 CHAMF 0.190 (4.82) 0.170 (4.32) 8 0 0.090 (2.29) PRINTED IN U.S.A. PIN 1 0.157 (3.99) 0.150 (3.81) 10 0 0.098 (0.2482) 0.075 (0.1905) 0.030 (0.76) 0.018 (0.46) All brand or product names mentioned are trademarks or registered trademarks of their respective holders. -16- REV. D