1
LT1167
Single Resistor Gain
Programmable, Precision
Instrumentation Amplifier
Gain Nonlinearity
The LT
®
1167 is a low power, precision instrumentation
amplifier that requires only one external resistor to set gains
of 1 to 10,000. The low voltage noise of 7.5nV/√Hz (at 1kHz)
is not compromised by low power dissipation (0.9mA typical
for ±2.3V to ±15V supplies).
The high accuracy of 10ppm maximum nonlinearity and
0.08% max gain error (G = 10) is not degraded even for load
resistors as low as 2k (previous monolithic instrumentation
amps used 10k for their nonlinearity specifications). The
LT1167 is laser trimmed for very low input offset voltage
(40µV max), drift (0.3µV/°C), high CMRR (90dB, G = 1) and
PSRR (105dB, G = 1). Low input bias currents of 350pA max
are achieved with the use of superbeta processing. The
output can handle capacitive loads up to 1000pF in any gain
configuration while the inputs are ESD protected up to 13kV
(human body). The LT1167 with two external 5k resistors
passes the IEC 1000-4-2 level 4 specification.
The LT1167, offered in 8-pin PDIP and SO packages, requires
significantly less PC board area than discrete multi op amp
and resistor designs. These advantages make the LT1167 the
most cost effective solution for precision instrumentation
amplifier applications.
■
Single Gain Set Resistor: G = 1 to 10,000
■
Gain Error: G = 10, 0.08% Max
■
Gain Nonlinearity: G = 10, 10ppm Max
■
Input Offset Voltage: G = 10, 60µV Max
■
Input Offset Voltage Drift: 0.3
µ
V/
°
C Max
■
Input Bias Current: 350pA Max
■
PSRR at G = 1: 105dB Min
■
CMRR at G = 1: 90dB Min
■
Supply Current: 1.3mA Max
■
Wide Supply Range: ±2.3V to ±18V
■
1kHz Voltage Noise: 7.5nV/√Hz
■
0.1Hz to 10Hz Noise: 0.28µV
P-P
■
Available in 8-Pin PDIP and SO Packages
■
Meets IEC 1000-4-2 Level 4 ESD Tests with
Two External 5k Resistors
FEATURES
DESCRIPTION
U
, LTC and LT are registered trademarks of Linear Technology Corporation.
■
Bridge Amplifiers
■
Strain Gauge Amplifiers
■
Thermocouple Amplifiers
■
Differential to Single-Ended Converters
■
Medical Instrumentation
APPLICATIONS
U
–
+
+
–
–
+
2
1
1
1
1
2
R5
392k
R4
50k
R3
50k
R8
100k
R6
1k
LT1634CCZ-1.25
8
4
1/2
LT1490
3
R
SET
0.2% ACCURACY AT 25°C
1.2% ACCURACY AT 0°C TO 60°C
V
S
= 8V TO 30V
5k
5k
5k
5k
V
S
5
4
3
2
–
+
7
1/2
LT1490
5
6
2
8
LUCAS NOVA SENOR
NPC-1220-015-A-3L
7
V
S
6
1167 TA01
5TO
4-DIGIT
DVM
4
R2
12Ω
LT1167
G = 60
R1
825Ω
3
6
R7
50k VOLTS
2.800
3.000
3.200
INCHES Hg
28.00
30.00
32.00
TYPICAL APPLICATION
U
Single Supply Barometer
G = 1000
R
L
= 1k
V
OUT
= ±10V
1167 TA02
OUTPUT VOLTAGE (2V/DIV)
NONLINEARITY (100ppm/DIV)
2
LT1167
ABSOLUTE MAXIMUM RATINGS
W
WW
U
ORDER PART
NUMBER
PACKAGE/ORDER INFORMATION
W
UU
S8 PART MARKING
LT1167ACN8
LT1167ACS8
LT1167AIN8
LT1167AIS8
LT1167CN8
LT1167CS8
LT1167IN8
LT1167IS8
1167A
1167AI
Consult factory for Military grade parts.
1
2
3
4
8
7
6
5
TOP VIEW
R
G
–IN
+IN
–V
S
R
G
+V
S
OUTPUT
REF
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
+
–
T
JMAX
= 150°C, θ
JA
= 130°C/ W (N8)
T
JMAX
= 150°C, θ
JA
= 190°C/ W (S8)
1167
1167I
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, TA = 25°C, RL = 2k, unless otherwise noted.
LT1167AC/LT1167AI LT1167C/LT1167I
SYMBOL PARAMETER CONDITIONS (Note 7) MIN TYP MAX MIN TYP MAX UNITS
G Gain Range G = 1 + (49.4k/R
G
) 1 10k 1 10k
Gain Error G = 1 0.008 0.02 0.015 0.03 %
G = 10 (Note 2) 0.010 0.08 0.020 0.10 %
G = 100 (Note 2) 0.025 0.08 0.030 0.10 %
G = 1000 (Note 2) 0.040 0.10 0.040 0.10 %
Gain Nonlinearity (Note 5) V
O
= ±10V, G = 1 1 6 1.5 10 ppm
V
O
= ±10V, G = 10 and 100 2 10 3 15 ppm
V
O
= ±10V, G = 1000 15 40 20 60 ppm
V
O
= ±10V, G = 1, R
L
= 600 5 12 6 15 ppm
V
O
= ±10V, G = 10 and 100, 6 15 7 20 ppm
R
L
= 600
V
O
= ±10V, G = 1000, 20 65 25 80 ppm
R
L
= 600
V
OST
Total Input Referred Offset Voltage V
OST
= V
OSI
+ V
OSO
/G
V
OSI
Input Offset Voltage G = 1000, V
S
= ±5V to ±15V 15 40 20 60 µV
V
OSO
Output Offset Voltage G = 1, V
S
= ±5V to ±15V 40 200 50 300 µV
I
OS
Input Offset Current 90 320 100 450 pA
I
B
Input Bias Current 50 350 80 500 pA
e
n
Input Noise Voltage, RTI 0.1Hz to 10Hz, G = 1 2.00 2.00 µV
P-P
0.1Hz to 10Hz, G = 10 0.50 0.50 µV
P-P
0.1Hz to 10Hz, G = 100 and 1000 0.28 0.28 µV
P-P
Total RTI Noise =
√
e
ni2
+ (e
no
/G)
2
e
ni
Input Noise Voltage Density, RTI f
O
= 1kHz 7.5 12 7.5 12 nV/√Hz
e
no
Output Noise Voltage Density, RTI f
O
= 1kHz (Note 3) 67 90 67 90 nV/√Hz
(Note 1)
Supply Voltage ...................................................... ±20V
Differential Input Voltage (Within the
Supply Voltage) ..................................................... ±40V
Input Voltage (Equal to Supply Voltage) ................ ±20V
Input Current (Note 3) ........................................ ±20mA
Output Short-Circuit Duration .......................... Indefinite
Operating Temperature Range ................ –40°C to 85°C
Specified Temperature Range
LT1167AC/LT1167C (Note 4) .................. 0°C to 70°C
LT1167AI/LT1167I ............................. –40°C to 85°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)..................300°C
3
LT1167
LT1167AC/LT1167AI LT1167C/LT1167I
SYMBOL PARAMETER CONDITIONS (Note 7) MIN TYP MAX MIN TYP MAX UNITS
i
n
Input Noise Current f
O
= 0.1Hz to 10Hz 10 10 pA
P-P
Input Noise Current Density f
O
= 10Hz 124 124 fA/√Hz
R
IN
Input Resistance V
IN
= ±10V 200 1000 200 1000 GΩ
C
IN(DIFF)
Differential Input Capacitance f
O
= 100kHz 1.6 1.6 pF
C
IN(CM)
Common Mode Input f
O
= 100kHz 1.6 1.6 pF
Capacitance
V
CM
Input Voltage Range G = 1, Other Input Grounded
V
S
= ±2.3V to ±5V –V
S
+ 1.9 +V
S
– 1.2 –V
S
+ 1.9 +V
S
– 1.2 V
V
S
= ±5V to ±18V –V
S
+ 1.9 +V
S
– 1.4 –V
S
+ 1.9 +V
S
– 1.4 V
CMRR Common Mode 1k Source Imbalance,
Rejection Ratio V
CM
= 0V to ±10V
G = 1 90 95 85 95 dB
G = 10 106 115 100 115 dB
G = 100 120 125 110 125 dB
G = 1000 126 140 120 140 dB
PSRR Power Supply V
S
= ±2.3 to ±18V
Rejection Ratio G = 1 105 120 100 120 dB
G = 10 125 135 120 135 dB
G = 100 131 140 126 140 dB
G = 1000 135 150 130 150 dB
I
S
Supply Current V
S
= ±2.3V to ±18V 0.9 1.3 0.9 1.3 mA
V
OUT
Output Voltage Swing R
L
= 10k
V
S
= ±2.3V to ±5V –V
S
+ 1.1 +V
S
– 1.2 –V
S
+ 1.1 +V
S
– 1.2 V
V
S
= ±5V to ±18V –V
S
+ 1.2 +V
S
– 1.3 –V
S
+ 1.2 +V
S
– 1.3 V
I
OUT
Output Current 20 27 20 27 mA
BW Bandwidth G = 1 1000 1000 kHz
G = 10 800 800 kHz
G = 100 120 120 kHz
G = 1000 12 12 kHz
SR Slew Rate G = 1, V
OUT
= ±10V 0.75 1.2 0.75 1.2 V/µs
Settling Time to 0.01% 10V Step
G = 1 to 100 14 14 µs
G = 1000 130 130 µs
R
REFIN
Reference Input Resistance 20 20 kΩ
I
REFIN
Reference Input Current V
REF
= 0V 50 50 µA
V
REF
Reference Voltage Range –V
S
+ 1.6 +V
S
– 1.6 –V
S
+ 1.6 +V
S
– 1.6 V
A
VREF
Reference Gain to Output 1 ± 0.0001 1 ± 0.0001
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, TA = 25°C, RL = 2k, unless otherwise noted.
4
LT1167
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, 0°C ≤ TA ≤ 70°C, RL = 2k, unless otherwise noted.
LT1167AC LT1167C
SYMBOL PARAMETER CONDITIONS (Note 7) MIN TYP MAX MIN TYP MAX UNITS
Gain Error G = 1 ●0.01 0.03 0.012 0.04 %
G = 10 (Note 2) ●0.08 0.30 0.100 0.33 %
G = 100 (Note 2) ●0.09 0.30 0.120 0.33 %
G = 1000 (Note 2) ●0.14 0.33 0.140 0.35 %
Gain Nonlinearity V
OUT
= ±10V, G = 1 ●1.5 10 2 15 ppm
V
OUT
= ±10V, G = 10 and 100 ●3 15 4 20 ppm
V
OUT
= ±10V, G = 1000 ●20 60 25 80 ppm
G/T Gain vs Temperature G < 1000 (Note 2) ●20 50 20 50 ppm/°C
V
OST
Total Input Referred V
OST
= V
OSI
+ V
OSO
/G
Offset Voltage
V
OSI
Input Offset Voltage V
S
= ±5V to ±15V ●18 60 23 80 µV
V
OSIH
Input Offset Voltage Hysteresis (Notes 3, 6) 3.0 3.0 µV
V
OSO
Output Offset Voltage V
S
= ±5V to ±15V ●60 380 70 500 µV
V
OSOH
Output Offset Voltage Hysteresis (Notes 3, 6) 30 30 µV
V
OSI
/T Input Offset Drift (RTI) (Note 3) ●0.05 0.3 0.06 0.4 µV/°C
V
OSO
/T Output Offset Drift (Note 3) ●0.7 3 0.8 4 µV/°C
I
OS
Input Offset Current ●100 400 120 550 pA
I
OS
/T Input Offset Current Drift ●0.3 0.4 pA/°C
I
B
Input Bias Current ●75 450 105 600 pA
I
B
/T Input Bias Current Drift ●0.4 0.4 pA/°C
V
CM
Input Voltage Range G = 1, Other Input Grounded
V
S
= ±2.3V to ±5V ●–V
S
+2.1 +V
S
–1.3 –V
S
+2.1 +V
S
–1.3 V
V
S
= ±5V to ±18V ●–V
S
+2.1 +V
S
–1.4 –V
S
+2.1 +V
S
–1.4 V
CMRR Common Mode 1k Source Imbalance,
Rejection Ratio V
CM
= 0V to ±10V
G = 1 ●88 92 83 92 dB
G = 10 ●100 110 97 110 dB
G = 100 ●115 120 113 120 dB
G = 1000 ●117 135 114 135 dB
PSRR Power Supply Rejection Ratio V
S
= ±2.3V to ±18V
G = 1 ●103 115 98 115 dB
G = 10 ●123 130 118 130 dB
G = 100 ●127 135 124 135 dB
G = 1000 ●129 145 126 145 dB
I
S
Supply Current V
S
= ±2.3V to ±18V ●1.0 1.5 1.0 1.5 mA
V
OUT
Output Voltage Swing R
L
= 10k
V
S
= ±2.3V to ±5V ●–V
S
+1.4 +V
S
–1.3 –V
S
+1.4 +V
S
–1.3 V
V
S
= ±5V to ±18V ●–V
S
+1.6 +V
S
–1.5 –V
S
+1.6 +V
S
–1.5 V
I
OUT
Output Current ●16 21 16 21 mA
SR Slew Rate G = 1, V
OUT
= ±10V ●0.65 1.1 0.65 1.1 V/µs
V
REF
REF Voltage Range (Note 3) ●–V
S
+1.6 +V
S
–1.6 –V
S
+1.6 +V
S
–1.6 V
5
LT1167
ELECTRICAL CHARACTERISTICS
VS = ±15V, VCM = 0V, –40°C ≤ TA ≤ 85°C, RL = 2k, unless otherwise noted. (Note 4)
LT1167AI LT1167I
SYMBOL PARAMETER CONDITIONS (Note 7) MIN TYP MAX MIN TYP MAX UNITS
Gain Error G = 1 ●0.014 0.04 0.015 0.05 %
G = 10 (Note 2) ●0.130 0.40 0.140 0.42 %
G = 100 (Note 2) ●0.140 0.40 0.150 0.42 %
G = 1000 (Note 2) ●0.160 0.40 0.180 0.45 %
G
N
Gain Nonlinearity (Notes 2, 4) V
O
= ±10V, G = 1 ●2 15 3 20 ppm
V
O
= ±10V, G = 10 and 100 ●5 20 6 30 ppm
V
O
= ±10V, G = 1000 ●26 70 30 100 ppm
G/T Gain vs Temperature G < 1000 (Note 2) ●20 50 20 50 ppm/°C
V
OST
Total Input Referred V
OST
= V
OSI
+ V
OSO
/G
Offset Voltage
V
OSI
Input Offset Voltage ●20 75 25 100 µV
V
OSIH
Input Offset Voltage Hysteresis (Notes 3, 6) 3.0 3.0 µV
V
OSO
Output Offset Voltage ●180 500 200 600 µV
V
OSOH
Output Offset Voltage Hysteresis (Notes 3, 6) 30 30 µV
V
OSI
/T Input Offset Drift (RTI) (Note 3) ●0.05 0.3 0.06 0.4 µV/°C
V
OSO
/T Output Offset Drift (Note 3) ●0.8 5 1 6 µV/°C
I
OS
Input Offset Current ●110 550 120 700 pA
I
OS
/T Input Offset Current Drift ●0.3 0.3 pA/°C
I
B
Input Bias Current ●180 600 220 800 pA
I
B
/T Input Bias Current Drift ●0.5 0.6 pA/°C
V
CM
Input Voltage Range V
S
= ±2.3V to ±5V ●–V
S
+ 2.1 +V
S
– 1.3 –V
S
+ 2.1 +V
S
– 1.3 V
V
S
= ±5V to ±18V ●–V
S
+ 2.1 +V
S
– 1.4 –V
S
+ 2.1 +V
S
– 1.4 V
CMRR Common Mode Rejection Ratio 1k Source Imbalance,
V
CM
= 0V to ±10V
G = 1 ●86 90 81 90 dB
G = 10 ●98 105 95 105 dB
G = 100 ●114 118 112 118 dB
G = 1000 ●116 133 112 133 dB
PSRR Power Supply Rejection Ratio V
S
= ±2.3V to ±18V
G = 1 ●100 112 95 112 dB
G = 10 ●120 125 115 125 dB
G = 100 ●125 132 120 132 dB
G = 1000 ●128 140 125 140 dB
I
S
Supply Current ●1.1 1.6 1.1 1.6 mA
V
OUT
Output Voltage Swing V
S
= ±2.3V to ±5V ●–V
S
+ 1.4 +V
S
– 1.3 –V
S
+ 1.4 +V
S
– 1.3 V
V
S
= ±5V to ±18V ●–V
S
+ 1.6 +V
S
– 1.5 –V
S
+ 1.6 +V
S
– 1.5 V
I
OUT
Output Current ●15 20 15 20 mA
SR Slew Rate G = 1, V
OUT
= ±10V ●0.55 0.95 0.55 0.95 V/µs
V
REF
REF Voltage Range (Note 3) ●–V
S
+ 1.6 +V
S
– 1.6 –V
S
+ 1.6 +V
S
– 1.6 V
The ● denotes specifications that apply over the full specified
temperature range.
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be imparied.
Note 2: Does not include the effect of the external gain resistor R
G
.
Note 3: This parameter is not 100% tested.
Note 4: The LT1167AC/LT1167C are designed, characterized and expected
to meet the industrial temperature limits, but are not tested at –40°C and
85°C. I-grade parts are guaranteed.
Note 5: This parameter is measured in a high speed automatic tester that
does not measure the thermal effects with longer time constants. The
magnitude of these thermal effects are dependent on the package used,
heat sinking and air flow conditions.
Note 6: Hysteresis in offset voltage is created by package stress that
differs depending on whether the IC was previously at a higher or lower
temperature. Offset voltage hysteresis is always measured at 25°C, but
the IC is cycled to 85°C I-grade (or 70°C C-grade) or –40°C I-grade
(0°C C-grade) before successive measurement. 60% of the parts will
pass the typical limit on the data sheet.
Note 7: Typical parameters are defined as the 60% of the yield parameter
distribution.
6
LT1167
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Gain Nonlinearity, G = 1 Gain Nonlinearity, G = 100
Gain Nonlinearity, G = 10
NONLINEARITY (1ppm/DIV)
OUTPUT VOLTAGE (2V/DIV)
G = 1
R
L
= 2k
V
OUT
= ±10V
1167 G01
NONLINEARITY (10ppm/DIV)
OUTPUT VOLTAGE (2V/DIV)
G = 10
R
L
= 2k
V
OUT
= ±10V
1167 G02
NONLINEARITY (10ppm/DIV)
OUTPUT VOLTAGE (2V/DIV)
G = 100
R
L
= 2k
V
OUT
= ±10V
1167 G03
Gain Nonlinearity, G = 1000
NONLINEARITY (100ppm/DIV)
OUTPUT VOLTAGE (2V/DIV)
G = 1000
R
L
= 2k
V
OUT
= ±10V
1167 G04
Gain Error vs Temperature
TEMPERATURE (°C)
–50
GAIN ERROR (%)
–0.20
–0.10
–0.05
0
50
0.20
1167 G06
–0.15
0
–25 75
G = 1
25 100
0.05
0.10
0.15
V
S
= ±15V
V
OUT
= ±10V
R
L
= 2k
*DOES NOT INCLUDE
TEMPERATURE EFFECTS
OF R
G
G = 10*
G = 1000*
G = 100*
Gain Nonlinearity vs Temperature
TEMPERATURE (°C)
–50
NONLINEARITY (ppm)
70
25
1167 G05
40
20
–25 0 50
10
0
80
60
50
30
75 100 150
G = 1000
G = 100
G = 1, 10
V
S
= ±15V
V
OUT
= –10V TO 10V
R
L
= 2k
Distribution of Input
Offset Voltage, TA = –40°CDistribution of Input
Offset Voltage, TA = 25°C
Distribution of Input
Offset Voltage, TA = 85°C
INPUT OFFSET VOLTAGE (µV)
–80
PERCENT OF UNITS (%)
20
30
1167 G40
10
0–60 –40 –20 20 40 60
0
40
15
25
5
35
V
S
= ±15V
G = 1000 137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
INPUT OFFSET VOLTAGE (µV)
–60 –40 –20 0 20 40 60
PERCENT OF UNITS (%)
20
25
30
1167 G41
15
10
0
5
V
S
= ±15V
G = 1000 137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
INPUT OFFSET VOLTAGE (µV)
–80
PERCENT OF UNITS (%)
20
30
1167 G42
10
0–60 –40 –20 20 40 60
0
40
15
25
5
35
V
S
= ±15V
G = 1000 137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
7
LT1167
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Distribution of Output
Offset Voltage, TA = 85°C
Warm-Up Drift
Distribution of Output Offset
Voltage Drift
OUTPUT OFFSET VOLTAGE (µV)
–200 –150 –100 –50 0 50 100 150 200
PERCENT OF UNITS (%)
20
30
1167 G44
10
0
15
25
5
V
S
= ±15V
G = 1
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
Distribution of Output
Offset Voltage, TA = 25°C
Distribution of Output
Offset Voltage, TA = –40°C
OUTPUT OFFSET VOLTAGE (µV)
–400 –300 –200 –100 0 100 200 300 400
PERCENT OF UNITS (%)
20
30
1167 G43
10
0
40
15
25
5
35
V
S
= ±15V
G = 1
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
OUTPUT OFFSET VOLTAGE (µV)
–400 –300 –200 –100 0 100 200 300 400
PERCENT OF UNITS (%)
20
30
1167 G45
10
0
40
15
25
5
35
V
S
= ±15V
G = 1
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
TIME AFTER POWER ON (MINUTES)
0
10
12 S8
N8
14
34
1167 G09
8
6
12 5
4
2
0
CHANGE IN OFFSET VOLTAGE (µV)
V
S
= ±15V
T
A
= 25°C
G = 1
OUTPUT OFFSET VOLTAGE (µV)
0
PERCENT OF UNITS (%)
5
10
15
20
40
012345–1–2–3–4–5
1167 G47
30
35
25
V
S
= ±15V
T
A
= –40°C TO 85°C
G = 1
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
INPUT OFFSET VOLTAGE (µV)
–0.4
0
PERCENT OF UNITS (%)
5
10
15
20
30
–0.3 –0.2 –0.1 0
1167 G46
0.1 0.2 0.3
25
V
S
= ±15V
T
A
= –40°C TO 85°C
G = 1000
137 N8 (2 LOTS)
165 S8 (3 LOTS)
302 TOTAL PARTS
Distribution of Input Offset
Voltage Drift
INPUT BIAS CURRENT (pA)
–100
PERCENT OF UNITS (%)
30
40
50
60
1167 G10
20
10
0–60 –20 20 100
V
S
= ±15V
T
A
= 25°C270 S8
122 N8
392 TOTAL PARTS
Input Bias Current
INPUT OFFSET CURRENT (pA)
–100
PERCENT OF UNITS (%)
30
40
50
60
1167 G11
20
10
0–60 –20 20 100
V
S
= ±15V
T
A
= 25°C270 S8
122 N8
392 TOTAL PARTS
Input Offset Current
Input Bias and Offset Current
vs Temperature
TEMPERATURE (°C)
–50–75
–500
INPUT BIAS AND OFFSET CURRENT (pA)
–400
–200
–100
0
500
200
050 75
1167 G12
–300
300
400
100
–25 25 100
I
OS
125
V
S
= ±15V
V
CM
= 0V
I
B
8
LT1167
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Supply Current vs Supply Voltage
SUPPLY VOLTAGE (±V)
0
SUPPLY CURRENT (mA)
1.00
1.25
85°C
25°C
–40°C
20
1167 G18
0.75
0.50 510 15
1.50
COMMON MODE INPUT VOLTAGE (V)
–15
INPUT BIAS CURRENT (pA)
100
300
500
9
1167 G13
–100
–300
0
200
400
–200
–400
–500 –9 –3 3
–12 12
–6 0615
–40°C
85°C
0°C
70°C
25°C
Input Bias Current
vs Common Mode Input Voltage
FREQUENCY (Hz)
0.1
NEGATIVE POWER SUPPLY REJECTION RATIO (dB)
60
80
100
100 10k
1167 G15
40
20
0110 1k
120
140
160
100k
G = 1000
G = 100
G = 10
G = 1
V
+
= 15V
T
A
= 25°C
Negative Power Supply Rejection
Ratio vs Frequency
FREQUENCY (Hz)
0.1
COMMON MODE REJECTION RATIO (dB)
60
80
100
100 10k
1167 G14
40
20
0110 1k
120
140
160
100k
G = 1000
G = 100
G = 10
G = 1
V
S
= ±15V
T
A
= 25°C
1k SOURCE
IMBALANCE
Common Mode Rejection Ratio
vs Frequency
FREQUENCY (Hz)
0.1
POSITIVE POWER SUPPLY REJECTION RATIO (dB)
60
80
100
100 10k
1167 G16
40
20
0110 1k
120
140
160
100k
G = 1000
G = 10
G = 1
V
–
= –15V
T
A
= 25°C
G = 100
Positive Power Supply Rejection
Ratio vs Frequency Gain vs Frequency
FREQUENCY (kHz)
0
GAIN (dB)
10
30
50
60
0.01 1 10 1000
1167 G17
–10
0.1 100
40
20
–20
G = 1000
G = 100
G = 10
G = 1
V
S
= ±15V
T
A
= 25°C
FREQUENCY (Hz)
1
0
100
1000
10 100 1k 100k10k
1167 G19
10
VOLTAGE NOISE DENSITY (nV√Hz)
V
S
= ±15V
T
A
= 25°C
1/f
CORNER
= 10Hz
1/f
CORNER
= 9Hz
1/f
CORNER
= 7Hz
GAIN = 1
GAIN = 10
GAIN = 100, 1000
BW LIMIT
GAIN = 1000
Voltage Noise Density
vs Frequency
TIME (SEC)
0
NOISE VOLTAGE (2µV/DIV)
8
1167 G20
24510
6
139
7
V
S
= ±15V
T
A
= 25°C
0.1Hz to 10Hz Noise Voltage,
G = 1
0.1Hz to 10Hz Noise Voltage, RTI
G = 1000
TIME (SEC)
0
NOISE VOLTAGE (0.2µV/DIV)
8
1167 G21
24510
6
139
7
V
S
= ±15V
T
A
= 25°C
9
LT1167
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Current Noise Density
vs Frequency
FREQUENCY (Hz)
1
10
CURRENT NOISE DENSITY (fA/√Hz)
100
1000
10 100 1000
1167 G22
V
S
= ±15V
T
A
= 25°C
R
S
TIME (SEC)
0
CURRENT NOISE (5pA/DIV)
8
1167 G23
24510
6
139
7
V
S
= ±15V
T
A
= 25°C
0.1Hz to 10Hz Current Noise
TIME FROM OUTPUT SHORT TO GROUND (MINUTES)
0
–50
(SINK) (SOURCE)
OUTPUT CURRENT (mA)
–40
–20
–10
0
50
20
12
1167 G24
–30
30
40
10
3
T
A
= –40°C
V
S
= ±15V
T
A
= –40°C
T
A
= 25°C
T
A
= 85°C
T
A
= 85°C
T
A
= 25°C
Short-Circuit Current vs Time
5V/DIV
10µs/DIV
Large-Signal Transient Response
1167 G28
20mV/DIV
10µs/DIV
G = 1
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
Small-Signal Transient Response
1167 G29
G = 1
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
5V/DIV
10µs/DIV
Large-Signal Transient Response
20mV/DIV
10µs/DIV
G = 10
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
Small-Signal Transient Response
1167 G32
G = 10
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
1167 G31
Overshoot vs Capacitive Load
CAPACITIVE LOAD (pF)
10
40
OVERSHOOT (%)
50
60
70
80
100 1000 10000
1167 G25
30
20
10
0
90
100
V
S
= ±15V
V
OUT
= ±50mV
R
L
=
∞
A
V
≥ 100
A
V
= 10
A
V
= 1
Output Impedance vs Frequency
FREQUENCY (kHz)
1
OUTPUT IMPEDANCE (Ω)
10
100
1000
10 100 1000
1167 G26
0.1 1
V
S
= ±15V
T
A
= 25°C
G = 1 TO 1000
10
LT1167
TYPICAL PERFOR A CE CHARACTERISTICS
UW
5V/DIV
10µs/DIV
Large-Signal Transient Response
1167 G34
G = 100
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
20mV/DIV
10µs/DIV
G = 100
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
Small-Signal Transient Response
1167 G35
50µs/DIV
Small-Signal Transient Response
G = 1000
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
20mV/DIV
1167 G38
50µs/DIV
Large-Signal Transient Response
G = 1000
V
S
= ±15V
R
L
= 2k
C
L
= 60pF
5V/DIV
1167 G37
Undistorted Output Swing
vs Frequency
FREQUENCY (kHz)
1
20
25
PEAK-TO-PEAK OUTPUT SWING (V)
30
35
10 100 1000
1167 G27
15
10
5
0
G = 1
G = 10, 100, 1000
VS = ±15V
TA = 25°C
Settling Time vs Gain
GAIN (dB)
1
1
SETTLING TIME (µs)
10
100
1000
10 100 1000
1167 G30
V
S
= ±15V
T
A
= 25°C
∆V
OUT
= 10V
1mV = 0.01%
Settling Time vs Step Size
SETTLING TIME (µs)
2
OUTPUT STEP (V)
2
6
10
10
1167 G33
–2
–6
0
4
8
–4
–8
–10 468
311
5
7
9
12
0V V
OUT
TO 0.1%
TO 0.1%
TO 0.01%
TO 0.01%
0V V
OUT
V
S
= ±15
G = 1
T
A
= 25°C
C
L
= 30pF
R
L
= 1k
Output Voltage Swing
vs Load Current
OUTPUT CURRENT (mA)
OUTPUT VOLTAGE SWING (V) 
(REFERRED TO SUPPLY VOLTAGE)
+V
S
+V
S
– 0.5
+V
S
– 1.0
+V
S
– 1.5
+V
S
– 2.0
–V
S
+ 2.0
–V
S
+ 1.5
–V
S
+ 1.0
–V
S
+ 0.5
–V
S
0.01 1 10 100
1167 G39
0.1
V
S
= ±15V 85°C
25°C
–40°C
SOURCE
SINK
Slew Rate vs Temperature
TEMPERATURE (°C)
–50 –25
0.8
SLEW RATE (V/µs)
1.2
1.8
050 75
1167 G36
1.0
1.6
1.4
25 100 125
V
S
= ±15V
V
OUT
= ±10V
G = 1
+SLEW
–SLEW
11
LT1167
BLOCK DIAGRAM
W
Q1
R
G
2
OUTPUT
6
REF
1167 F01
5
7
–
+
A1
–
+
A3
VB
R1
24.7k
R3
400Ω
R4
400Ω
C1
1
R
G
8
R7
10k R8
10k
R5
10k R6
10k
DIFFERENCE AMPLIFIER STAGEPREAMP STAGE
+IN
–IN
3
–
+
A2
VB
R2
24.7k
C2
V
+
V
–
V
–
V
+
V
–
Q2 V
–
V
+
4V
–
with programmed gain. Therefore, the bandwidth does not
drop proportional to gain.
The input transistors Q1 and Q2 offer excellent matching,
which is inherent in NPN bipolar transistors, as well as
picoampere input bias current due to superbeta process-
ing. The collector currents in Q1 and Q2 are held constant
due to the feedback through the Q1-A1-R1 loop and
Q2-A2-R2 loop which in turn impresses the differential
input voltage across the external gain set resistor R
G
.
Since the current that flows through R
G
also flows through
R
1 and R2, the ratios provide a gained-up differential volt-
age, G = (R1 + R2)/R
G
, to the unity-gain difference
amplifier
A3. The common mode voltage is removed by A3, result-
ing in a single-ended output voltage referenced to the
voltage on the REF pin. The resulting gain equation is:
V
OUT
– V
REF
= G(V
IN+
– V
IN–
)
where:
G = (49.4kΩ/R
G
) + 1
solving for the gain set resistor gives:
R
G
= 49.4kΩ/(G – 1)
THEORY OF OPERATIO
U
The LT1167 is a modified version of the three op amp
instrumentation amplifier. Laser trimming and monolithic
construction allow tight matching and tracking of circuit
parameters over the specified temperature range. Refer to
the block diagram (Figure 1) to understand the following
circuit description. The collector currents in Q1 and Q2 are
trimmed to minimize offset voltage drift, thus assuring a
high level of performance. R1 and R2 are trimmed to an
absolute value of 24.7k to assure that the gain can be set
accurately (0.05% at G = 100) with only one external
resistor R
G
. The value of R
G
in parallel with R1 (R2)
determines the transconductance of the preamp stage. As
R
G
is reduced for larger programmed gains, the transcon-
ductance of the input preamp stage increases to that of the
input transistors Q1 and Q2. This increases the open-loop
gain when the programmed gain is increased, reducing
the input referred gain related errors and noise. The input
voltage noise at gains greater than 50 is determined only
by Q1 and Q2. At lower gains the noise of the difference
amplifier and preamp gain setting resistors increase the
noise. The gain bandwidth product is determined by C1,
C2 and the preamp transconductance which increases
Figure 1. Block Diagram
12
LT1167
Input and Output Offset Voltage
The offset voltage of the LT1167 has two components: the
output offset and the input offset. The total offset voltage
referred to the input (RTI) is found by dividing the output
offset by the programmed gain (G) and adding it to the
input offset. At high gains the input offset voltage domi-
nates, whereas at low gains the output offset voltage
dominates. The total offset voltage is:
Total input offset voltage (RTI)
= input offset + (output offset/G)
Total output offset voltage (RTO)
= (input offset • G) + output offset
Reference Terminal
The reference terminal is one end of one of the four 10k
resistors around the difference amplifier. The output volt-
age of the LT1167 (Pin 6) is referenced to the voltage on
the reference terminal (Pin 5). Resistance in series with
the REF pin must be minimized for best common mode
rejection. For example, a 2Ω resistance from the REF pin
to ground will not only increase the gain error by 0.02%
but will lower the CMRR to 80dB.
Single Supply Operation
For single supply operation, the REF pin can be at the same
potential as the negative supply (Pin 4) provided the
output of the instrumentation amplifier remains inside the
specified operating range and that one of the inputs is at
least 2.5V above ground. The barometer application on the
front page of this data sheet is an example that satisfies
these conditions. The resistance R
b
from the bridge trans-
ducer to ground sets the operating current for the bridge
and also has the effect of raising the input common mode
voltage. The output of the LT1167 is always inside the
specified range since the barometric pressure rarely goes
low enough to cause the output to rail (30.00 inches of Hg
corresponds to 3.000V). For applications that require the
output to swing at or below the REF potential, the voltage
on the REF pin can be level shifted. An op amp is used to
buffer the voltage on the REF pin since a parasitic series
resistance will degrade the CMRR. The application in the
back of this data sheet, Four Digit Pressure Sensor, is an
example.
THEORY OF OPERATIO
U
Output Offset Trimming
The LT1167 is laser trimmed for low offset voltage so that
no external offset trimming is required for most applica-
tions. In the event that the offset needs to be adjusted, the
circuit in Figure 2 is an example of an optional offset adjust
circuit. The op amp buffer provides a low impedance to the
REF pin where resistance must be kept to minimum for
best CMRR and lowest gain error.
Input Bias Current Return Path
The low input bias current of the LT1167 (350pA) and the
high input impedance (200GΩ) allow the use of high
impedance sources without introducing additional offset
voltage errors, even when the full common mode range is
required. However, a path must be provided for the input
bias currents of both inputs when a purely differential
signal is being amplified. Without this path the inputs will
float to either rail and exceed the input common mode
range of the LT1167, resulting in a saturated input stage.
Figure 3 shows three examples of an input bias current
path. The first example is of a purely differential signal
source with a 10kΩ input current path to ground. Since the
impedance of the signal source is low, only one resistor is
needed. Two matching resistors are needed for higher
impedance signal sources as shown in the second
example. Balancing the input impedance improves both
common mode rejection and DC offset. The need for input
resistors is eliminated if a center tap is present as shown
in the third example.
–
+
2
–IN
OUTPUT
+IN
1
8
10k
100Ω
100Ω
–10mV
1167 F02
V
–
V
+
10mV
5
2
3
1
6
1/2
LT1112
±10mV
ADJUSTMENT RANGE
R
G
3
–
+
LT1167
REF
Figure 2. Optional Trimming of Output Offset Voltage
13
LT1167
THEORY OF OPERATIO
U
Figure 3. Providing an Input Common Mode Current Path
A 2N4393 drain/source to gate is a good low leakage diode
for use with 1k resistors, see Figure 4. The input resistors
should be carbon and not metal film or carbon film.
RFI Reduction
In many industrial and data acquisition applications,
instrumentation amplifiers are used to accurately amplify
small signals in the presence of large common mode
voltages or high levels of noise. Typically, the sources of
these very small signals (on the order of microvolts or
millivolts) are sensors that can be a significant distance
from the signal conditioning circuit. Although these sen-
sors may be connected to signal conditioning circuitry,
using shielded or unshielded twisted-pair cabling, the ca-
bling may act as antennae, conveying very high frequency
interference directly into the input stage of the LT1167.
The amplitude and frequency of the interference can have
an adverse effect on an instrumentation amplifier’s input
stage by causing an unwanted DC shift in the amplifier’s
input offset voltage. This well known effect is called RFI
rectification and is produced when out-of-band interfer-
ence is coupled (inductively, capacitively or via radiation)
and rectified by the instrumentation amplifier’s input tran-
sistors. These transistors act as high frequency signal
detectors, in the same way diodes were used as RF
envelope detectors in early radio designs. Regardless of
the type of interference or the method by which it is
coupled into the circuit, an out-of-band error signal ap-
pears in series with the instrumentation amplifier’s inputs.
APPLICATIONS INFORMATION
WUU U
The LT1167 is a low power precision instrumentation
amplifier that requires only one external resistor to accu-
rately set the gain anywhere from 1 to 1000. The output
can handle capacitive loads up to 1000pF in any gain
configuration and the inputs are protected against ESD
strikes up to 13kV (human body).
Input Protection
The LT1167 can safely handle up to ±20mA of input
current in an overload condition. Adding an external 5k
input resistor in series with each input allows DC input
fault voltages up to ±100V and improves the ESD immu-
nity to 8kV (contact) and 15kV (air discharge), which is the
IEC 1000-4-2 level 4 specification. If lower value input
resistors are needed, a clamp diode from the positive
supply to each input will maintain the IEC 1000-4-2
specification to level 4 for both air and contact discharge.
Figure 4. Input Protection
10k
R
G
R
G
R
G
1167 F03
THERMOCOUPLE
200k
MICROPHONE,
HYDROPHONE,
ETC
200k
CENTER-TAP PROVIDES
BIAS CURRENT RETURN
–
+
LT1167
–
+
LT1167
–
+
LT1167
V
EE
1167 F04
V
CC
V
CC
V
CC
J2
2N4393
J1
2N4393
OUT
OPTIONAL FOR HIGHEST
ESD PROTECTION
R
G
R
IN
R
IN
–
+
LT1167 REF
14
LT1167
APPLICATIONS INFORMATION
WUU U
signal to be processed, set the common mode time
constant an order of magnitude (or more) larger than the
differential mode time constant. To avoid any possibility of
common mode to differential mode signal conversion,
match the common mode time constants to 1% or better.
If the sensor is an RTD or a resistive strain gauge, then the
series resistors R
S1, 2
can be omitted, if the sensor is in
proximity to the instrumentation amplifier.
“Roll Your Own”—Discrete vs Monolithic LT1167
Error Budget Analysis
The LT1167 offers performance superior to that of “roll
your own” three op amp discrete designs. A typical appli-
cation that amplifies and buffers a bridge transducer’s
differential output is shown in Figure 6. The amplifier, with
its gain set to 100, amplifies a differential, full-scale output
voltage of 20mV over the industrial range. To make the
comparison challenging, the low cost version of the LT1167
will be compared to a discrete instrumentation amp made
with the A grade of one of the best precision quad op amps,
the LT1114A. The LT1167C outperforms the discrete
amplifier that has lower V
OS
, lower I
B
and comparable V
OS
drift. The error budget comparison in Table 1 shows how
various errors are calculated and how each error affects
the total error budget. The table shows the greatest
differences between the discrete solution and the LT1167
are input offset voltage and CMRR. Note that for the
discrete solution, the noise voltage specification is multi-
plied by √2 which is the RMS sum of the uncorelated noise
of the two input amplifiers. Each of the amplifier errors is
referenced to a full-scale bridge differential voltage of
20mV. The common mode range of the bridge is 5V. The
LT1114 data sheet provides offset voltage, offset voltage
drift and offset current specifications for the matched op
amp pairs used in the error-budget table. Even with an
excellent matching op amp like the LT1114, the discrete
solution’s total error is significantly higher than the
LT1167’s total error. The LT1167 has additional advan-
tages over the discrete design, including lower compo-
nent cost and smaller size.
To significantly reduce the effect of these out-of-band
signals on the input offset voltage of instrumentation
amplifiers, simple lowpass filters can be used at the
inputs. This filter should be located very close to the input
pins of the circuit. An effective filter configuration is
illustrated in Figure 5, where three capacitors have been
added to the inputs of the LT1167. Capacitors C
XCM1
and
C
XCM2
form lowpass filters with the external series resis-
tors R
S1, 2
to any out-of-band signal appearing on each of
the input traces. Capacitor C
XD
forms a filter to reduce any
unwanted signal that would appear across the input traces.
An added benefit to using C
XD
is that the circuit’s AC
common mode rejection is not degraded due to common
mode capacitive imbalance. The differential mode and
common mode time constants associated with the capaci-
tors are:
t
DM(LPF)
= (2)(R
S
)(C
XD
)
t
CM(LPF)
= (R
S1, 2
)(C
XCM1, 2
)
Setting the time constants requires a knowledge of the
frequency, or frequencies of the interference. Once this
frequency is known, the common mode time constants
can be set followed by the differential mode time constant.
To avoid any possibility of inadvertently affecting the
V
–
V
+
IN
+
IN
–
1167 F05
V
OUT
R
G
C
XCM1
100pF
C
XCM2
100pF
C
XD
10pF
R
S1
1.6k
R
S2
1.6k
EXTERNAL RFI
FILTER
–
+
LT1167
Figure 5. Adding a Simple RC Filter at the Inputs to an
Instrumentation Amplifier is Effective in Reducing Rectification
of High Frequency Out-of-Band Signals
15
LT1167
APPLICATIONS INFORMATION
WUU U
Figure 6. “Roll Your Own” vs LT1167
“ROLL YOUR OWN”’ CIRCUIT
ERROR SOURCE LT1167C CIRCUIT CALCULATION CALCULATION LT1167C “ROLL YOUR OWN”
Absolute Accuracy at T
A
= 25
°
C
Input Offset Voltage, µV60µV/20mV 100µV/20mV 3000 5000
Output Offset Voltage, µV (300µV/100)/20mV [(60µV)(2)/100]/20mV 150 60
Input Offset Current, nA [(450pA)(350/2)Ω]/20mV [(450pA)(350Ω)/2]/20mV 4 4
CMR, dB 110dB→[(3.16ppm)(5V)]/20mV [(0.02% Match)(5V)]/20mV 790 500
Total Absolute Error 3944 5564
Drift to 85
°
C
Gain Drift, ppm/°C (50ppm + 10ppm)(60°C) (100ppm/°C Track)(60°C) 3600 6000
Input Offset Voltage Drift, µV/°C [(0.4µV/°C)(60°C)]/20mV [(1.6µV/°C)(60°C)]/20mV 1200 4800
Output Offset Voltage Drift, µV/°C[6µV/°C)(60°C)]/100/20mV [(1.1µV/°C)(2)(60°C)]/100/20mV 180 66
Total Drift Error 4980 10866
Resolution
Gain Nonlinearity, ppm of Full Scale 15ppm 10ppm 15 10
Typ 0.1Hz to 10Hz Voltage Noise, µV
P-P
0.28µV
P-P
/20mV (0.3µV
P-P
)(√2)/20mV 14 21
Total Resolution Error 29 31
Grand Total Error 8953 16461
G = 100, V
S
= ±15V
All errors are min/max and referred to input.
Table 1. “Roll Your Own” vs LT1167 Error Budget
ERROR, ppm OF FULL SCALE
Current Source
Figure 7 shows a simple, accurate, low power program-
mable current source. The differential voltage across Pins
2 and 3 is mirrored across R
G
. The voltage across R
G
is
amplified and applied across R
X
, defining the output
current. The 50µA bias current flowing from Pin 5 is
buffered by the LT1464 JFET operational amplifier. This
has the effect of improving the resolution of the current
source to 3pA, which is the maximum I
B
of the LT1464A.
Replacing R
G
with a programmable resistor greatly
increases the range of available output currents.
–
+
–
+
–
+
350Ω
350Ω
350Ω
350Ω
10V
10k**
PRECISION BRIDGE TRANSDUCER LT1167 MONOLITHIC
INSTRUMENTATION AMPLIFIER
G = 100, R
G
= ±10ppm TC
SUPPLY CURRENT = 1.3mA MAX
“ROLL YOUR OWN” INST AMP, G = 100
* 0.02 RESISTOR MATCH, 3ppm/°C TRACKING
** DISCRETE 1% RESISTOR, ±100ppm/°C TC
100ppm TRACKING
SUPPLY CURRENT = 1.35mA FOR 3 AMPLIFIERS
1167 F06
R
G
499Ω
1/4
LT1114A
1/4
LT1114A
1/4
LT1114A
10k**
100Ω**
10k*
10k* 10k*
10k*
–
+
LT1167C
REF
16
LT1167
APPLICATIONS INFORMATION
WUU U
important, R6 and C2 make up a 0.3Hz highpass filter.
The AC signal at LT1112’s Pin 5 is amplified by a gain of
101 set by (R7/R8) +1. The parallel combination of C3
and R7 form a lowpass filter that decreases this gain at
frequencies above 1kHz. The ability to operate at ±3V on
0.9mA of supply current makes the LT1167 ideal for
battery-powered applications. Total supply current for
this application is 1.7mA. Proper safeguards, such as
isolation, must be added to this circuit to protect the
patient from possible harm.
Low I
B
Favors High Impedance Bridges,
Lowers Dissipation
The LT1167’s low supply current, low supply voltage
operation and low input bias currents optimize it for
battery-powered applications. Low overall power dissi-
pation necessitates using higher impedance bridges. The
single supply pressure monitor application (Figure 9)
shows the LT1167 connected to the differential output of
a 3.5k bridge. The bridge’s impedance is almost an order
of magnitude higher than that of the bridge used in the
error-budget table. The picoampere input bias currents
keep the error caused by offset current to a negligible
level. The LT1112 level shifts the LT1167’s reference pin
and the ADC’s analog ground pins above ground. The
LT1167’s and LT1112’s combined power dissipation is
still less than the bridge’s. This circuit’s total supply
current is just 2.8mA.
–
+
3
+IN
RX
VX
IL
–IN
8
1
1167 F07
–VS
VS
5
2
3
4
7
6
1/2
LT1464
RG
2
1
LOAD
IL = = [(+IN) – (–IN)]G
RX
VX
RX
G =  + 1
49.4kΩ
RG
–
+
LT1167 REF
Figure 8. Nerve Impulse Amplifier
2
2
–IN
PATIENT
GROUND
OUTPUT
1V/mV
+IN
1
1
8
R6
1M
R7
10k
R8
100Ω
1167 F08
A
V
= 101
POLE AT 1kHz
5
5
4
–3V –3V
3V
3V
7
68
4
7
6
–
+
1/2
LT1112
1/2
LT1112
R4
30k
R3
30k
R1
12k
C1
0.01µF
R
G
6k
3
3
R2
1M
C2
0.47µF
0.3Hz
HIGHPASS
C3
15nF
PATIENT/CIRCUIT
PROTECTION/ISOLATION
–
+
LT1167
G = 10
+
–
Nerve Impulse Amplifier
The LT1167’s low current noise makes it ideal for high
source impedance EMG monitors. Demonstrating the
LT1167’s ability to amplify low level signals, the circuit in
Figure 8 takes advantage of the amplifier’s high gain and
low noise operation. This circuit amplifies the low level
nerve impulse signals received from a patient at Pins 2
and 3. R
G
and the parallel combination of R3 and R4 set
a gain of ten. The potential on LT1112’s Pin 1 creates a
ground for the common mode signal. C1 was chosen to
maintain the stability of the patient ground. The LT1167’s
high CMRR ensures that the desired differential signal is
amplified and unwanted common mode signals are at-
tenuated. Since the DC portion of the signal is not
Figure 7. Precision Voltage-to-Current Converter
17
LT1167
APPLICATIONS INFORMATION
WUU U
–
+
23
2
1
1
1
1/2
LT1112
3.5k
5V
3.5k
3.5k
3.5k
87
6
1167 F09
5
40k
20k
40k
DIGITAL
DATA
OUTPUT
4
G = 200
249Ω
3
REF
IN
AGND
ADC
LTC
®
1286
BI TECHNOLOGIES
67-8-3 R40KQ
(0.02% RATIO MATCH)
–
+
LT1167
Figure 9. Single Supply Pressure Monitor
AC Coupled Instrumentation Amplifier
2
–IN
OUTPUT
+IN
1
8R1
500k
1167 TA04
2
3
5
1
6
C1
0.3µF
–
+
1/2
LT1112
R
G
3
f
–3dB
= 1
(2π)(R1)(C1)
= 1.06Hz
–
+
LT1167
REF
TYPICAL APPLICATION
U
18
LT1167
PACKAGE DESCRIPTION
U
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
N8 1197
0.100 ± 0.010
(2.540 ± 0.254)
0.065
(1.651)
TYP
0.045 – 0.065
(1.143 – 1.651)
0.130 ± 0.005
(3.302 ± 0.127)
0.020
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.125
(3.175)
MIN
12 34
8765
0.255 ± 0.015*
(6.477 ± 0.381)
0.400*
(10.160)
MAX
0.009 – 0.015
(0.229 – 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.325 +0.035
–0.015
+0.889
–0.381
8.255
()
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
19
LT1167
PACKAGE DESCRIPTION
U
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
0.406 – 1.270
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 0996
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH 
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD 
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
20
LT1167
PART NUMBER DESCRIPTION COMMENTS
LTC1100 Precision Chopper-Stabilized Instrumentation Amplifier Best DC Accuracy
LT1101 Precision, Micropower, Single Supply Instrumentation Amplifier Fixed Gain of 10 or 100, I
S
< 105µA
LT1102 High Speed, JFET Instrumentation Amplifier Fixed Gain of 10 or 100, 30V/µs Slew Rate
LTC®1418 14-Bit, Low Power, 200ksps ADC with Serial and Parallel I/O Single Supply 5V or ±5V Operation, ±1.5LSB INL
and ±1LSB DNL Max
LT1460 Precision Series Reference Micropower; 2.5V, 5V, 10V Versions; High Precision
LT1468 16-Bit Accurate Op Amp, Low Noise Fast Settling 16-Bit Accuracy at Low and High Frequencies, 90MHz GBW,
22V/µs, 900ns Settling
LTC1562 Active RC Filter Lowpass, Bandpass, Highpass Responses; Low Noise,
Low Distortion, Four 2nd Order Filter Sections
LTC1605 16-Bit, 100ksps, Sampling ADC Single 5V Supply, Bipolar Input Range: ±10V,
Power Dissipation: 55mW Typ
1167f LT/GP 1298 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
4-Digit Pressure Sensor
TYPICAL APPLICATION
U
–
+
+
–
2
1
1
1
1
2
R8
392k
LT1634CCZ-1.25
4
11
1/4
LT1114
–
+
1/4
LT1114
3
5k
5k
5k
5k
9V
9V
5
4
3
2
28
LUCAS NOVA SENOR
NPC-1220-015A-3L
7
6
1167 TA03
5TO
4-DIGIT
DVM
4
8
R5
100k
R3
51k
R4
100k
R1
825Ω
R2
12Ω
C1
1µF
R9
1k
R
SET
R6
50k
R7
180k
3
12
14
13
6
–
+
1/4
LT1114
10
9
–
+
LT1167
G = 60
0.2% ACCURACY AT ROOM TEMP
1.2% ACCURACY AT 0°C TO 60°C
VOLTS
2.800
3.000
3.200
INCHES Hg
28.00
30.00
32.00
RELATED PARTS
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507
●
www.linear-tech.com