Agilent HCNR200/201
High-Linearity Analog Optocouplers
Data Sheet
CAUTION: It is advised that normal static precautions be taken in handling and assembly of this component to
prevent damage and/or degradation which may be induced by ESD.
3
4
1
2
VF
–
+IF
IPD1
6
5
IPD2
8
7
NC
NC
PD2 CATHODE
PD2 ANODE
LED CATHODE
LED ANODE
PD1 CATHODE
PD1 ANODE
Features
• Low nonlinearity: 0.01%
•K
3 (IPD2/IPD1) transfer gain
HCNR200: ±15%
HCNR201: ±5%
• Low gain temperature coefficient:
-65 ppm/°C
• Wide bandwidth – DC to >1 MHz
• Worldwide safety approval
- UL 1577 recognized
(5 kV rms/1 min rating)
- CSA approved
- IEC/EN/DIN EN 60747-5-2
approved
V
IORM = 1414 V peak
(option #050)
• Surface mount option available
(Option #300)
• 8-Pin DIP package - 0.400" spacing
• Allows flexible circuit design
• Special selection for HCNR201:
tighter K1, K3 and lower
nonlinearity available
Applications
• Low cost analog isolation
• Telecom: Modem, PBX
• Industrial process control:
Transducer isolator
Isolator for thermocouples 4 mA to
20 mA loop isolation
• SMPS feedback loop, SMPS
feedforward
• Monitor motor supply voltage
• Medical
Schematic
Description
The HCNR200/201 high-linearity
analog optocoupler consists of a
high-performance AlGaAs LED that
illuminates two closely matched
photodiodes. The input photodiode
can be used to monitor, and
therefore stabilize, the light output
of the LED. As a result, the non-
linearity and drift characteristics
of the LED can be virtually
eliminated. The output photodiode
produces a photocurrent that is
linearly related to the light output
of the LED. The close matching of
the photodiodes and advanced
design of the package ensure the
high linearity and stable gain
characteristics of the optocoupler.
The HCNR200/201 can be used to
isolate analog signals in a wide
variety of applications that
require good stability, linearity,
bandwidth and low cost. The
HCNR200/201 is very flexible and,
by appropriate design of the
application circuit, is capable of
operating in many different
modes, including: unipolar/
bipolar, ac/dc and inverting/non-
inverting. The HCNR200/201 is an
excellent solution for many analog
isolation problems.
2
Ordering Information
HCNR20x
0 = ±15% Transfer Gain, 0.25% Maximum Nonlinearity
1 = ±5% Transfer Gain, 0.05% Maximum Nonlinearity
Option yyyy
050 = IEC/EN/DIN EN 60747-5-2 VIORM = 1414 V peak Option
300 = Gull Wing Surface Mount Lead Option
500 = Tape/Reel Package Option (1 k min.)
XXXE = Lead Free Option
Option data sheets available. Contact your Agilent Technologies sales representative or authorized distributor
for information.
Remarks: The notation “#” is used for existing products, while (new) products launched since 15th July 2001
and lead free option will use “-”
Package Outline Drawings
Figure 1.
0.40 (0.016)
0.56 (0.022)
1
2
3
4
8
7
6
5
1.70 (0.067)
1.80 (0.071)
2.54 (0.100) TYP.
0.51 (0.021) MIN.
5.10 (0.201) MAX.
3.10 (0.122)
3.90 (0.154)
DIMENSIONS IN MILLIMETERS AND (INCHES).
NC
PD1
K1
11.30 (0.445)
MAX.
PIN
ONE 1.50
(0.059)
MAX.
A
HCNR200Z
YYWW
OPTION
CODE*
DATE
CODE
8 7 6 5
12 3 4
9.00
(0.354)
TYP.
0.20 (0.008)
0.30 (0.012)
0°
15°
11.00
(0.433)
MAX.
10.16
(0.400)
TYP.
K2
PD2
NC
LED
* MARKING CODE LETTER FOR OPTION NUMBERS.
"V" = OPTION 050
OPTION NUMBERS 300 AND 500 NOT MARKED.
NOTE: FLOATING LEAD PROTRUSION IS 0.25 mm (10 mils) MAX.
3
Gull Wing Surface Mount Option #300
1.00 ± 0.15
(0.039 ± 0.006)
7° NOM.
12.30 ± 0.30
(0.484 ± 0.012)
0.75 ± 0.25
(0.030 ± 0.010)
11.00
(0.433)
5
6
7
8
4
3
2
1
11.15 ± 0.15
(0.442 ± 0.006)
9.00 ± 0.15
(0.354 ± 0.006)
1.3
(0.051)
13.56
(0.534)
2.29
(0.09)
LAND PATTERN RECOMMENDATION
1.78 ± 0.15
(0.070 ± 0.006)
4.00
(0.158)MAX.
1.55
(0.061)
MAX.
2.54
(0.100)
BSC
DIMENSIONS IN MILLIMETERS (INCHES).
LEAD COPLANARITY = 0.10 mm (0.004 INCHES).
NOTE: FLOATING LEAD PROTRUSION IS
0.25 mm (10 mils) MAX.
0.254 + 0.076
- 0.0051
(0.010+ 0.003)
- 0.002)
MAX.
4
Solder Reflow Temperature Profile
Regulatory Information
The HCNR200/201 optocoupler
features a 0.400" wide, eight pin
DIP package. This package was
specifically designed to meet
worldwide regulatory require-
ments. The HCNR200/201 has
been approved by the following
organizations:
UL
Recognized under UL 1577,
Component Recognition Program,
FILE E55361
CSA
Approved under CSA Component
Acceptance Notice #5, File CA
88324
IEC/EN/DIN EN 60747-5-2
Approved under
IEC 60747-5-2:1997 + A1:2002
EN 60747-5-2:2001 + A1:2002
DIN EN 60747-5-2 (VDE 0884
Teil 2):2003-01
(Option 050 only)
Recommended Pb-Free IR Profile
0
TIME (SECONDS)
TEMPERATURE (°C)
200
100
50 150100 200 250
300
0
30
SEC.
50 SEC.
30
SEC.
160°C
140°C
150°C
PEAK
TEMP.
245°C
PEAK
TEMP.
240°CPEAK
TEMP.
230°C
SOLDERING
TIME
200°C
PREHEATING TIME
150°C, 90 + 30 SEC.
2.5°C ± 0.5°C/SEC.
3°C + 1°C/–0.5°C
TIGHT
TYPICAL
LOOSE
ROOM
TEMPERATURE
PREHEATING RATE 3°C + 1°C/–0.5°C/SEC.
REFLOW HEATING RATE 2.5°C ± 0.5°C/SEC.
217 °C
RAMP-DOWN
6 °C/SEC. MAX.
RAMP-UP
3 °C/SEC. MAX.
150 - 200 °C
260 +0/-5 °C
t 25 °C to PEAK
60 to 150 SEC.
20-40 SEC.
TIME WITHIN 5 °C of ACTU AL
PEAK TEMPERA TURE
t
p
t
s
PREHEAT
60 to 180 SEC.
t
L
T
L
T
smax
T
smin
25
T
p
TIME
TEMPERATURE
NOTES:
THE TIME FROM 25 °C to PEAK TEMPERATURE = 8 MINUTES MAX.
T
smax
= 200 °C, T
smin
= 150 °C
5
IEC/EN/DIN EN 60747-5-2 Insulation Characteristics (Option #050 Only)
Description Symbol Characteristic Unit
Installation classification per DIN VDE 0110/1.89, Table 1
For rated mains voltage ≤600 V rms I-IV
For rated mains voltage ≤1000 V rms I-III
Climatic Classification (DIN IEC 68 part 1) 55/100/21
Pollution Degree (DIN VDE 0110 Part 1/1.89) 2
Maximum Working Insulation Voltage VIORM 1414 V peak
Input to Output Test Voltage, Method b* VPR 2651 V peak
VPR = 1.875 x VIORM, 100% Production Test with
tm = 1 sec, Partial Discharge < 5 pC
Input to Output Test Voltage, Method a* VPR 2121 V peak
VPR = 1.5 x VIORM, Type and sample test, tm = 60 sec,
Partial Discharge < 5 pC
Highest Allowable Overvoltage* VIOTM 8000 V peak
(Transient Overvoltage, tini = 10 sec)
Safety-Limiting Values
(Maximum values allowed in the event of a failure,
also see Figure 11)
Case Temperature TS150 °C
Current (Input Current IF, PS = 0) IS400 mA
Output Power PS,OUTPUT 700 mW
Insulation Resistance at TS, VIO = 500 V RS>109Ω
*Refer to the front of the Optocoupler section of the current catalog for a more detailed description of IEC/EN/DIN EN 60747-5-2 and other product
safety regulations.
Note: Optocouplers providing safe electrical separation per IEC/EN/DIN EN 60747-5-2 do so only within the safety-limiting values to which they are
qualified. Protective cut-out switches must be used to ensure that the safety limits are not exceeded.
Insulation and Safety Related Specifications
Parameter Symbol Value Units Conditions
Min. External Clearance L(IO1) 9.6 mm Measured from input terminals to output
(External Air Gap) terminals, shortest distance through air
Min. External Creepage L(IO2) 10.0 mm Measured from input terminals to output
(External Tracking Path) terminals, shortest distance path along body
Min. Internal Clearance 1.0 mm Through insulation distance conductor to
(Internal Plastic Gap) conductor, usually the direct distance
between the photoemitter and photodetector
inside the optocoupler cavity
Min. Internal Creepage 4.0 mm The shortest distance around the border
(Internal Tracking Path) between two different insulating materials
measured between the emitter and detector
Comparative Tracking Index CTI 200 V DIN IEC 112/VDE 0303 PART 1
Isolation Group IIIa Material group (DIN VDE 0110)
Option 300 – surface mount classification is Class A in accordance with CECC 00802.
6
Absolute Maximum Ratings
Storage Temperature ......................................................... -55°C to +125°C
Operating Temperature (TA) ............................................. -55°C to +100°C
Junction Temperature (TJ) ................................................................. 125°C
Reflow Temperature Profile ...... See Package Outline Drawings Section
Lead Solder Temperature .......................................................260°C for 10s
(up to seating plane)
Average Input Current - IF.................................................................. 25 mA
Peak Input Current - IF........................................................................ 40 mA
(50 ns maximum pulse width)
Reverse Input Voltage - VR......................................................................2.5 V
(IR = 100 µA, Pin 1-2)
Input Power Dissipation ............................................... 60 mW @ TA = 85°C
(Derate at 2.2 mW/°C for operating temperatures above 85°C)
Reverse Output Photodiode Voltage ......................................................30 V
(Pin 6-5)
Reverse Input Photodiode Voltage .........................................................30 V
(Pin 3-4)
Recommended Operating Conditions
Storage Temperature ........................................................... -40°C to +85°C
Operating Temperature ....................................................... -40°C to +85°C
Average Input Current - IF............................................................. 1 - 20 mA
Peak Input Current - IF........................................................................ 35 mA
(50% duty cycle, 1 ms pulse width)
Reverse Output Photodiode Voltage................................................. 0 - 15 V
(Pin 6-5)
Reverse Input Photodiode Voltage ................................................... 0 - 15 V
(Pin 3-4)
7
Electrical Specifications
TA = 25°C unless otherwise specified.
Parameter Symbol Device Min. Typ. Max. Units Test Conditions Fig. Note
Transfer Gain K3HCNR200 0.85 1.00 1.15 5 nA < IPD < 50 µA, 2,3 1
0 V < VPD < 15 V
HCNR201 0.95 1.00 1.05 5 nA < IPD < 50 µA, 1,2
0 V < VPD < 15 V
HCNR201 0.93 1.00 1.07 -40°C < TA < 85°C, 1,2
5 nA < IPD < 50 µA,
0 V < VPD < 15 V
Temperature ∆K3/∆TA -65 ppm/°C-40°C < TA < 85°C, 2,3
Coefficient of 5 nA < IPD < 50 µA,
Transfer Gain 0 V < VPD < 15 V
DC NonLinearity NLBF HCNR200 0.01 0.25 % 5 nA < IPD < 50 µA, 4,5, 3
(Best Fit) 0 V < VPD < 15 V 6
HCNR201 0.01 0.05 5 nA < IPD < 50 µA, 2,3
0 V < VPD < 15 V
HCNR201 0.01 0.07 -40°C < TA < 85°C, 2,3
5 nA < IPD < 50 µA,
0 V < VPD < 15 V
DC Nonlinearity NLEF 0.016 5 nA < IPD < 50 µA, 4
(Ends Fit) 0 V < VPD < 15 V
Input Photo- K1HCNR200 0.25 0.50 0.75 % IF = 10 mA, 7 2
diode Current 0 V < VPD1 < 15 V
Transfer Ratio HCNR201 0.36 0.48 0.72
(IPD1/IF)
Temperature ∆K1/∆TA-0.3 %/°C-40°C < TA < 85°C, 7
Coefficient IF = 10 mA
of K10 V < VPD1 < 15 V
Photodiode ILK 0.5 25 nA IF = 0 mA, 8
Leakage Current 0 V < VPD < 15 V
Photodiode BVRPD 30 150 V IR = 100 µA
Reverse Break-
down Voltage
Photodiode CPD 22 pF VPD = 0 V
Capacitance
LED Forward VF1.3 1.6 1.85 V IF = 10 mA 9,
Voltage 10
1.2 1.6 1.95 IF = 10 mA,
-40°C < TA < 85°C
LED Reverse BVR2.5 9 V IF = 100 µA
Breakdown
Voltage
Temperature ∆VF/∆TA-1.7 mV/°CI
F = 10 mA
Coefficient of
Forward Voltage
LED Junction CLED 80 pF f = 1 MHz,
Capacitance VF = 0 V
8
AC Electrical Specifications
TA = 25°C unless otherwise specified.
Test
Parameter Symbol Device Min. Typ. Max. Units Conditions Fig. Note
LED Bandwidth f -3dB 9 MHz IF = 10 mA
Application Circuit Bandwidth:
High Speed 1.5 MHz 16 7
High Precision 10 kHz 17 7
Application Circuit: IMRR
High Speed 95 dB freq = 60 Hz 16 7, 8
Package Characteristics
TA = 25°C unless otherwise specified.
Test
Parameter Symbol Device Min. Typ. Max. Units Conditions Fig. Note
Input-Output VISO 5000 V rms RH ≤50%, 5, 6
Momentary-Withstand t = 1 min.
Voltage*
Resistance RI-O 1012 1013 ΩVO = 500 VDC 5
(Input-Output)
1011 TA = 100°C, 5
VIO = 500 VDC
Capacitance CI-O 0.4 0.6 pF f = 1 MHz 5
(Input-Output)
Notes:
1. K3 is calculated from the slope of the best fit
line of IPD2 vs. IPD1 with eleven equally
distributed data points from 5 nA to 50 µA.
This is approximately equal to IPD2/IPD1 at
IF = 10 mA.
2. Special selection for tighter K1, K3 and lower
Nonlinearity available.
3. BEST FIT DC NONLINEARITY (NLBF) is the
maximum deviation expressed as a
percentage of the full scale output of a “best
fit” straight line from a graph of IPD2 vs. IPD1
with eleven equally distributed data points
from 5 nA to 50 µA. IPD2 error to best fit line is
the deviation below and above the best fit
line, expressed as a percentage of the full
scale output.
4. ENDS FIT DC NONLINEARITY (NLEF) is the
maximum deviation expressed as a
percentage of full scale output of a straight
line from the 5 nA to the 50 µA data point on
the graph of IPD2 vs. IPD1.
5. Device considered a two-terminal device:
Pins 1, 2, 3, and 4 shorted together and pins
5, 6, 7, and 8 shorted together.
6. In accordance with UL 1577, each
optocoupler is proof tested by applying an
insulation test voltage of ≥6000 V rms for ≥1
second (leakage detection current limit, II-O of
5 µA max.). This test is performed before the
100% production test for partial discharge
(method b) shown in the IEC/EN/DIN EN
60747-5-2 Insulation Characteris-tics Table
(for Option #050 only).
7. Specific performance will depend on circuit
topology and components.
8. IMRR is defined as the ratio of the signal gain
(with signal applied to VIN of Figure 16) to the
isolation mode gain (with VIN connected to
input common and the signal applied
between the input and output commons) at
60 Hz, expressed in dB.
*The Input-Output Momentary Withstand Voltage is a dielectric voltage rating that should not be interpreted as an input-output continuous voltage
rating. For the continuous voltage rating refer to the VDE 0884 Insulation Characteristics Table (if applicable), your equipment level safety specification,
or Application Note 1074, “Optocoupler Input-Output Endurance Voltage.”
9
Figure 5. NLBF vs. temperature.
Figure 2. Normalized K3 vs. input IPD. Figure 3. K3 drift vs. temperature. Figure 4. IPD2 error vs. input IPD (see note 4).
Figure 6. NLBF drift vs. temperature. Figure 7. Input photodiode CTR vs. LED input
current.
Figure 8. Typical photodiode leakage vs.
temperature. Figure 9. LED input current vs. forward
voltage. Figure 10. LED forward voltage vs.
temperature.
I
LK
– PHOTODIODE LEAKAGE – nA
10.0
4.0
0.0
T
A
– TEMPERATURE – °C
6.0
2.0
8.0
-25-55 5 35 65 95 125
V
PD
= 15 V
DELTA K3 – DRIFT OF K3 TRANSFER GAIN
0.02
-0.005
-0.02
T
A
– TEMPERATURE – °C
0.01
0.005
-0.01
-0.015 = DELTA K3 MEAN
= DELTA K3 MEAN ± 2 • STD DEV
0.0
0.015
-25-55 5 35 65 95 125
0 V < V
PD
< 15 V
DELTA NL
BF
– DRIFT OF BEST-FIT NL – % PTS
0.02
-0.005
-0.02
T
A
– TEMPERATURE – °C
0.01
0.005
-0.01
-0.015 = DELTA NL
BF
MEAN
= DELTA NL
BF
MEAN ± 2 • STD DEV
0.0
0.015
-25-55 5 35 65 95 125
0 V < V
PD
< 15 V
5 nA < I
PD
< 50 µA
NORMALIZED K1 – INPUT PHOTODIODE CTR
0.0
0.5
0.2
I
F
– LED INPUT CURRENT – mA
2.0 6.0 12.0
0.6
0.4
0.3
4.0 8.0 10.0
0.7
0.8
0.9
1.0
1.1
1.2
14.0 16.0
-55°C
25°C
-40°C
85°C
100°C
NORMALIZED TO K1 CTR
AT I
F
= 10 mA, T
A
= 25°C
0 V < V
PD1
< 15 V
V
F
– LED FORWARD VOLTAGE – V
1.5
1.2
T
A
– TEMPERATURE – °C
1.8
1.7
1.4
1.3
1.6
-25-55 5 35 65 95 125
I
F
= 10 mA
NORMALIZED K3 – TRANSFER GAIN
0.0
1.06
1.00
0.94
I
PD1
– INPUT PHOTODIODE CURRENT – µA
10.0 30.0 60.0
1.04
1.02
0.98
0.96
20.0 40.0 50.0
= NORM K3 MEAN
= NORM K3 MEAN ± 2 • STD DEV
NORMALIZED TO BEST-FIT K3 AT T
A
= 25°C,
0 V < V
PD
< 15 V
0.0
0.03
0.00
-0.03
I
PD1
– INPUT PHOTODIODE CURRENT – µA
10.0 30.0 60.0
0.02
0.01
-0.01
-0.02
20.0 40.0 50.0
= ERROR MEAN
= ERROR MEAN ± 2 • STD DEV
I
PD2
ERROR FROM BEST-FIT LINE (% OF FS)
T
A
= 25 °C, 0 V < V
PD
< 15 V
NL
BF
– BEST-FIT NON-LINEARITY – %
0.015
0.00
T
A
– TEMPERATURE – °C
0.03
0.025
0.01
0.005
= NL
BF
50TH PERCENTILE
= NL
BF
90TH PERCENTILE
0.02
0.035
-25-55 5 35 65 95 125
0 V < V
PD
< 15 V
5 nA < I
PD
< 50 µA
1.20
100
0.1
0.0001
V
F
– FORWARD VOLTAGE – VOLTS
1.30 1.50
10
1
0.01
0.001
1.40 1.60
I
F
– FORWARD CURRENT – mA
T
A
= 25°C
10
Figure 12. Basic isolation amplifier.
I
F
LED
I
PD1
PD1
R1
V
IN
A1
+
-I
PD2
PD2
R2
A2
-
+V
OUT
PD1
R1
V
IN
A1
-
+PD2 PD2
R2
A2
-
+V
OUT
A) BASIC TOPOLOGY
B) PRACTICAL CIRCUIT
C1
R3
V
CC
LED C2
Figure 11. Thermal derating curve dependence of safety limiting value
with case temperature per IEC/EN/DIN EN 60747-5-2.
-
+
V
IN
-
+
V
OUT
V
IN
-
+
-
+V
OUT
A) POSITIVE INPUT
V
CC
B) POSITIVE OUTPUT
C) NEGATIVE INPUT D) NEGATIVE OUTPUT
Figure 13. Unipolar circuit topologies.
0
800
300
0
T
S
– CASE TEMPERATURE – °C
25 75 150
600
500
200
100
50 100 125
P
S
OUTPUT POWER – mV
I
S
INPUT CURRENT – mA
400
700
900
1000
175
11
Figure 15. Loop-powered 4-20 mA current loop circuits.
Figure 14. Bipolar circuit topologies.
-
+
-
+
V
OUT
V
IN
-
+
-
+V
OUT
A) SINGLE OPTOCOUPLER
V
CC1
B) DUAL OPTOCOUPLER
V
CC1
IOS1
V
CC2
IOS2
V
IN
-
+
V
CC
-
+V
OUT
+I
IN
-
+
-
+
+I
OUT
A) RECEIVER
B) TRANSMITTER
PD2
V
IN
-
+
V
CC
-I
IN
R1
R3
PD1
LED
D1
R2
R1
PD1
LED
-I
OUT
R2
R3
PD2
D1 Q1
12
Figure 18. Bipolar isolation amplifier.
Figure 16. High-speed low-cost analog isolator.
VIN
VCC1 +5 V
R1
68 K
PD1
LED
R3
10 K
Q1
2N3906 R4
10
Q2
2N3904
VCC2 +5 V
R2
68 K
PD2
R5
10 K
Q3
2N3906 R6
10
Q4
2N3904
R7
470
VOUT
-
+
PD1 2
3A1
7
4
R1
200 K
INPUT
BNC 1%
C3
0.1µ
V
CC1
+15 V
C1
47
P
LT1097
R6
6.8 K
R4
2.2 K R5
270
Q1
2N3906
V
EE1
-15 V
C4
0.1µ R3
33 K
LED D1
1N4150
-
+PD2
2
3
A2
7
4
C2
33
P
OUTPUT
BNC
174 K
LT1097
50 K
1 %
V
EE2
-15 V
C6
0.1µ
R2
C5
0.1µ
V
CC2
+15 V
6
6
Figure 17. Precision analog isolation amplifier.
-
+V
MAG
-
+
V
IN
OC1
PD1
+
-
OC2
PD1
R1
50 K
D2
C2 10 pf
C1 10 pf
D1 R4
680
R5
680
OC1
LED
OC2
LED
R3
180 K
R2
180 K
BALANCE
C3 10 pf
OC1
PD2
R6
180 K R7
50 K
GAIN
OC2
PD2
V
CC1
= +15 V
V
EE1
= -15 V
13
Figure 20. SPICE model listing.
Figure 19. Magnitude/sign isolation amplifier.
H
.SUBCKT HCNR200
-
+VMAG
-
+
VIN OC1
PD1
+
-D4
C2 10 pf
C1 10 pf
D3
R4
680
OC1
LED
R1
220 K
C3 10 pf
OC1
PD2
R5
180 K R6
50 K
GAIN
R2
10 K R3
4.7 K
D1
-
+
D2
+
-R7
6.8 K
VCC
R8
2.2 K
VSIGN
OC2
6N139
VCC1 = +15 V
VEE1 = -15 V
14
Figure 21. 4 to 20 mA HCNR200 receiver circuit.
Figure 22. 4 to 20 mA HCNR200 transmitter circuit.
-
+VOUT
-
+
VCC
5.5 V
R1
10 kΩ
+ILOOP
HCNR200
PD 1
-ILOOP
R2
10 kΩ
R4
180 Ω
2N3906
Z1
5.1 V
0.1 µF
R3
25 Ω
0.001 µF
R5
80 kΩ
LM158
HCNR200
PD 2
0.001 µF
2
HCNR200
LED
LM158
DESIGN EQUATIONS:
VOUT / ILOOP = K3 (R5 R3) / R1 + R3)
K3 = K2 / K1 = CONSTANT = 1
NOTE: THE TWO OP-AMPS SHOWN ARE TWO SEPARATE LM158, AND NOT TWO CHANNELS IN A SINGLE
DUAL PACKAGE, OTHERWISE THE LOOP SIDE AND OUTPUT SIDE WILL NOT BE PROPERLY ISOLATED.
-
+
V
CC
R3
10 kΩ
+ILOOP
HCNR200
PD 2
-ILOOP
R4
10 kΩ
R6
140 Ω
2N3904
Z1
5.1 V
0.1 µF
R7
3.2 kΩ0.001 µF
R1
80 kΩ
LM158
HCNR200
PD 1
0.001 µF
1
R8
100 kΩ
VIN
V
CC
5.5 V
R2
150 Ω
HCNR200
LED
2N3906 2N3904
2N3904
R5
25 Ω
LM158
-
+
DESIGN EQUATIONS:
(ILOOP / V
IN
) = K3 (R5 + R3) / R5 R1)
K3 = K2 / K1 = CONSTANT = 1
NOTE: THE TWO OP-AMPS SHOWN ARE TWO SEPARATE LM158, AND NOT TWO CHANNELS IN A SINGLE
DUAL PACKAGE, OTHERWISE THE LOOP SIDE AND OUTPUT SIDE WILL NOT BE PROPERLY ISOLATED.
15
Theory of Operation
Figure 1 illustrates how the
HCNR200/201 high-linearity opto-
coupler is configured. The basic
optocoupler consists of an LED
and two photodiodes. The LED
and one of the photodiodes (PD1)
is on the input leadframe and the
other photodiode (PD2) is on the
output leadframe. The package of
the optocoupler is constructed so
that each photodiode receives
approximately the same amount of
light from the LED.
An external feedback amplifier
can be used with PD1 to monitor
the light output of the LED and
automatically adjust the LED
current to compensate for any
non-linearities or changes in light
output of the LED. The feedback
amplifier acts to stabilize and
linearize the light output of the
LED. The output photodiode then
converts the stable, linear light
output of the LED into a current,
which can then be converted back
into a voltage by another
amplifier.
Figure 12a illustrates the basic
circuit topology for implementing
a simple isolation amplifier using
the HCNR200/201 optocoupler.
Besides the optocoupler, two
external op-amps and two
resistors are required. This simple
circuit is actually a bit too simple
to function properly in an actual
circuit, but it is quite useful for
explaining how the basic isolation
amplifier circuit works (a few
more components and a circuit
change are required to make a
practical circuit, like the one
shown in Figure 12b).
The operation of the basic circuit
may not be immediately obvious
just from inspecting Figure 12a,
particularly the input part of the
circuit. Stated briefly, amplifier A1
adjusts the LED current (IF), and
therefore the current in PD1
(IPD1), to maintain its “+” input
terminal at 0 V. For example,
increasing the input voltage
would tend to increase the voltage
of the “+” input terminal of A1
above 0 V. A1 amplifies that
increase, causing IF to increase, as
well as IPD1. Because of the way
that PD1 is connected, IPD1 will
pull the “+” terminal of the op-
amp back toward ground. A1 will
continue to increase IF until its
“+” terminal is back at 0 V.
Assuming that A1 is a perfect op-
amp, no current flows into the
inputs of A1; therefore, all of the
current flowing through R1 will
flow through PD1. Since the “+”
input of A1 is at 0 V, the current
through R1, and therefore IPD1 as
well, is equal to VIN/R1.
Essentially, amplifier A1 adjusts
IF so that
IPD1 = VIN/R1.
Notice that IPD1 depends ONLY on
the input voltage and the value of
R1 and is independent of the light
output characteristics of the LED.
As the light output of the LED
changes with temperature, ampli-
fier A1 adjusts IF to compensate
and maintain a constant current
in PD1. Also notice that IPD1 is
exactly proportional to VIN, giving
a very linear relationship between
the input voltage and the
photodiode current.
The relationship between the
input optical power and the
output current of a photodiode is
very linear. Therefore, by stabiliz-
ing and linearizing IPD1, the light
output of the LED is also
stabilized and linearized. And
since light from the LED falls on
both of the photodiodes, IPD2 will
be stabilized as well.
The physical construction of the
package determines the relative
amounts of light that fall on the
two photodiodes and, therefore,
the ratio of the photodiode
currents. This results in very
stable operation over time and
temperature. The photodiode
current ratio can be expressed as
a constant, K, where
K = IPD2/IPD1.
Amplifier A2 and resistor R2 form
a trans-resistance amplifier that
converts IPD2 back into a voltage,
VOUT, where
VOUT = IPD2*R2.
Combining the above three
equations yields an overall
expression relating the output
voltage to the input voltage,
VOUT/VIN = K*(R2/R1).
Therefore the relationship
between VIN and VOUT is constant,
linear, and independent of the
light output characteristics of the
LED. The gain of the basic isola-
tion amplifier circuit can be
adjusted simply by adjusting the
ratio of R2 to R1. The parameter K
(called K3 in the electrical
specifications) can be thought of
as the gain of the optocoupler and
is specified in the data sheet.
Remember, the circuit in
Figure 12a is simplified in order
to explain the basic circuit opera-
tion. A practical circuit, more like
Figure 12b, will require a few
additional components to stabilize
the input part of the circuit, to
limit the LED current, or to
optimize circuit performance.
Example application circuits will
be discussed later in the data
sheet.
16
example circuits that operate with
bipolar input signals are
discussed in the next section.
As a final example of circuit
design flexibility, the simplified
schematics in Figure 15 illustrate
how to implement 4-20 mA analog
current-loop transmitter and
receiver circuits using the
HCNR200/201 optocoupler. An
important feature of these circuits
is that the loop side of the circuit
is powered entirely by the loop
current, eliminating the need for
an isolated power supply.
The input and output circuits in
Figure 15a are the same as the
negative input and positive output
circuits shown in Figures 13c and
13b, except for the addition of R3
and zener diode D1 on the input
side of the circuit. D1 regulates
the supply voltage for the input
amplifier, while R3 forms a
current divider with R1 to scale
the loop current down from 20
mA to an appropriate level for the
input circuit (<50 µA).
As in the simpler circuits, the
input amplifier adjusts the LED
current so that both of its input
terminals are at the same voltage.
The loop current is then divided
between R1 and R3. IPD1 is equal
to the current in R1 and is given
by the following equation:
IPD1 = ILOOP*R3/(R1+R3).
Combining the above equation
with the equations used for Figure
12a yields an overall expression
relating the output voltage to the
loop current,
VOUT/ILOOP = K*(R2*R3)/(R1+R3).
Again, you can see that the
relationship is constant, linear,
and independent of the charac-
teristics of the LED.
Circuit Design Flexibility
Circuit design with the HCNR200/
201 is very flexible because the
LED and both photodiodes are
accessible to the designer. This
allows the designer to make perf-
ormance trade-offs that would
otherwise be difficult to make with
commercially available isolation
amplifiers (e.g., bandwidth vs.
accuracy vs. cost). Analog isolation
circuits can be designed for
applications that have either
unipolar (e.g., 0-10 V) or bipolar
(e.g., ±10 V) signals, with positive
or negative input or output
voltages. Several simplified circuit
topologies illustrating the design
flexibility of the HCNR200/201 are
discussed below.
The circuit in Figure 12a is
configured to be non-inverting with
positive input and output voltages.
By simply changing the polarity of
one or both of the photodiodes, the
LED, or the op-amp inputs, it is
possible to implement other circuit
configurations as well. Figure 13
illustrates how to change the basic
circuit to accommodate both
positive and negative input and
output voltages. The input and
output circuits can be matched to
achieve any combination of
positive and negative voltages,
allowing for both inverting and
non-inverting circuits.
All of the configurations described
above are unipolar (single polar-
ity); the circuits cannot accommo-
date a signal that might swing both
positive and negative. It is possible,
however, to use the HCNR200/201
optocoupler to implement a
bipolar isolation amplifier. Two
topologies that allow for bipolar
operation are shown in Figure 14.
The circuit in Figure 14a uses two
current sources to offset the signal
so that it appears to be unipolar to
the optocoupler. Current source
IOS1 provides enough offset to
ensure that IPD1 is always positive.
The second current source, IOS2,
provides an offset of opposite
polarity to obtain a net circuit
offset of zero. Current sources IOS1
and IOS2 can be implemented
simply as resistors connected to
suitable voltage sources.
The circuit in Figure 14b uses two
optocouplers to obtain bipolar
operation. The first optocoupler
handles the positive voltage
excursions, while the second
optocoupler handles the negative
ones. The output photodiodes are
connected in an antiparallel
configuration so that they
produce output signals of
opposite polarity.
The first circuit has the obvious
advantage of requiring only one
optocoupler; however, the offset
performance of the circuit is
dependent on the matching of IOS1
and IOS2 and is also dependent on
the gain of the optocoupler.
Changes in the gain of the opto-
coupler will directly affect the
offset of the circuit.
The offset performance of the
second circuit, on the other hand,
is much more stable; it is inde-
pendent of optocoupler gain and
has no matched current sources
to worry about. However, the
second circuit requires two
optocouplers, separate gain
adjustments for the positive and
negative portions of the signal,
and can exhibit crossover distor-
tion near zero volts. The correct
circuit to choose for an applica-
tion would depend on the
requirements of that particular
application. As with the basic
isolation amplifier circuit in
Figure 12a, the circuits in Figure
14 are simplified and would
require a few additional compo-
nents to function properly. Two
17
The 4-20 mA transmitter circuit in
Figure 15b is a little different
from the previous circuits, partic-
ularly the output circuit. The
output circuit does not directly
generate an output voltage which
is sensed by R2, it instead uses Q1
to generate an output current
which flows through R3. This
output current generates a voltage
across R3, which is then sensed
by R2. An analysis similar to the
one above yields the following
expression relating output
current to input voltage:
ILOOP/VIN = K*(R2+R3)/(R1*R3).
The preceding circuits were pre-
sented to illustrate the flexibility
in designing analog isolation
circuits using the HCNR200/201.
The next section presents several
complete schematics to illustrate
practical applications of the
HCNR200/201.
Example Application Circuits
The circuit shown in Figure 16 is
a high-speed low-cost circuit
designed for use in the feedback
path of switch-mode power
supplies. This application
requires good bandwidth, low cost
and stable gain, but does not
require very high accuracy. This
circuit is a good example of how a
designer can trade off accuracy to
achieve improvements in
bandwidth and cost. The circuit
has a bandwidth of about 1.5 MHz
with stable gain characteristics
and requires few external
components.
Although it may not appear so at
first glance, the circuit in Figure
16 is essentially the same as the
circuit in Figure 12a. Amplifier A1
is comprised of Q1, Q2, R3 and
R4, while amplifier A2 is
comprised of Q3, Q4, R5, R6 and
R7. The circuit operates in the
same manner as well; the only
difference is the performance of
amplifiers A1 and A2. The lower
gains, higher input currents and
higher offset voltages affect the
accuracy of the circuit, but not the
way it operates. Because the basic
circuit operation has not changed,
the circuit still has good gain
stability. The use of discrete
transistors instead of op-amps
allowed the design to trade off
accuracy to achieve good
bandwidth and gain stability at
low cost.
To get into a little more detail
about the circuit, R1 is selected to
achieve an LED current of about
7-10 mA at the nominal input
operating voltage according to the
following equation:
IF = (VIN/R1)/K1,
where K1 (i.e., IPD1/IF) of the
optocoupler is typically about
0.5%. R2 is then selected to
achieve the desired output voltage
according to the equation,
VOUT/VIN = R2/R1.
The purpose of R4 and R6 is to
improve the dynamic response
(i.e., stability) of the input and
output circuits by lowering the
local loop gains. R3 and R5 are
selected to provide enough
current to drive the bases of Q2
and Q4. And R7 is selected so that
Q4 operates at about the same
collector current as Q2.
The next circuit, shown in
Figure 17, is designed to achieve
the highest possible accuracy at a
reasonable cost. The high
accuracy and wide dynamic range
of the circuit is achieved by using
low-cost precision op-amps with
very low input bias currents and
offset voltages and is limited by
the performance of the opto-
coupler. The circuit is designed to
operate with input and output
voltages from 1 mV to 10 V.
The circuit operates in the same
way as the others. The only major
differences are the two compensa-
tion capacitors and additional
LED drive circuitry. In the high-
speed circuit discussed above, the
input and output circuits are
stabilized by reducing the local
loop gains of the input and output
circuits. Because reducing the
loop gains would decrease the
accuracy of the circuit, two
compensation capacitors, C1 and
C2, are instead used to improve
circuit stability. These capacitors
also limit the bandwidth of the
circuit to about 10 kHz and can be
used to reduce the output noise of
the circuit by reducing its
bandwidth even further.
The additional LED drive
circuitry (Q1 and R3 through R6)
helps to maintain the accuracy
and bandwidth of the circuit over
the entire range of input voltages.
Without these components, the
transconductance of the LED
driver would decrease at low
input voltages and LED currents.
This would reduce the loop gain
of the input circuit, reducing
circuit accuracy and bandwidth.
D1 prevents excessive reverse
voltage from being applied to the
LED when the LED turns off
completely.
No offset adjustment of the circuit
is necessary; the gain can be
adjusted to unity by simply
adjusting the 50 kohm poten-
tiometer that is part of R2. Any
OP-97 type of op-amp can be used
in the circuit, such as the LT1097
from Linear Technology or the
AD705 from Analog Devices, both
of which offer pA bias currents,
µV offset voltages and are low
cost. The input terminals of the
op-amps and the photodiodes are
connected in the circuit using
Kelvin connections to help ensure
the accuracy of the circuit.
The next two circuits illustrate
how the HCNR200/201 can be
used with bipolar input signals.
The isolation amplifier in
Figure 18 is a practical implemen-
tation of the circuit shown in
Figure 14b. It uses two opto-
couplers, OC1 and OC2; OC1
handles the positive portions of
the input signal and OC2 handles
the negative portions.
Diodes D1 and D2 help reduce
crossover distortion by keeping
both amplifiers active during both
positive and negative portions of
the input signal. For example,
when the input signal positive,
optocoupler OC1 is active while
OC2 is turned off. However, the
amplifier controlling OC2 is kept
active by D2, allowing it to turn
on OC2 more rapidly when the
input signal goes negative,
thereby reducing crossover
distortion.
Balance control R1 adjusts the
relative gain for the positive and
negative portions of the input
signal, gain control R7 adjusts the
overall gain of the isolation
amplifier, and capacitors C1-C3
provide compensation to stabilize
the amplifiers.
The final circuit shown in
Figure 19 isolates a bipolar analog
signal using only one optocoupler
and generates two output signals:
an analog signal proportional to
the magnitude of the input signal
and a digital signal corresponding
to the sign of the input signal.
This circuit is especially useful for
applications where the output of
the circuit is going to be applied
to an analog-to-digital converter.
The primary advantages of this
circuit are very good linearity and
offset, with only a single gain
adjustment and no offset or
balance adjustments.
18
To achieve very high linearity for
bipolar signals, the gain should be
exactly the same for both positive
and negative input polarities. This
circuit achieves excellent linearity
by using a single optocoupler and
a single input resistor, which
guarantees identical gain for both
positive and negative polarities of
the input signal. This precise
matching of gain for both polari-
ties is much more difficult to
obtain when separate components
are used for the different input
polarities, such as is the previous
circuit.
The circuit in Figure 19 is actually
very similar to the previous
circuit. As mentioned above, only
one optocoupler is used. Because
a photodiode can conduct current
in only one direction, two diodes
(D1 and D2) are used to steer the
input current to the appropriate
terminal of input photodiode PD1
to allow bipolar input currents.
Normally the forward voltage
drops of the diodes would cause a
serious linearity or accuracy
problem. However, an additional
amplifier is used to provide an
appropriate offset voltage to the
other amplifiers that exactly
cancels the diode voltage drops to
maintain circuit accuracy.
Diodes D3 and D4 perform two
different functions; the diodes
keep their respective amplifiers
active independent of the input
signal polarity (as in the previous
circuit), and they also provide the
feedback signal to PD1 that
cancels the voltage drops of
diodes D1 and D2.
Either a comparator or an extra
op-amp can be used to sense the
polarity of the input signal and
drive an inexpensive digital
optocoupler, like a 6N139.
It is also possible to convert this
circuit into a fully bipolar circuit
(with a bipolar output signal) by
using the output of the 6N139 to
drive some CMOS switches to
switch the polarity of PD2
depending on the polarity of the
input signal, obtaining a bipolar
output voltage swing.
HCNR200/201 SPICE Model
Figure 20 is the net list of a SPICE
macro-model for the HCNR200/
201 high-linearity optocoupler.
The macro-model accurately
reflects the primary
characteristics of the HCNR200/
201 and should facilitate the
design and understanding of
circuits using the HCNR200/201
optocoupler.
www.agilent.com/semiconductors
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distributors, please go to our web site.
For technical assistance call:
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Data subject to change.
Copyright © 2005 Agilent Technologies, Inc.
Obsoletes 5989-0286EN
October 7, 2005
5989-2137EN
