ets Ms 19-0160; Rev 2; 4/97 rit) hey KIT D MIAAILM Dual-Output Power-Supply Controller for Notebook Computers General Description The MAX786 is a system-engineered power-supply controller for notebook computers or similar battery- powered equipment. It provides two high-performance step-down (buck) pulse-width modulators (PWMs) for +3.3V and +5V. Other features include dual, low-dropout, micropower linear regulators for CMOS/RTC back-up, and two precision low-battery- detection comparators. High efficiency (95% at 2A; greater than 80% at loads from 5mA to 3A) is achieved through synchronous recti- fication and PWM operation at heavy loads, and Idle Mode operation at light loads. The MAX786 uses physically small components, thanks to high operating frequencies (300kKHz/200kHz) and a new current-mode PWM architecture that allows for output filter capacitors as small as 30UF per ampere of load. Line- and load- transient responses are terrific, with a high 60kHz unity- gain crossover frequency allowing output transients to be corrected within four or five clock cycles. Low sys- tem cost is achieved through a high level of integration and the use of low-cost, external N-channel MOSFETs. Other features include low-noise, fixed-frequency PWM operation at moderate to heavy loads, and a synchro- nizable oscillator for noise-sensitive applications such as electromagnetic pen-based systems and communi- cating computers. The MAX786 is a monolithic, BiCMOS IC available in fine-pitch, surface-mount SSOP packages. Applications Notebook Computers Portable Data Terminals Communicating Computers Pen-Entry Systems ______iTypical Application Diagram Features Dual PWM Buck Controllers (+3.3V and +5V) Two Precision Comparators or Level Translators 95% Efficiency 420pA Quiescent Current, 70pA in Standby (linear regulators alive) --.hUcOHmhU OH @ 25yA Shutdown Current (+5V linear alive) # 5.5V to 30V Input Range Small SSOP Package @ Fixed Output Voltages: 3.3V (standard) 3.45V (High-Speed Pentium) 3.6V (PowerPC) Ordering Information PART TEMP. RANGE PIN-PACKAGE VouT MAX786CAI 0C to +70C 28 SSOP 3.3V MAX786RCAI 0C to +70C 28 SSOP 3.45V n 43.3V 5.5V POWER BP TO = | AMAXIA SECTION, | MEMORY 30 +5] PERIPHERALS| SHUTDOWN ONER GOOD 5V ON/OFF p 4 OF 3.3V No LOW. BATTERY WARNING SUSPEND POWER Ordering Information continued at end of data sheet. Pin Configuration TCP VIBV css [4] 29] FEB ans [3] 26] LB 01 [4] ALAXLAA [25 8st vH[6| 3] Ve o[7| pa] WL a [a] [21] FES an [9 /20] PGND FEF [10 i 9] DLs sync [11] 1a] BST SHDN [12] 17] 6 ss [14 15] C86 SSOP Idle Mode is a trademark of Maxim Integrated Products. Pentium is a trademark of Intel Corp. PowerPC is a trademark of IBM Corp. MAXIM Maxim Integrated Products 1 For free samples & the latest literature: http://)www.maxim-ic.com, or phone 1-800-998-8800. For small orders, phone 1-800-835-8769. 98ZXVINMAX 786 Dual-Output Power-Supply Controller for Notebook Computers ABSOLUTE MAXIMUM RATINGS Vt tO GND oo cece teeter reer renee eeeeeaeeae se seeseeeeaes -0.3V to 36V DH3 to LX8 oo. eee eee eee teeeeee eee eeeeeeteaeees -0.3V to (BST3 + 0.3V) PGND to GND 0... e ete eeeeeeeee eee teeter eee eeaeeaeeeaeeaeee eee eaeeee +2V DHS to LX5 ooo ee cece eee cee cess tee eeeeeeeaeees -0.3V to (BST5 + 0.3V) VL to GND 2... ceccecceecesessssesaeeeeceeseeeeeeseesesessssssssaeeeeeeees -0.3V to 7V REF, VL Short to GND...........:cccsscceccecceeeeeeeeeessssessseeees Momentary BST3, BST5 to GND... -0.3V to 36V REF Current LX3 tO BSTS oot cece ences eens enaneneneee -7V to 0.3V VL CUITON oe cece cece eee e eee tee ener eee teeta eeaeeae teeter eeeneaeeaeees LX5 tO BST5 woe eee eens sees eeeasceenanenenes -7V to 0.3V Continuous Power Dissipation (T, = + 70C) Inputs/Outputs to GND SSOP (derate 9.52mW/C above +70C) wo... cee 762mW (D1, D2, SHDN, ON5, REF, SS5, CS5, Operating Temperature Ranges FB5, SYNC, CS3,FB3, SS3, ON3)............ -0.3V to (VL + 0.3V) MAX786CAI/MAX786_CALI ....... eee ee rere 0C to +70C VH to GND 00. eeeeee ence eee eeeeeeeeeeeeeeseeeeeeeeeeeeeeeteeeeetes -0.3V to 20V MAX786EAI/MAX786_EAL ..........::eeeeee 40C to +85C Q1, Q2 to GND... .-0.3V to (VH + 0.3V) Lead Temperature (soldering, 10SC) oo... eee +300C DL3, DL5 to PGND... eee eee -0.3V to (VL + 0.3V) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (V+ = 15V, GND = PGND = OV, ly, = Iper = OMA, SHDN = ONS = ONS = 5V, other digital input levels are OV or +5V, Ta = Ton tO Tax, unless otherwise noted.) PARAMETER | CONDITIONS MIN TYP MAX | UNITS 3.3V AND 5V STEP-DOWN CONTROLLERS Input Supply Range 5.5 30 Vv FB5 Output Voltage OmV < (CSS-FBS) < 70mV, BV < V+ < SOV 480 5.08 5.20 Vv (includes load and line regulation) MAX786 3.17 3.35 3.46 FB3 Output Voltage OmV < (CS3-FB3) < 70mV, BV < V+ < SOV anager 3.32 3.50 3.60 Vv (includes load and line regulation) MAX786S 3.46 3.65 3.75 Load Regulation Either controller (CS_-FB_= OmV to 70mV) 2.5 % Line Regulation Either controller (V+ = 6V to 30V) 0.03 %IN Current-Limit Voltage CS3-FB3 or CS5-FB5 80 100 120 mV $S3/SS5 Source Current 2.5 4.0 6.5 pA $S3/SS5 Fault Sink Current 2 mA INTERNAL REGULATOR AND REFERENCE VL Output Voltage ONS = ON3 = OV, 5.5V < V+ < 30V, OMA < IL < 25mA 4.5 5.5 Vv VL Fault Lockout Voltage Falling edge, hysteresis = 1% 3.6 4.2 Vv VL/FB5 Switchover Voltage Rising edge of FB5, hysteresis = 1% 4.2 4.7 Vv REF Output Voltage No external load (Note 1) 3.24 3.36 Vv REF Fault Lockout Voltage Falling edge 2.4 3.2 Vv REF Load Regulation OmA < IL < 5mA (Note 2) 30 75 mV V+ Shutdown Current SHDN = D1 = D2 = ON3 = ONS = OV, V+ = 30V 25 40 pA V+ Standby Current D1 = D2 = ON3 = ON5 = OV, V+ = 30V 70 120 pA Quiescent Power Consumption D1 = D2 = OV, FB5 = CS5 = 5.25V, 55 86 mW (both PWM controllers on) FB3 = CS3 = 3.5V V+ Off Current FB5 = CS5 = 5.25V, VL switched over to FB5 30 60 pA COMPARATORS D1, D2 Trip Voltage Falling edge, hysteresis = 1% 1.61 1.69 Vv D1, D2 Input Current D1 = D2 = OV, 5V +100 nA 2 MAXUM(V+ = 15V, GND = PGND = OV, ly, = Iper = OMA, SHDN = ONS = ONS = 5V, other digital input levels are OV or +5V, Dual-Output Power-Supply Controller for Notebook Computers ELECTRICAL CHARACTERISTICS (continued) Ta = Tun tO Tyax, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX | UNITS Q1, Q2 Source Current VH = 15V, Vout = 2.5V 12 20 30 pA Q1, Q2 Sink Current VH = 15V, Vout = 2.5V 200 500 1000 pA Q1, Q2 Output High Voltage IsouRCE = SHA, VH = 3V VH -0.5 Vv Q1, Q2 Output Low Voltage Igink = 20HA, VH = 3V 0.4 Vv Quiescent VH Current VH = 18V, D1 = D2 = 5V, no external load 4 10 pA OSCILLATOR AND INPUTS/OUTPUTS Oscillator Frequency SYNC = 3.3V 270 300 390 kHz SYNC = OV, 5V 170 200 230 SYNC High Pulse Width 200 ns SYNC Low Pulse Width 200 ns SYNC Rise/Fall Time Not tested 200 ns Oscillator SYNC Range 240 350 kHz Maximum Duty Cycle SYNC = 3.3V 89 92 % SYNC = OV or 5V 92 95 Input Low Voltage SHDN, ON3, ONS, SYNC 0.8 Vv Input High Voltage SHDN, ONS, ONS 24 SYNC VL-0.5 Input Current SHDN, ON3, ONS Vix = OV, 5V +1 LA DL3/DL5 Sink/Source Current | Voy7 = 2V 1 A DH3/DH5 Sink/Source Current | BST3-LX3 = BST5-LX5 = 4.5V, Voy7 = 2V 1 A DL3/DL5 On-Resistance High or low 7 Q DH3/DH5 On-Resistance High or low, BST3-LX3 = BST5-LX5 = 4.5V 7 Q Note 1: Since the reference uses VL as its supply, its V+ line regulation error is insignificant. Note 2: The main switching outputs track the reference voltage. Loading the reference reduces the main outputs slightly according to the closed-loop gain (AVcL) and the reference voltage load-regulation error. AVc_ for the +3.3V supply is unity gain. AVc_ for the +5V supply is 1.54. MAXIM 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers (Circuit of Figure 1, TA = +25C, unless otherwise noted.) EFACIENCY vs. +5V QUTPUT EFFICIENCY vs. +5V QUTPUT EFFICGENCY vs. +3.3V QUTPUT Typical Operating Characteristics CURRENT, 200kHz CURRENT, 300kHz CURRENT, 200kHz TTT lB OTT . Wee Lert nt Vin= 15V Vin=6V Lt oN ett SS eet oof TIN ne SoS 9 [7] Vin=8 Cone 4 NI 20 WS Tt | 7) viv= 18 SN MT To = TT Vive Vin= 15 gS rT = el = 0 VIN=90V sw = Z LH a0V a0 iil Tye TAS oO al 2 LW J SYNC = OV, 43.3VCFF | 60 y | 60 /) 60 SYNC=0V, SVN ol alt . 1m 40m 400m 1 10 1m 40m 400m 1 10 1m 40m 400m 1 10 45V CUTPUT CLPRENT (A) 45V CUTPUT CLPRENT (A) 43.3V CUTPUT CUFFENT (A) EFACIENCY vs. +3.3V CUIPUT QUIESCENT SUPPLY CURRENT vs. STANDBY SUPPLY CURRENT vs. CURRENT, 300kHz SUPPLY VOLTAGE SUPPLY VOLTAGE 100 19 25 HTM TT] VIN= 18V 90 STN z,|_| E 20 Wr Lael ns r Vin=6V YW Ws Lamond f I Sy font mt [7 ay ONB = CNB = HGH 15 5 Ya] Yin= SV : 2 o , CNB =CN5 -0V oO 7 a E 1.0 45VCN a 1 = 6 LU 5 | NI = as el [aetna 59 WW 0 0 1m 40m 400m 1 10 0 6 12 18 24 30 0 6 12 18 A 8 43.3V CUTPUT CURRENT (A) SUPPLY VOLTAGE(V) SUPPLY VOLTAGE(V) SHUTDOWN SUPPLY CURRENT vs. MINIMUM Vj TO Vo yy DIFFERENTIAL S\MTCHINGFREGUENCY vs. SUPPLY VOLTAGE vs. +5V QUTPUT CURRENT LOAD CURRENT 500 1.0 1000 TT > / SYNC = REF (300kHz) s ~ ONG = ONS = 5V = wo |4 E os Hy = sop LAU dl Date A / 5 45V, Vin=75V Wf M1 300 GB as "| S| | We x Lo re SHON=0V 3 a J H& 10 F 45V, Vin=30V TW 3 0 Lamont A O Wil 200 =, 04 DA. 20K 2 ve LTO INI | p WLU) +3.3V, Viy = 7.5V 5 | leadatet 5 4 ee A ly .3V, Vin = 7.5V | 4100 S 02 46V CUTPUT a er lL a 2 STILL REGULATING Le = WLU rT 0 0 0.1 0 6 42 18 A 30 im 10m 100m 1 10 100 1m 10m 100m 1 SUPPLY VOLTAGE() +5V QUTPUT CUPFENT (A) LOAD CURRENT (A) 4 MA AXIMADual-Output Power-Supply Controller for Notebook Computers Typical Operating Characteristics (continued) (Circuit of Figure 1, TA = +25C, unless otherwise noted.) IDLE MODE WAVEFORMS PULSE Wi DTH MODULATION MODE WAVEFORMS +5V QUTPUT all SOr\/Gv LX10Vdiv eV i " | sev CLIPUT 50rVdiv 200us/div Vin=15V MAXUM 5 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers (Circuit of Figure 1, TA = +25C, unless otherwise noted.) +5V LINE: TRANSIENT RESPONSE, RISING ee ee 45VCUIPUT i DON div Midv Typical Operating Characteristics (continued) +5V LINE- TRANSIENT RESPONSE, FALLING Aidiv MAXIMADual-Output Power-Supply Controller for Notebook Computers Pin Description PIN NAME FUNCTION 1 CS3 Current-sense input for +3.3V; +100mV = current limit level referenced to FB3. 2 SS3 Soft-start input for +3.3V. Ramp time to full current limit is 1ms/nF of capacitance to GND. 3 ON3 ON/OFF control input disables the +3.3V PWM. Tie directly to VL for automatic start-up. 4 D1 #1 level-translator/comparator noninverting input, threshold = +1.650V. Controls Q1. Tie to GND if unused. 5 D2 #2 level-translator/comparator noninverting input (see D1) 6 VH External positive supply-voltage input for the level translators/comparators 7 Qe #2 level-translator/comparator output. Sources 20UA from VH when D2 is high. Sinks 500pA to GND when D2 is low, even with VH = OV. 8 Q1 #1 level translator/comparator output (see Q2) GND Low-current analog ground 10 REF 3.3V reference output. Sources up to 5mA for external loads. Bypass to GND with 1pF/mA of load or 0.22uF minimum. 11 SYNC Oscillator control/synchronization input. Connect to VL or GND for 200kHz; connect to REF for 300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition causes a new cycle to start. 12 SHON Shutdown control input, active low. Tie to VL for automatic Start-up. The 5V VL supply stays active in shutdown, but all other circuitry is disabled. Do not force SHDN higher than VL + 0.3V. 13 ON5 ON/OFF control input disables the +5V PWM supply. Tie to VL for automatic start-up. 14 S35 Soft-start control input for +5V. Ramp time to full current limit is 1ms/nF of capacitance to GND. 15 Css Current-sense input for +5V; +100mV = current-limit level referenced to FBS. 16 DH5 Gate-drive output for the +5V high-side MOSFET 17 LX5 Inductor connection for the +5V supply 18 BST5 Boost capacitor connection for the +5V supply (0.1pF) 19 DL5 Gate-drive output for the +5V low-side MOSFET 20 PGND Power ground 21 FB5 Feedback and current-sense input for the +5V PWM 22 VL 5V logic supply voltage for internal circuitry. VL is always on and can source 5mA for external loads. 23 V+ Supply voltage input from battery, 5.5V to 30V 24 DL3 Gate-drive output for the +3.3V low-side MOSFET 25 BST3 Boost capacitor connection for the +3.3V supply (0.1pF) 26 LX3 Inductor connection for the +3.3V supply 27 DH3 Gate-drive output for the +3.3V high-side MOSFET 28 FB3 Feedback and current-sense input for the +5V PWM MAXUM 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers Detailed Description The MAX786 converts a 5.5V to 30V input to four outputs (Figure 1). It produces two high-power, PWM, switch- mode supplies, one at +5V and the other at +3.3V. The two supplies operate at either 300kHz or 200kHz, allowing for small external components. Output current capability depends on external components, and can exceed 6A on each supply. An internal 5V, 5mA supply (VL) and a 3.3V, 5mA reference voltage are also gener- ated via linear regulators, as shown in Figure 2. Fault protection circuitry shuts off the PWMs when the inter- nal supplies lose regulation. Two precision voltage comparators are also included. Their output stages permit them to be used as level translators for driving external N-channel MOSFETs in load-switching applications, or for more conventional logic signals. The MAX786 has two close relatives: the MAX782 and the MAX783. The MAX782 and MAX783 each include an extra flyback winding regulator and linear regulators for dual, +12V/programmable PCMCIA VPP outputs. The MAX782/MAX783 data sheet contains extra appli- cations information on the MAX786 not found in this data sheet. +3.3V Switch-Mode Supply The +3.3V supply is generated by a current-mode, PWM step-down regulator using two N-channel MOSFETs, a rectifier, and an LC output filter (Figure 1). The gate-drive signal to the high-side MOSFET, which must exceed the battery voltage, is provided by a boost circuit that uses a 100nF capacitor connected to BSTS. INPUT SV TOSOV e (NEN) cr _|+ e it co arp za | = Ve Ww 2 +5V AT 5A CPA a te N14 14148, 4mF 4 T & 25 18 OAyF = p | OtRF 27 a =P 16 | BV M 45V ay me 10H cal " 1Z FN or AT 3A be Ds 10.4H 25m 3A > 4 WET os Tt T DI 24 19 Ds T SQUF == inssig 4B NB |___4 03 O15 -o _____| NA ingeig = C7 Cl2 MA AXLAA 150uF 150uF =s 1 MAX?86 16 = = cs3 css XX 281 pas as f2! XX (NOTE2) 5 1 (NOTE2) ee a = _ 0.01~F = 3 ss ss 6 = OotuF 483V0NCRE A" Q VH COCMPARATCRSUPPLY INPUT 13 4 45VCNOF ON6 DI IN OcMPARATCR SHUTDOWN 12) aN a fe our 11 5 CEC SYNC , SYNC re : IN cowpapaTore | @ Our | NOTE: INPUT VOLTAGE RANGES 5VTOS0V | ONS AS SHON SEELOMVCLTAGE | 9 [10 [2 (6-CELL) CPERATIONSECTICN FOR | = = DEIALS. 433VATSmA NOTE2: USESHORT, KELVIN-OCNNECTED PC It 3 BOARD TRACES PLACED VERY CLCBE Te TOCKEANOTHER = Figure 1. MAX786 Application Circuit MAXIMADual-Output Power-Supply Controller for Notebook Computers A synchronous rectifier at LX3 keeps efficiency high by clamping the voltage across the rectifier diode. Maximum current limit is set by an external low-value sense resistor, which prevents excessive inductor cur- rent during start-up or under short-circuit conditions. Programmable soft-start is set by an optional external capacitor; this reduces in-rush surge currents upon start-up and provides adjustable power-up times for power-supply sequencing purposes. AESZHRE PGND a AES SHRE Figure 2. MAX786 Block Diagram MAXUM 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers +5V Switch-Mode Supply The +5V output is produced by a current-mode, PWM step-down regulator, which is nearly identical to the +3.3V supply. The +5V supplys dropout voltage, as configured in Figure 1, is typically 400mV at 2A. As V+ approaches 5V, the +5V output gracefully falls with V+ until the VL regulator output hits its undervoltage- lockout threshold at 4V. At this point, the +5V supply turns off. The default frequency for both PWM controllers is 300kHz (with SYNC connected to REF), but 200kHz may be used by connecting SYNC to GND or VL. +3.3V and +5V PWM Buck Controllers The two current-mode PWM controllers are identical except for different preset output voltages (Figure 3). Each PWM is independent except for being synchro- nized to a master oscillator and sharing a common ref- erence (REF) and logic supply (VL). Each PWM can be turned on and off separately via ON3 and ON5. The PWMs are a direct-summing type, lacking a tradi- tional integrator error amplifier and the phase shift associated with it. They therefore do not require any external feedback compensation components if the fil- ter capacitor ESR guidelines given in the Design Procedure are followed. The main gain block is an open-loop comparator that sums four input signals: an output voltage error signal, current-sense signal, slope-compensation ramp, and precision voltage reference. This direct-summing method approaches the ideal of cycle-by-cycle control of the output voltage. Under heavy loads, the controller operates in full PWM mode. Every pulse from the oscil- lator sets the output latch and turns on the high-side switch for a period determined by the duty cycle (approximately Voyt/Vin). As the high-side switch turns off, the synchronous rectifier latch is set and, 60Ons later, the low-side switch turns on (and stays on until the beginning of the next clock cycle, in continuous mode, or until the inductor current crosses through zero, in discontinuous mode). Under fault conditions where the inductor current exceeds the 100mV current-limit threshold, the high-side latch is reset and the high-side switch is turned off. At light loads, the inductor current fails to exceed the 25mvV threshold set by the minimum current comparator. When this occurs, the PWM goes into idle mode, skip- ping most of the oscillator pulses in order to reduce the switching frequency and cut back switching losses. The oscillator is effectively gated off at light loads because the minimum current comparator immediately resets the high-side latch at the beginning of each cycle, unless the FB_ signal falls below the reference voltage level. 10 Soft-Start/SS_ Inputs Connecting capacitors to SS3 and SS5 allows gradual build-up of the +3.3V and +5V supplies after ON3 and ONS are driven high. When ON3 or ONS is low, the appropriate SS capacitors are discharged to GND. When ONS or ONS5 is driven high, a 4uA constant cur- rent source charges these capacitors up to 4V. The resulting ramp voltage on the SS_ pins linearly increas- es the current-limit comparator setpoint so as to increase the duty cycle to the external power MOSFETs up to the maximum output. With no SS capacitors, the circuit will reach maximum current limit within 10ps. Soft-start greatly reduces initial in-rush current peaks and allows start-up time to be programmed externally. Synchronous Rectifiers Synchronous rectification allows for high efficiency by reducing the losses associated with the Schottky rectifiers. When the external power MOSFET N1 (or N2) turns off, energy stored in the inductor causes its terminal volt- age to reverse instantly. Current flows in the loop formed by the inductor, Schottky diode, and loadan action that charges up the filter capacitor. The Schottky diode has a forward voltage of about 0.5V which, although small, represents a significant power loss, degrading efficiency. A synchronous rectifier, N3 (or N4), parallels the diode and is turned on by DL3 (or DL5) shortly after the diode conducts. Since the on resistance (rpgon)) of the synchronous rectifier is very low, the losses are reduced. The synchronous rectifier MOSFET is turned off when the inductor current falls to zero. Cross conduction (or shoot-through) occurs if the high-side switch turns on at the same time as the syn- chronous rectifier. The MAX786s internal break-before- make timing ensures that shoot-through does not occur. The Schottky rectifier conducts during the time that nei- ther MOSFET is on, which improves efficiency by pre- venting the synchronous-rectifier MOSFETs lossy body diode from conducting. The synchronous rectifier works under all operating condi- tions, including discontinuous-conduction and idle mode. Boost Gate-Driver Supply Gate-drive voltage for the high-side N-channel switch is generated with a flying-capacitor boost circuit as shown in Figure 4. The capacitor is alternately charged from the VL supply via the diode and placed in parallel with the high-side MOSFETs gate-source terminals. On start- up, the synchronous rectifier (low-side) MOSFET forces LX_to OV and charges the BST_ capacitor to 5V. On the MAXIMADual-Output Power-Supply Controller for Notebook Computers second half-cycle, the PWM turns on the high-side MOSFET by connecting the capacitor to the MOSFET gate by closing an internal switch between BST_ and DH_. This provides the necessary enhancement voltage to turn on the high-side switch, an action that boosts the 5V gate-drive signal above the battery voltage. Ringing seen at the high-side MOSFET gates (DH3 and DHS) in discontinuous-conduction mode (light loads) is a natural operating condition caused by the residual energy in the tank circuit formed by the inductor and stray capacitance at the LX_ nodes. The gate driver negative rail is referred to LX_, so any ringing there is directly coupled to the gate-drive supply. MA + Ww MINMLM - CURFENT (IDLEMCD Figure 3. PWM Controller Block Diagram MAXUM 11 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers Modes of Operation PWM Mode Under heavy loadsover approximately 25% of full load the +3.3V and +5V supplies operate as continuous- current PWM supplies (see Typical Operating Char- acteristics). The duty cycle (%ON) is approximately: Current flows continuously in the inductor: First, it ramps up when the power MOSFET conducts; then, it ramps down during the flyback portion of each cycle as energy is put into the inductor and then dis- charged into the load. Note that the current flowing into the inductor when it is being charged is also flow- ing into the load, so the load is continuously receiving current from the inductor. This minimizes output rip- ple and maximizes inductor use, allowing very small physical and electrical sizes. Output ripple is primarily a function of the filter capacitor (C7 or C6) effective series resistance (ESR) and is typically under 50mV (see the Design Procedure section). Output ripple is worst at light load and maximum input voltage. Idle Mode Under light loads (<25% of full load), efficiency is fur- ther enhanced by turning the drive voltage on and off for only a single clock period, skipping most of the clock pulses entirely. Asynchronous switching, seen as ghosting on an oscilloscope, is thus a normal operating condition whenever the load current is less than approximately 25% of full load. At certain input voltage and load conditions, a transition region exists where the controller can pass back and forth from idle mode to PWM mode. In this situation, short bursts of pulses occur that make the current waveform look erratic, but do not materially affect the output ripple. Efficiency remains high. Current Limiting The voltage between CS3 (CS5) and FB3 (FBS) is contin- uously monitored. An external, low-value shunt resistor is connected between these pins, in series with the inductor, allowing the inductor current to be continuously measured throughout the switching cycle. Whenever this voltage exceeds 100mV, the drive voltage to the external high-side MOSFET is cut off. This protects the MOSFET, the load, and the battery in case of short circuits or tem- porary load surges. The current-limiting resistors R1 and R2 are typically 25mQ for 3A load current. Oscillator Frequency; SYNC Input The SYNC input controls the oscillator frequency. Connecting SYNC to GND or to VL selects 200kHz opera- tion; connecting to REF selects 300kHz operation. SYNC 12 can also be driven with an external 240kHz to 350kHz CMOS/TTL source to synchronize the internal oscillator. Normally, 300kHz is used to minimize the inductor and filter capacitor sizes, but 200kHz may be necessary for low input voltages (see Low-Voltage (6-Cell) Operation). Comparators Two noninverting comparators can be used as precision voltage comparators or high-side drivers. The supply for these comparators (VH) is brought out and may be connected to any voltage between +3V and +19V irrespective of V+. The noninverting inputs (D1-D2) are high impedance, and the inverting input is internally con- nected to a 1.650V reference. Each output (Q1-Q2) sources 20yA from VH when its input is above 1.650V, and sinks 500A to GND when its input is below 1.650V. The Q1-Q2 outputs can be fixed together in wired-OR configuration since the pull-up current is only 20pA. Connecting VH to a logic supply (5V or 3V) allows the comparators to be used as low-battery detectors. For driving N-channel power MOSFETs to turn external loads on and off, VH should be 6V to 12V higher than the load voltage. This enables the MOSFETs to be fully turned on and results in low rpgon)- The comparators are always active when V+ is above +4V, even when VH is OV. Thus, Q1-@Q2 will sink current to GND even when VH is OV, but they will only source current from VH when VH is above approximately 1.5V. If Q1 or Q2 is externally pulled above VH, an internal diode conducts, pulling VH a diode drop below the output and powering anything connected to VH. This voltage will also power the other comparator outputs. BATTERY INPUT + tBen | 8 PAM LX VL DL Figure 4. Boost Supply for Gate Drivers MAXIMADual-Output Power-Supply Controller for Notebook Computers Table 1. Surface-Mount Components (See Figure 1 for Standard Application Circuit.) COMPONENT SPECIFICATION MANUFACTURER PART NO. . AVX TPSE226M035R0100 C1, C10 33uF, 35V | HF, SOV tantalum capacitors Sprague 595D336X0035R . AVX TAJB475M016 C2 4.7uF, 6V | HF, GV tantalum capacitor Sprague 595D475X0016A . AVX TAJA105M025 C3 1pF, 20V | HF, 20V tantalum capacitor Sprague 595D105X0020A2B C4, C5 0.1pF, 16V ceramic capacitors Murata-Erie GRM42-6X7R104K50V C6 330pF, 10V tantalum capacitor Sprague 595D337X0010R C7, C12 150pF, 10V tantalum capacitors Sprague 595D157X0010D C8, C9 0.01pF, 16V ceramic capacitors Murata-Erie GRM42-6X7R103K50V D2A, D2B 1N4148-type dual diodes Central Semiconductor CMPD2836 D1, D3 1N5819 SMT diodes Nihon EC10QS04 L1, Le 1OPH, 2.65A inductors Sumida CDR125-100 Ni-N4 N-channel MOSFETs (SO-8) Siliconix Si9410DY R1, R2 0.025Q, 1% (SMT) resistors IRC LR2010-01 -RO25-F Table 2. Component Suppliers FACTORY FAX COMPANY [COUNTRY CODE] | USA PHONE 803) 946-0690 AVX [1] (803) 626-3123 | (g99) 989-4975 (803) 300) Central Semiconductor | [1] (616) 435-1824 | (616) 435-1110 IRC [1] (612) 992-3377 | (512) 992-7900 [1] (814) 238-0490 | (814) 237-1431 (805) (408) (603) (247) Murata-Erie Nihon [81] 3-3494-7414 805) 867-2555 Siliconix [1] (408) 970-3950 | (408) 988-8000 Sprague [1] (603) 224-1430 | (603) 224-1961 Sumida [81] 3-3607-5144 847) 956-0666 Internal VL and REF Supplies An internal linear regulator produces the 5V used by the internal control circuits. This regulator's output is avail- able on pin VL and can source 5mA for external loads. Bypass VL to GND with 4.7uF. To save power, when the +5V switch-mode supply is above 4.5V, the internal lin- ear regulator is turned off and the high-efficiency +5V switch-mode supply output is connected to VL. The internal 3.3V bandgap reference (REF) is powered by the internal 5V VL supply. It can furnish up to 5mA. Bypass REF to GND with 0.22uF, plus 1pF/mA of load MAXUM current. The main switching outputs track the reference voltage. Loading the reference will reduce the main outputs slightly, according to the reference load-regula- tion error. Both the VL and REF outputs remain active, even when the switching regulators are turned off, to supply mem- ory keep-alive power (see Shutdown Mode section). These linear-regulator outputs can be directly connected to the corresponding step-down regulator outputs (i.e., REF to +3.3V, VL to +5V) to keep the main supplies alive in standby mode. However, to ensure start-up, standby load currents must not exceed 5mA on each supply. Fault Protection The +3.3V and +5V PWM supplies and the compara- tors are disabled when either of two faults is present: VL < +4.0V or REF < +2.8V (85% of its nominal value). Design Procedure Figure 1s schematic and Table 2s component list show values suitable for a 3A, +5V supply and a 3A, +3.3V supply. This circuit operates with input voltages from 6.5V to 30V, and maintains high efficiency with output currents between 5mA and 3A (see the Typical Operating Characteristics). This circuit's components may be changed if the design guidelines described in this section are used but before beginning the design, the following information should be firmly established: 13 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers Vin(max), the maximum input (battery) voltage. This value should include the worst-case conditions under which the power supply is expected to function, such as no-load (standby) operation when a battery charger is connected but no battery is installed. Vingmax) Cannot exceed 30V. Vincminy, the minimum input (battery) voltage. This value should be taken at the full-load operating cur- rent under the lowest battery conditions. If Vinminy is below about 6.5V, the filter capacitance required to maintain good AC load regulation increases, and the current limit for the +5V supply has to be increased for the same load level. Inductor (L1, L2) Three inductor parameters are required: the inductance value (L), the peak inductor current (ILpeax), and the coil resistance (R,). The inductance is: (Vout) (Vinqax) - Vout) (Vinamax)) (f) (lout) (LIR) Vout = output voltage (3.3V or 5V); Vin(max) = Maximum input voltage (V); f = switching frequency, normally 300kHz; lout = maximum DC load current (A); LIR = ratio of inductor peak-to-peak AC current to average DC load current, typically 0.3. A higher value of LIR allows smaller inductance, but results in higher losses and higher ripple. where: The highest peak inductor current (I peax) equals the DC load current (Igy) plus half the peak-to-peak AC inductor current (l_pp). The peak-to-peak AC inductor current is typically chosen as 30% of the maximum DC load cur- rent, so the peak inductor current is 1.15 times loyr. The peak inductor current at full load is given by: (Vout) (Vin(max) - Vout) (2) (f) (L) (Vinamax)) The coil resistance should be as low as possible, preferably in the low milliohms. The coil is effectively in series with the load at all times, so the wire losses alone are approximately: Power loss = (Igy) (RL). In general, select a standard inductor that meets the L, lLpEAK, and R, requirements (see Tables 1 and 2). Ifa standard inductor is unavailable, choose a core with an Ll2 parameter greater than (L) (ILpeax2), and use the largest wire that will fit the core. ILPEAK = lout + 14 Current-Sense Resistors (R1, R2) The sense resistors must carry the peak current in the inductor, which exceeds the full DC load current. The internal current limiting starts when the voltage across the sense resistors exceeds 100mV nominally, 80mV minimum. Use the minimum value to ensure adequate output current capability: For the +3.3V supply, R1 80mV / (1.15 xX loyt); for the +5V supply, R2 80mV/(1.15 xX lout), assuming that LIR = 0.3. Since the sense resistance values (e.g., R1 = 25mQ for lout = 3A) are similar to a few centimeters of narrow traces on a printed circuit board, trace resistance can contribute significant errors. To prevent this, Kelvin con- nect the CS_ and FB_ pins to the sense resistors; i.e., use separate traces not carrying any of the inductor or load current, as shown in Figure 5. Run these traces parallel at minimum spacing from one another. The wiring layout for these traces is critical for stable, low-ripple outputs (see the Layout and Grounding section). MOSFET Switches (N1-N4) The four N-channel power MOSFETs are usually iden- tical and must be logic-level FETs; that is, they must be fully on (have low rpgon)) ) with only 4V gate- source drive voltage. The MOSFET rpgyony should ideally be about twice the value of the sense resistor. MOSFETs with even lower rpgiony have higher gate capacitance, which increases switching time and transition losses. MOSFETs with low gate-threshold voltage specifica- tions (i.e., maximum VegTH) = 2V rather than 3V) are preferred, especially for high-current (5A) applications. Output Filter Capacitors (C6, C7, C12) The output filter capacitors determine the loop stability and output ripple voltage. To ensure stability, the mini- mum capacitance and maximum ESR values are: VREF Ce > A (Vout) (Res) (2) (2) (GBWP) and, (Vout) (Res) ESRcr < VREF where: Cpr = output filter capacitance (F); Vrer = reference voltage, 3.3V; Vout = output voltage, 3.3V or 5V; Res = sense resistor (Q); GBWP = gain-bandwidth product, 60kHz; ESRor = output filter capacitor ESR (Q). MAXIMADual-Output Power-Supply Controller for Notebook Computers Be sure to select output capacitors that satisfy both the minimum capacitance and maximum ESR require- ments. To achieve the low ESR required, it may be appropriate to use a capacitance value 2 or 3 times larger than the calculated minimum. The output ripple in continuous-current mode is: VouT(RPL) = ILPP(MAX) X (ESRep + 1/(2 x wx FX Cr) ). In idle-mode, the ripple has a capacitive and resistive component: (4) (10-4) (L) VouT(RPL)(C) = _ x (Rog?) (Cr) 1 1 + _ | Volts Vout Vin- Vout (0.02) (ESRep) VouT(RPL)(R) = Volts Res The total ripple, VoutirpL), Can be approximated as follows: (RPL)(R) < 9.5 VoutiRPLy(), then VoutRpPL) = Vout(RPL)(), ( ( R RPL) = 9.5 VoutiApL)() + RPL)(R). Diodes D1 and D3 Use 1N5819s or similar Schottky diodes. D1 and D3 conduct only about 3% of the time, so the 1N5819s 1A current rating is conservative. The voltage rating of D1 and D3 must exceed the maximum input supply voltage from the battery. These diodes must be Schottky diodes to prevent the lossy MOSFET body diodes from turning on, and they must be placed physically close to their associated synchro- nous rectifier MOSFETs. Soft-Start Capacitors (C8, C9) A capacitor connected from GND to either SS pin causes that supply to ramp up slowly. The ramp time to full current limit, tgg, is approximately ims for every nF of capacitance on SS_, with a minimum value of 10us. Typical capacitor values are in the 10nF to 100nF range; a 5V rating is sufficient. Because this ramp is applied to the current-limit circuit, the actual time for the output voltage to ramp up depends on the load current and output capacitor value. Using Figure 1s circuit with a 2A load and no SS capacitor, full output voltage is reached about 600us after ON_ is driven high. Boost Capacitors (C4, C5) Capacitors C4 and C5 store the boost voltage and pro- vide the supply for the DH3 and DH6 drivers. Use 0.1pF and place each within 10mm of the BST_ and LX_pins. Boost Diodes (D1A, D1B) Use high-speed signal diodes; e.g., 1N4148 or equivalent. Bypass Capacitors Input Filter Capacitors (C1, C10) Use at least 3LF/W of output power for the input filter capacitors, C1 and C10. They should have less than 150mQ ESR, and should be located no further than 10mm from N1 and N2 to prevent ringing. Connect the negative terminals directly to PGND. Do not exceed the surge current ratings of input bypass capacitors. Shutdown Mode Shutdown (SHDN = low) forces both PWMs off and dis- ables the REF output and both comparators (Q1 = Q2 = OV). Supply current in shutdown mode is typically 25uA. The VL supply remains active and can source 25mA for external loads. Note that the VL load capabili- ty is higher in shutdown and standby modes than when the PWMs are operating (25mA vs. 5mA). Standby mode is achieved by holding ON3 and ON5 low while SHDN is high. This disables both PWMs, but keeps VL, REF, and the precision comparators alive. Supply current in standby mode is typically 70pA. Figure 5. Kelvin Connections for the Current-Sense Resistors 15 MAXUM 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers Table 3. EV Kit Power-Supply Controls (SW1) Other ways to shut down the MAX786 are suggested in the applications section of the MAX782/MAX783 ON OFF d h SWITCH| NAME FUNCTION SETTING | SETTING ata sheet. 1 SON Enable shutdown Operate | Shutdown Applications Information Low-Voltage (6-Cell) Operation 2 ON3 Enable 3.3V 3.3V ON | 3.3V OFF The standard application circuit can be configured to power supply accept input voltages from 5.5V to 12V by changing Enable 5.0V the oscillator frequency to 200kHz and increasing the 8 ONS power supply SV ON SVOFF +5V filter capacitor to 660LF. This allows stable opera- tion at 5V loads up to 2A (the 3.3V side requires no 4 SYNC Oscillator 200kHz | 300kHz changes and still delivers 3A). Ve . cr T+ uF TT sv = Fe 23 to 1k ol ML e VH be Rio - CPN OA Le] ests BSTS RI lu viel 21) os DHS ay 0.0260 10uH 25) bg LS et a Dt 24 tr rt 1N5819 | 08 Oe G op MAXIM 150uF 150uF = 1 MAX786 sw 1 CS VF (3.3) os 2 ee m, NI-N4=Si94t10D as OG 1M T 20V Oo panel CRTWOINA psito 1 = SYNC SHDN a D1 @ te O.01pF Figure 6. MAX786 EV Kit Schematic 16 MAXIMADual-Output Power-Supply Controller for Notebook Computers EV Kit Information The MAX786 evaluation kit (EV kit) embodies the standard application circuit, with some extra pull- up and pull-down resistors needed to set default logic signal levels. The board comes configured to accept battery input voltages between 6.5V and 30V, and pro- vides up to 25W of output power. All functions are con- trolled by standard CMOS/TTL logic levels or DIP switches. The kit can be reconfigured for lower battery voltages by setting the oscillator to 200kHz and increasing the 5V output filter capacitor value. The D1 and D2 comparators can be used as precision voltage detectors by installing resistor dividers at each input. a MIAKIWVI @ a MAX786 r @ _ EVALUATION KIT i = a7 oo (comen| (mm + Lunntiiutte +-(m (a) + so lao? eel . eer ST 2mm: aap Dg PE SE e| e a= e. o @ @ ONS . ws zs 2& ao 25 a SHDN ONS ware FF F CO NH = @ wo MAOE IN THE USA /++-- 1.0" Figure 7. MAX786 EV Kit Top Component Layout and Silk Screen, Top View Figure 8. MAX786 EV Kit Ground Plane (Layers 2 and 3), Top View awh UL +10 Figure 9. MAX786 EV Kit Top Layer (Layer 1}, Top View MAXUM 17 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers BR @ G0. tA e ==* oo Ro ne tt Ke e e a= Be.5: vt oP ...., e eo ap f G iso ane GD oi in ap .. eco om Oe: en b-_ 1.0" _H Figure 10. MAX786 EV Kit, Bottom Component Layout and Silk Screen, Bottom View Figure 11. MAX786 EV Kit, Bottom Layer (Layer 4), Top View 18 MAXIMADual-Output Power-Supply Controller for Notebook Computers Chip Topography SS3 CS3 FB3 DH3 D1 D2 BST3 DL3 VH V+ 0.181" oe at Me (4.597mm) GND PGND DL5 REF SYNC SHDN ONS ss5 CS5 DH5 0.109" (2.769mm) TRANSISTOR COUNT: 1294 SUBSTRATE CONNECTED TO GND MAXUM 19 98ZXVINMAX786 Dual-Output Power-Supply Controller for Notebook Computers _ Ordering Information (continued) PART TEMP. RANGE PIN-PACKAGE VouT MAX786SCAI 0C to +70C 28 SSOP 3.6V MAX786C/D 0C to +70C Dice* _ MAX786EAI -40C to +85C 28 SSOP 3.3V MAX786REAI -40C to +85C 28 SSOP 3.45V MAX786SEAI -40C to +85C 28 SSOP 3.6V EV KIT TEMP. RANGE BOARD TYPE MAX786EVKIT-SO 0C to +70C Surface Mount *Contact factory for dice specifications. Package Information . a INCHES MILLIMETERS z "A 0 5e8 0 78 1 = 1 33 MIN MAX MIN MAX | | | | | | | Al 0.002 0.008 0.05 021 Dl 0.239 |0.249 | 6.07 | 6.33 | 14L 1 5 Tooto tons to25 tose | (ploese lozds | 6.07 | 633 | 16L 010 0. 25 |0. Dlo.27810.289| 7.07| 7.33| 20L + C_| 0.004 10.008 10.09 10.20 | |r a 317 [o.328| 8.07/ 8.33] 24 z SEE _VARTATIONS 0.39710.407 |10.07|10.33 E H E |0.205 |0.209 | 5.20 | 5.38 D}0. - - - esl e [0.0256 BSC |0.65 BSC H |0,301 |0.311 7.65 | 7.90 L |0.025 |0.037 | 0.63 | 0.95 a o 8 0 8 { . NOTES: > 1. D&E DO NOT INCLUDE MOLD FLASH. VIA X [ VI e. MOLD FLASH OR PROTRUSIONS NOT TO TITLE: EXCEED .i5mm (.006%) PACKAGE OUTLINE, SSOP, 5.3X.65mm APPROVAL DOCUMENT CONTROL NO REV 1 3, CONTROLLING DIMENSION: MILLIMETER i bose 0 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 1997 Maxim Integrated Products Printed USA MAXIM is a registered trademark of Maxim Integrated Products.