LM4961LQBD - National Semiconductor

December 2006

LM4961

Ceramic Speaker Driver

General Description

The LM4961 is an audio power amplifier primarily designed

for driving Ceramic Speaker for applications in Cell Phone

and PDAs. It integrates a boost converter, with variable output

voltage, with an audio power amplifier. It is capable of driving

15Vp-p in BTL mode to 2uF+ 30 ohms load, continuous aver-

age power, with less than 1% distortion (THD+N) from a

3.2VDC power supply.

Boomer audio power amplifiers were designed specifically to

provide high quality output power with a minimal number of

external components. The LM4961 does not require boot-

strap capacitors, or snubber circuits therefore it is ideally

suited for portable applications requiring high voltage output

to drive capacitive loads like Ceramic Speakers. The LM4961

features a low-power consumption shutdown mode. Addition-

ally, the LM4961 features an internal thermal shutdown pro-

tection mechanism.

The LM4961 contains advanced pop & click circuitry that

eliminates noises which would otherwise occur during turn-on

and turn-off transitions.

The LM4961 is unity-gain stable and can be configured by

external gain-setting resistors.

Key Specifications

■Quiescent Power Supply Current 7mA (typ)

■Voltage Swing in BTL at 1% THD 15Vp-p (typ)

■Shutdown current 0.1μA (typ)

Features

■Pop & click circuitry eliminates noise during turn-on and

turn-off transitions

■Low current shutdown mode

■Low quiescent current

■Mono 15Vp-p BTL output, RL = 2μF+30Ω, f = 1kHz

■Thermal shutdown protection

■Unity-gain stable

■External gain configuration capability

■Including Band exchange SW

■Including Leakage cut SW

Applications

■Cellphone

■PDA

Connection Diagram

LM4961LQ (5x5)

20094084

Top View

Order Number LM4961LQ

See NS Package Number LQA28A

Boomer® is a registered trademark of National Semiconductor Corporation.

LM4961 Ceramic Speaker Driver

Typical Application

20094001

* RC is needed for over/under voltage protection. If inputs are less than VDD +0.3V and greater than –0.3V, and if inputs are disabled

when in shutdown mode, then RC can be shorted.

** VFB = 1.23V

FIGURE 1. Typical Audio Amplifier Application Circuit

Shutdown 1 Shutdown 2 Band-SW

Receiver Mode (BW2) —high low

Ringer Mode (BW1) high high high

Shutdown low low low

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LM4961

Absolute Maximum Ratings (Notes 1, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (Vdd)6.0V

Amplifier Supply Voltage (V1)9.5V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJA (LLP) 66°C/W

See AN-1187 'Leadless Leadframe Packaging (LLP).'

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 3.0V < VDD < 5.0V

Amplifier Supply Voltage (V1) 2.7V < V1 < 9.0V

Electrical Characteristics VDD = 4.2V

The following specifications apply for VDD = 4.2V, AV-BTL = 26dB, RL = 2µF+30Ω, Cb = 1.0μF, Band-SW = VDD unless otherwise

specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions LM4961 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

IDD Quiescent Power Supply Current VIN = 0V, No Load

Band-SW = VDD

7 14 mA (max)

Iddrcv Iq in receiver mode VIN = 0V, No Load

Band-SW = GND

2 4 mA (max)

ISD Shutdown Current VSHUTDOWN1 = VSHUTDOWN2 = GND

Band-SW = GND (Note 9)

0.1 2.0 µA (max)

VLH Logic High Threshold Voltage For Shutdown 1, Shutdown 2, and

Band-SW

1.5 V (min)

VLL Logic Low Threshold Voltage For Shutdown 1, Shutdown 2, and

Band-SW

0.4 V (max)

RPULLDOWN Pulldown Resistor For Shutdown 2 and Band-SW 70k 50k Ω (min)

TSD Thermal Shutdown Temperature 125 °C (min)

Vout Output Voltage Swing THD = 1%, f = 1kHz

RL = 2μF+30Ω Mono BTL 15 14 Vp-p (min)

THD+N Total Harmomic Distortion + Noise Vout = 14Vp-p, f = 1kHz 0.05 1.0 % (max)

εOS Output Noise A-Weighted Filter, VIN = 0V (Note

10)

115 µV

PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz 80 65 dB (min)

Ron-sw-out On Resistance on SW-Out Band SW “High” Isink = 100µA

(Between pin 1 and pin 28)

170 220 Ω (max)

TWUA Amplifier Wake-up Time CB = 1μF25 35 ms (max)

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LM4961

Electrical Characteristics VDD = 3.2V

The following specifications apply for VDD = 3.2V, AV-BTL = 26dB, RL = 2µF+30Ω, Cb = 1.0μF, Band-SW = VDD unless otherwise

specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions LM4961 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

IDD Quiescent Power Supply Current VIN = 0V, No Load

Band-SW = VDD

9 15 mA (max)

Iddrcv Iq in receiver mode VIN = 0V, No Load

Band-SW = GND

2 4 mA (max)

ISD Shutdown Current VSHUTDOWN1 = VSHUTDOWN2 = GND

Band-SW = GND (Note 9)

0.1 2.0 µA (max)

VLH Logic High Threshold Voltage For Shutdown 1, Shutdown 2, and

Band-SW

1.5 V (min)

VLL Logic Low Threshold Voltage For Shutdown 1, Shutdown 2, and

Band-SW

0.4 V (max)

RPULLDOWN Pulldown Resistor For Shutdown 2 and Band-SW 70k 50k Ω (min)

TSD Thermal Shutdown Temperature 125 °C (min)

Vout Output Voltage Swing THD = 1%, f = 1kHz

RL = 2μF+30Ω Mono BTL 15 14 Vp-p (min)

THD+N Total Harmomic Distortion + Noise Vout = 14Vp-p, f = 1kHz 0.1 1.0 % (max)

εOS Output Noise A-Weighted Filter, VIN = 0V (Note

10)

125 µV

PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz 80 65 dB (min)

Ron-sw-out On Resistance on SW-Out Band SW “High” Isink = 100µA

(Between pin 1 and pin 28)

170 220 Ω (max)

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions

which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters

where no limit is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4961 typical application

(shown in Figure 1) with VDD = 4.2V, RL = 2μF+30Ω mono BTL operation the maximum power dissipation is 232mW. θJA = 66°C/W.

Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25°C and represent the parametric norm.

Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to GND for minimum shutdown

current.

Note 10: Noise measurements are dependent on the absolute values of closed loop gain setting resistors (input and feedback resistors).

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LM4961

Typical Performance Characteristics

THD+N vs Frequency

VDD = 4.2V, VO = 14VP-P, RL = 2μF+30Ω

20094087

THD+N vs Frequency

VDD = 3.2V, VO = 14VP-P, RL = 2μF+30Ω

200940c3

THD+N vs Output Voltage

VDD = 4.2V, RL = 2μF + 30Ω

20094002

THD+N vs Output Voltage

VDD = 3.2V, RL = 2μF + 30Ω

20094003

PSRR vs Frequency

VDD = 4.2V, RL = 8Ω, VRIPPLE = 200mVP-P

20094089

PSRR vs Frequency

VDD = 3.2V, RL = 8Ω, VRIPPLE = 20mVP-P

20094090

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LM4961

Power Dissipation vs Output Power

VDD = 4.2V, RL = 2μF + 30Ω, f = 1kHz

20094091

Power Dissipation vs Output Power

VDD = 3.2V, RL = 2μF + 30Ω, f = 1kHz

20094092

Supply Current vs Supply Voltage

RL = 2μF + 30Ω, VIN = 0V, RSOURCE = 50Ω

20094093

Frequency Response vs Input Capacitor Size

RL = 8Ω

from top to bottom:

Ci = 1.0μF, Ci = 0.39μF, Ci = 0.039μF

20094094

Switch Current Limit vs

Duty Cycle

20094095

Oscillator Frequency vs

Temperature

20094096

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LM4961

Feedback Voltage vs

Temperature

20094097

Feedback Bias Current vs

Temperature

20094098

Max. Duty Cycle vs

Temperature - ”X”

200940a0

RDS (ON) vs

Temperature

200940a1

RDS (ON) vs

VDD

200940a2

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LM4961

Application Information

BRIDGE CONFIGURATION EXPLANATION

The Audio Amplifier portion of the LM4961 has two internal

amplifiers allowing different amplifier configurations. The first

amplifier’s gain is externally configurable, whereas the sec-

ond amplifier is internally fixed in a unity-gain, inverting con-

figuration. The closed-loop gain of the first amplifier is set by

selecting the ratio of Rf to Ri while the second amplifier’s gain

is fixed by the two internal 20kΩ resistors. Figure 1 shows that

the output of amplifier one serves as the input to amplifier two.

This results in both amplifiers producing signals identical in

magnitude, but out of phase by 180°. Consequently, the dif-

ferential gain for the Audio Amplifier is

AVD = 2 *(Rf/Ri)

By driving the load differentially through outputs Vo1 and Vo2,

an amplifier configuration commonly referred to as “bridged

mode” is established. Bridged mode operation is different

from the classic single-ended amplifier configuration where

one side of the load is connected to ground.

A bridge amplifier design has a few distinct advantages over

the single-ended configuration. It provides differential drive to

the load, thus doubling the output swing for a specified supply

voltage. Four times the output power is possible as compared

to a single-ended amplifier under the same conditions.

The bridge configuration also creates a second advantage

over single-ended amplifiers. Since the differential outputs,

Vo1 and Vo2, are biased at half-supply, no net DC voltage

exists across the load. This eliminates the need for an output

coupling capacitor which is required in a single supply, single-

ended amplifier configuration. Without an output coupling

capacitor, the half-supply bias across the load would result in

both increased internal IC power dissipation and also possible

loudspeaker damage.

BOOST CONVERTER POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch

FET means the maximum output current will be determined

by power dissipation within the LM4961 FET switch. The

switch power dissipation from ON-time conduction is calcu-

lated by Equation 2.

PD(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (1)

where DC is the duty cycle.

There will be some switching losses as well, so some derating

needs to be applied when calculating IC power dissipation.

MAXIMUM AMPLIFIER POWER DISSIPATION

Power dissipation is a major concern when designing a suc-

cessful amplifier, whether the amplifier is bridged or single-

ended. A direct consequence of the increased power

delivered to the load by a bridge amplifier is an increase in

internal power dissipation. Since the amplifier portion of the

LM4961 has two operational amplifiers, the maximum internal

power dissipation is 4 times that of a single-ended amplifier.

The maximum power dissipation for a given BTL application

can be derived from Equation 1.

PDMAX(AMP) = (2VDD2) / (π2RL)(2)

where

RL = Ro1 + Ro2

MAXIMUM TOTAL POWER DISSIPATION

The total power dissipation for the LM4961 can be calculated

by adding Equation 1 and Equation 2 together to establish

Equation 3:

PDMAX(TOTAL) = (2VDD2) / (π2EFF2RL)(3)

where

EFF = Efficiency of boost converter

RL = Ro1 + Ro2

The result from Equation 3 must not be greater than the power

dissipation that results from Equation 4:

PDMAX = (TJMAX - TA) / θJA (4)

For the LQA28A, θJA = 66°C/W. TJMAX = 125°C for the

LM4961. Depending on the ambient temperature, TA, of the

system surroundings, Equation 4 can be used to find the

maximum internal power dissipation supported by the IC

packaging. If the result of Equation 3 is greater than that of

Equation 4, then either the supply voltage must be increased,

the load impedance increased or TA reduced. For the typical

application of a 4.2V power supply, with a 2uF+30Ω load, the

maximum ambient temperature possible without violating the

maximum junction temperature is approximately 109°C pro-

vided that device operation is around the maximum power

dissipation point. Thus, for typical applications, power dissi-

pation is not an issue. Power dissipation is a function of output

power and thus, if typical operation is not around the maxi-

mum power dissipation point, the ambient temperature may

be increased accordingly. Refer to the Typical Performance

Characteristics curves for power dissipation information for

lower output levels.

EXPOSED-DAP PACKAGE PCB MOUNTING

CONSIDERATIONS

The LM4961’s exposed-DAP (die attach paddle) package

(LD) provides a low thermal resistance between the die and

the PCB to which the part is mounted and soldered. The low

thermal resistance allows rapid heat transfer from the die to

the surrounding PCB copper traces, ground plane, and sur-

rounding air. The LD package should have its DAP soldered

to a copper pad on the PCB. The DAP’s PCB copper pad may

be connected to a large plane of continuous unbroken copper.

This plane forms a thermal mass, heat sink, and radiation

area. Further detailed and specific information concerning

PCB layout, fabrication, and mounting an LD (LLP) package

is found in National Semiconductor’s Package Engineering

Group under application note AN1187.

SHUTDOWN FUNCTION

In many applications, a microcontroller or microprocessor

output is used to control the shutdown circuitry to provide a

quick, smooth transition into shutdown. Another solution is to

use a single-pole, single-throw switch connected between

VDD and Shutdown pins.

BAND SWITCH FUNCTION

The LM4961 features a Band Switch function which allows

the user to use one amplifier for both receiver (earpiece)

mode and ringer/loudspeaker mode. When a logic high

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LM4961

(VDD) is applied to the Band-SW pin (pin 19) the amplifier is

in ringer mode. This enables the boost converter and sets the

externally configurable closed loop gain selection to BW1. If

the Band-SW pin has a logic low (GND) applied to its terminal

then the device is in receiver mode. In this mode the boost

converter is disabled and the gain selection is switched to

BW2. This allows the amplifier to be powered directly from the

battery minus the voltage drop across the Schottky diode.

REDUCING TRANSIENT CURRENT SPIKE

Due to the quick turn-on time of the Boost Converter, a tran-

sient supply current spike is observed on shutdown release.

To reduce the rise time of the output voltage (V1), thus re-

ducing the value of the supply current spike, please refer to

application circuit in Figure 2. Using this configuration will al-

low the user to reduce the transient supply current spike

without the Boost Converter experiencing any stability issues.

200940a3

FIGURE 2. Transient Current Spike Reduction Configuration

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using

integrated power amplifiers, and switching DC-DC convert-

ers, is critical for optimizing device and system performance.

Consideration to component values must be used to maxi-

mize overall system quality.

The best capacitors for use with the switching converter por-

tion of the LM4961 are multi-layer ceramic capacitors. They

have the lowest ESR (equivalent series resistance) and high-

est resonance frequency, which makes them optimum for

high frequency switching converters.

When selecting a ceramic capacitor, only X5R and X7R di-

electric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor. High-quality ceramic capacitors

can be obtained from Taiyo-Yuden.

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LM4961

POWER SUPPLY BYPASSING

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor location on both V1 and VDD pins should be as close

to the device as possible.

SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER

One of the major considerations is the closedloop bandwidth

of the amplifier. To a large extent, the bandwidth is dictated

by the choice of external components shown in Figure 1. The

input coupling capacitor, Ci, forms a first order high pass filter

which limits low frequency response. This value should be

chosen based on needed frequency response for a few dis-

tinct reasons.

High value input capacitors are both expensive and space

hungry in portable designs. Clearly, a certain value capacitor

is needed to couple in low frequencies without severe atten-

uation. But ceramic speakers used in portable systems,

whether internal or external, have little ability to reproduce

signals below 100Hz to 150Hz. Thus, using a high value input

capacitor may not increase actual system performance.

In addition to system cost and size, click and pop performance

is affected by the value of the input coupling capacitor, Ci. A

high value input coupling capacitor requires more charge to

reach its quiescent DC voltage (nominally 1/2 VDD). This

charge comes from the output via the feedback and is apt to

create pops upon device enable. Thus, by minimizing the ca-

pacitor value based on desired low frequency response, turn-

on pops can be minimized.

SELECTING BYPASS CAPACITOR FOR AUDIO

AMPLIFIER

Besides minimizing the input capacitor value, careful consid-

eration should be paid to the bypass capacitor value. Bypass

capacitor, CB, is the most critical component to minimize turn-

on pops since it determines how fast the amplifer turns on.

The slower the amplifier’s outputs ramp to their quiescent DC

voltage (nominally 1/2 VDD), the smaller the turn-on pop.

Choosing CB equal to 1.0µF along with a small value of Ci (in

the range of 0.039µF to 0.39µF), should produce a virtually

clickless and popless shutdown function. Although the device

will function properly, (no oscillations or motorboating), with

CB equal to 0.1µF, the device will be much more susceptible

to turn-on clicks and pops. Thus, a value of CB equal to 1.0µF

is recommended in all but the most cost sensitive designs.

SELECTING FEEDBACK CAPACITOR FOR AUDIO

AMPLIFIER

The LM4961 is unity-gain stable which gives the designer

maximum system flexibility. However, to drive ceramic speak-

ers, a typical application requires a closed-loop differential

gain of 10. In this case a feedback capacitor (Cf2) will be

needed as shown in Figure 1 to bandwidth limit the amplifier.

This feedback capacitor creates a low pass filter that elimi-

nates possible high frequency noise. Care should be taken

when calculating the -3dB frequency because an incorrect

combination of Rf and Cf2 will cause rolloff before the desired

frequency

SELECTING VALUE FOR RC

The audio power amplifier integrated in the LM4961 is de-

signed for very fast turn on time. The Cchg pin allows the input

capacitors (CinA and CinB) to charge quickly to improve click/

pop performance. Rchg1 and Rchg2 protect the Cchg pins

from any over/under voltage conditions caused by excessive

input signal or an active input signal when the device is in

shutdown. The recommended value for Rchg1 and Rchg2 is

1kΩ. If the input signal is less than VDD+0.3V and greater than

-0.3V, and if the input signal is disabled when in shutdown

mode, Rchg1 and Rchg2 may be shorted out.

SELECTING OUTPUT CAPACITOR (CO) FOR BOOST

CONVERTER

A single 4.7µF to 10µF ceramic capacitor will provide suffi-

cient output capacitance for most applications. If larger

amounts of capacitance are desired for improved line support

and transient response, tantalum capacitors can be used.

Aluminum electrolytics with ultra low ESR such as Sanyo Os-

con can be used, but are usually prohibitively expensive.

Typical AI electrolytic capacitors are not suitable for switching

frequencies above 500 kHz because of significant ringing and

temperature rise due to self-heating from ripple current. An

output capacitor with excessive ESR can also reduce phase

margin and cause instability.

In general, if electrolytics are used, we recommended that

they be paralleled with ceramic capacitors to reduce ringing,

switching losses, and output voltage ripple.

SELECTING INPUT CAPACITOR (Cs1) FOR BOOST

CONVERTER

An input capacitor is required to serve as an energy reservoir

for the current which must flow into the coil each time the

switch turns ON. This capacitor must have extremely low

ESR, so ceramic is the best choice. We recommend a nomi-

nal value of 4.7µF, but larger values can be used. Since this

capacitor reduces the amount of voltage ripple seen at the

input pin, it also reduces the amount of EMI passed back

along that line to other circuitry.

SETTING THE OUTPUT VOLTAGE (V1) OF BOOST

CONVERTER

The output voltage is set using the external resistors R2 and

R3 (see Figure 1). A value of approximately 13.3kΩ is recom-

mended for R3 to establish a divider current of approximately

92µA. R2 is calculated using the formula:

V1 = VFB [1 + R2(R3 + 170)] (5)

FEED-FORWARD COMPENSATION FOR BOOST

CONVERTER

Although the LM4961's internal Boost converter is internally

compensated, the external feed-forward capacitor Cf is re-

quired for stability (see Figure 1). Adding this capacitor puts

a zero in the loop response of the converter. The recom-

mended frequency for the zero fz should be approximately

6kHz. Cf1 can be calculated using the formula:

Cf1 = 1 / (2π x R1 x fz) (6)

SELECTING DIODES

The external diode used in Figure 1 should be a Schottky

diode. A 20V diode such as the MBR0520 from Fairchild

Semiconductor is recommended.

The MBR05XX series of diodes are designed to handle a

maximum average current of 0.5A. For applications exceed-

ing 0.5A average but less than 1A, a Microsemi UPS5817 can

be used.

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LM4961

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined as:

Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW

This applies for continuous mode operation.

INDUCTANCE VALUE

The first question we are usually asked is: “How small can I

make the inductor.” (because they are the largest sized com-

ponent and usually the most costly). The answer is not simple

and involves trade-offs in performance. Larger inductors

mean less inductor ripple current, which typically means less

output voltage ripple (for a given size of output capacitor).

Larger inductors also mean more load power can be delivered

because the energy stored during each switching cycle is:

E = L/2 x (lp)2

Where “lp” is the peak inductor current. An important point to

observe is that the LM4961 will limit its switch current based

on peak current. This means that since lp(max) is fixed, in-

creasing L will increase the maximum amount of power avail-

able to the load. Conversely, using too little inductance may

limit the amount of load current which can be drawn from the

output.

Best performance is usually obtained when the converter is

operated in “continuous” mode at the load current range of

interest, typically giving better load regulation and less output

ripple. Continuous operation is defined as not allowing the in-

ductor current to drop to zero during the cycle. It should be

noted that all boost converters shift over to discontinuous op-

eration as the output load is reduced far enough, but a larger

inductor stays “continuous” over a wider load current range.

To better understand these trade-offs, a typical application

circuit (5V to 12V boost with a 10µH inductor) will be analyzed.

We will assume:

VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V

Since the frequency is 1.6MHz (nominal), the period is ap-

proximately 0.625µs. The duty cycle will be 62.5%, which

means the ON-time of the switch is 0.390µs. It should be not-

ed that when the switch is ON, the voltage across the inductor

is approximately 4.5V. Using the equation:

V = L (di/dt)

We can then calculate the di/dt rate of the inductor which is

found to be 0.45 A/µs during the ON-time. Using these facts,

we can then show what the inductor current will look like dur-

ing operation:

20094099

FIGURE 3. 10μH Inductor Current

5V - 12V Boost (LM4961X)

During the 0.390µs ON-time, the inductor current ramps up

0.176A and ramps down an equal amount during the OFF-

time. This is defined as the inductor “ripple current”. It can also

be seen that if the load current drops to about 33mA, the in-

ductor current will begin touching the zero axis which means

it will be in discontinuous mode. A similar analysis can be

performed on any boost converter, to make sure the ripple

current is reasonable and continuous operation will be main-

tained at the typical load current values. Taiyo-Yudens

NR4012 inductor series is recommended.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the current

limiter cuts in is dependent on duty cycle of the application.

This is illustrated in a graph in the typical performance char-

acterization section which shows typical values of switch

current as a function of effective (actual) duty cycle.

CALCULATING OUTPUT CURRENT OF BOOST

CONVERTER (IAMP)

As shown in Figure 2 which depicts inductor current, the load

current is related to the average inductor current by the rela-

tion:

ILOAD = IIND(AVG) x (1 - DC) (7)

Where "DC" is the duty cycle of the application. The switch

current can be found by:

ISW = IIND(AVG) + 1/2 (IRIPPLE) (8)

Inductor ripple current is dependent on inductance, duty cy-

cle, input voltage and frequency:

IRIPPLE = DC x (VIN-VSW) / (f x L) (9)

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/2FL (10)

The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-OFF switching

losses of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET "ON" voltage (referred to as VSW in

equations 7 thru 10) is dependent on load current. A good

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LM4961

approximation can be obtained by multiplying the "ON Resis-

tance" of the FET times the average inductor current.

FET on resistance increases at VIN values below 5V, since

the internal N-FET has less gate voltage in this input voltage

range (see Typical Performance Characteristics curves).

Above VIN = 5V, the FET gate voltage is internally clamped to

5V.

The maximum peak switch current the device can deliver is

dependent on duty cycle. For higher duty cycles, see Typical

Performance Characteristics curves.

INDUCTOR SUPPLIERS

The recommended inductors for the LM4961 is the Taiyo-Yu-

den NR4012. When selecting an inductor, make certain that

the continuous current rating is high enough to avoid satura-

tion at peak currents. A suitable core type must be used to

minimize core (switching) losses, and wire power losses must

be considered when selecting the current rating.

PCB LAYOUT GUIDELINES

High frequency boost converters require very careful layout

of components in order to get stable operation and low noise.

All components must be as close as possible to the LM4961

device. It is recommended that a 4-layer PCB be used so that

internal ground planes are available. See Figures 4–7 for de-

mo board reference schematic and layout.

Some additional guidelines to be observed:

1. Keep the path between L1, D1, and Co extremely short.

Parasitic trace inductance in series with D1 and Co will in-

crease noise and ringing.

2. The feedback components R1, R2 and Cf 1 must be kept

close to the FB pin of U1 to prevent noise injection on the FB

pin trace.

3. If internal ground planes are available (recommended) use

vias to connect directly to ground at pin 2 of U1, as well as the

negative sides of capacitors Cs1 and Co.

GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION

This section provides practical guidelines for mixed signal

PCB layout that involves various digital/analog power and

ground traces. Designers should note that these are only

"rule-of-thumb" recommendations and the actual results will

depend heavily on the final layout.

Power and Ground Circuits

For 2 layer mixed signal design, it is important to isolate the

digital power and ground trace paths from the analog power

and ground trace paths. Star trace routing techniques (bring-

ing individual traces back to a central point rather than daisy

chaining traces together in a serial manner) can have a major

impact on low level signal performance. Star trace routing

refers to using individual traces to feed power and ground to

each circuit or even device. This technique will take require a

greater amount of design time but will not increase the final

price of the board. The only extra parts required may be some

jumpers.

Single-Point Power / Ground Connection

The analog power traces should be connected to the digital

traces through a single point (link). A "Pi-filter" can be helpful

in minimizing high frequency noise coupling between the ana-

log and digital sections. It is further recommended to place

digital and analog power traces over the corresponding digital

and analog ground traces to minimize noise coupling.

Placement of Digital and Analog Components

All digital components and high-speed digital signals traces

should be located as far away as possible from analog com-

ponents and circuit traces.

Avoiding Typical Design / Layout Problems

Avoid ground loops or running digital and analog traces par-

allel to each other (side-by-side) on the same PCB layer.

When traces must cross over each other do it at 90 degrees.

Running digital and analog traces at 90 degrees to each other

from the top to the bottom side as much as possible will min-

imize capacitive noise coupling and crosstalk.

www.national.com 12

LM4961

Schematic Board Layout

200940a4

FIGURE 4. Demo Board Schematic

13 www.national.com

LM4961

Demonstration Board Layout

200940c1

FIGURE 5. Recommended TS SE PCB Layout:

Top Silkscreen

200940c0

FIGURE 6. Recommended TS SE PCB Layout:

Top Layer

www.national.com 14

LM4961

200940c2

FIGURE 7. Recommended TS SE PCB Layout:

Bottom Layer

15 www.national.com

LM4961

Revision History

Rev Date Description

1.0 08/25/04 Initial WEB.

1.1 11/14/05 Replaced graphics 83, C4, and C5 with 01, 02, and 03), then WEB.

1.2 08/30/06 Added the TWUA row in the 4.2V Elect. Char table, then released the D/S to the WEB.

1.3 09/11/06 Added the “Selecting Value For Rc” in the Apps section, then released to the WEB.

www.national.com 16

LM4961

Physical Dimensions inches (millimeters) unless otherwise noted

LQ Package

Order Number LM4961LQ

NS Package Number LQA28A

17 www.national.com

LM4961

Notes

LM4961 Ceramic Speaker Driver

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